KR20220020788A - Ultrathin electromagnetic wave absorber - Google Patents

Abstract

The ability to completely absorb an electromagnetic wave by a material with a thickness much smaller than the resonant wavelength is an essential requirement for micro-miniature, flexible and lightweight electromagnetic wave devices. The present inventors combine a metal layer array with an inverse cross metal antenna through a dielectric separator and place a reflector at the bottom, to demonstrate a perfect absorber of metamaterial electromagnetic waves, especially microwaves, as thin as 1/1250 of the target wavelength. Compared to the conventional three-layer absorber, the modified design of the present application reconfigures a path of an induced current without changing a patch and a dielectric layer, thereby reversing the direction of a dipole magnetic field and significantly increasing the magnetic flux, to reduce a resonant frequency by a factor of up to 10. The ultra-thin film electromagnetic wave absorber of the present application includes a stack together with the reflector to realize the full absorption under critical coupling conditions, and the critical coupling condition is identified by monitoring the reflection phase shift across the resonant frequency. The present invention provides guidance for further reducing the thickness, and the present invention can be extended to other frequencies.

Description

Ultra-thin electromagnetic wave absorber {ULTRATHIN ELECTROMAGNETIC WAVE ABSORBER}

The present application relates to an ultra-thin electromagnetic wave absorber and a device including the same.

Sub-wavelength electromagnetic wave (EM wave) absorbers have attracted considerable attention across all regions of the electromagnetic spectrum, particularly in the MHz and GHz regions, which are practical in stealth technology, thermal sensing, sensors, and communications. important because of its importance. In particular, an electromagnetic wave absorber plays an important role because an autonomous vehicle or aeronautical system may malfunction if it does not sufficiently block a stray electromagnetic wave signal, thereby posing a serious risk to human safety. Accordingly, a high-performance electromagnetic wave absorber is an essential component of a complex electronic system. From a practical point of view, the absorber should not only exhibit 100% absorption efficiency, but also be lightweight and ultra-thin, providing mechanical flexibility, improving portability, and helping to improve the performance of the host device.

Reducing electromagnetic wave absorbers to extreme subwavelength thicknesses is an area that has been actively studied over the past few decades. In general, a carefully designed high-impedance structure or lossy dielectric, suspended or placed on a metal plane, is combined with a Salisbury screen and Dallenbach absorber with a host thickness of one-fourth the operating wavelength. Each can be represented by the same classic design. Considering the centimeter to metric scale wavelengths of GHz and MHz radiation, especially for modern electrical and telecommunication devices, the quarter wavelength thickness is large, resulting in a strong angle of incidence dependence. Recently, a significant reduction in thickness has been achieved with the advent of metamaterials or metasurfaces, composed of arrays of artificially designed sub-wavelength elements. Typically, the array is supported by a dielectric layer on a solid metal plane that prevents electromagnetic wave transmission and allows reflections to be canceled out by proper structuring of the elements. Landy et al. reported a “perfect” absorber, the first metamaterial to provide independently tunable effective permittivity and permeability through structural modification of split-ring-shaped components. did Complete absorption was achieved by impedance matching the structure to free space (ie, ε = μ). In the absence of reflection, incident electromagnetic waves can be dissipated within the lossy dielectric layer of the metamaterial, allowing the structure to remain as thin as λ ₀ /35 [where λ ₀ is the free space operating wavelength. wavelength)]. Since then, thinner split ring metamaterials exhibiting thicknesses of λ ₀ /40 and λ ₀ /75 have been reported.

As one of several types of elements used in metamaterials, patch antennas are simple and practical. Implemented decades ago by radio wave engineers, a patch antenna array on a dielectric layer supported by a metal plane allows the target frequency to remain fixed at large angles of incidence, and by its four-fold symmetry To maintain polarization independent. This design has been used for resonant electromagnetic wave absorbers featuring extreme sub-wavelength thicknesses. It was shown that closely spaced circular disks achieved an absorber thickness of λ ₀ /130.

An obvious strategy to achieve subwavelength absorber thickness is to redshift the resonance by broadening the patch antenna area to increase capacitance. However, this approach may cause problems for applications where the patch antenna size must be fixed or kept at sub-wavelengths. Another way to increase the capacitance is to decrease the thickness of the dielectric layer. However, this entails a reduction in inductance, which offsets the gain due to the increased capacitance, thus maintaining the insensitivity of the resonant frequency to dielectric thickness.

A striking feature of the design in fundamental resonance is the appearance of a dipole magnetic field between the patch and the bottom metal plane, perpendicular to the applied electric field and parallel to the plane. The magnetic field arises from antiparallel currents induced in the top and bottom planes and exhibits the ability to store magnetic energy that defines the associated inductance.

Y. Ra'di, C. R. Simovski, S. A. Tretyakov, Thin Perfect Absorbers for Electromagnetic Waves: Theory, Design, and Realizations. Physical Review Applied 3, 037001 (2015).

An object of the present application is to provide an ultra-thin electromagnetic wave absorber and a device including the same.

The present application provides an ultra-thin electromagnetic wave absorber using a meta-material that exhibits effective electromagnetic properties when the units are arranged with a certain period. A popular design for metamaterial absorbers is a simple and practical patch antenna. In order to achieve a sub-wavelength absorber, it is necessary to increase the capacitance of the patch antenna by enlarging the patch size. However, if the size of the patch antenna is enlarged, flexibility is reduced and there may be a problem in application of electronic devices. Another way to increase the capacitance is to reduce the thickness of the dielectric layer. However, this reduces the inductance and consequently the resonance frequency does not act sensitively to the thickness of the dielectric layer. Therefore, in the present application, an inverse structure is introduced to increase the inductance induced by the changed current movement path to achieve a sharp increase in the wavelength-to-thickness ratio. A metal ground plane for complete absorption is introduced into the combined patch and inverted antenna array to eliminate reflected waves due to wavelength interference. In addition, perfect absorption is achieved by finding an appropriate dielectric layer thickness range that satisfies the critical coupling condition for complete absorption.

However, the problems to be solved by the present application are not limited to the problems mentioned above, and other problems not mentioned will be clearly understood by those skilled in the art from the following description.

A first aspect of the present disclosure provides an ultra-thin electromagnetic wave absorber comprising a stack and a reflector, the stack comprising a metal layer, a dielectric layer and an inverse cross structure.

A second aspect of the present application provides a device including the ultra-thin electromagnetic wave absorber according to the first aspect.

The ultra-thin electromagnetic wave absorber according to the embodiments of the present application replaces a longitudinal dipolar magnetic field with an asymmetric pair of a transverse dipole magnetic field to increase the total inductance by 10 times or more, thereby significantly increasing the wavelength-to-thickness ratio Demonstrate that an increase can be achieved. This change introduces an appropriately tailored aperture in the metal plane that redefines inductance, devising antiparallel currents into in-plane current loops without modifying the thickness of the patch antenna or dielectric layer. can be realized by

Using ultra-thin electromagnetic wave absorbers according to embodiments herein, the inventors have found that absorption of at least about 80%, at least about 90%, at least about 95%, at least about 98%, or at least about 99%, particularly at least about 99.2% Efficiency and metamaterial absorbers with unprecedented sub-wavelength thicknesses as small as thicknesses from about λ ₀ /500 to about λ ₀ /2500, in particular λ ₀ /1250, have been experimentally demonstrated. Here, we set the stack thickness to be about 10 ³ times smaller than the resonance wavelength, using a three-layer stack comprising a metal layer, a dielectric layer, and a quadruple symmetric inverse metal cross structure. By providing an inverse structure cross arm, an intuitive form of coupling with respect to key aperture parameters, a boost in inductance resulting in a large resonant frequency reduction was described. In addition, in order to achieve complete absorption of electromagnetic waves, the stack was suspended on a metal plane to cancel reflected waves using wave interference. For a given intrinsic loss amount of the dielectric layer, the present inventors found a suitable dielectric layer thickness range that satisfies the critical coupling condition to induce perfect absorption. With extremely sub-wavelength thick stacks, the entire ultra-thin electromagnetic wave absorber including the stack and the reflector, especially the spacer between the metal ground, is much thinner than conventional patch antenna absorbers, and is insensitive to a wide range of angles of incidence. can keep

Figure 1 confirms the role of aperture in the design of the absorber of extreme sub-wavelength thickness of the present application. , an improved three-layer stack comprising a metal rod, a dielectric layer, and a metal plane, characterized by a 90° rotated inverted rod, and an inverted structure suspended on a metal ground plane A schematic diagram of a unit cell of an improved stack comprising rods is shown (where red and blue arrows indicate induced current and dipole magnetic field, respectively). The insets of FIGS. 1A and 1B show, respectively, a pink magnetic flux region, where the magnetic flux region was greatly increased with the introduction of the inverse structure rod, and the inset of FIG. 1C shows the cancellation of transmission and reflection under critical coupling conditions. indicates 1D is a graph of the calculated reflection spectra of the three-layer stack according to FIGS. 1A and 1B, respectively, and FIG. 1E is a graph of the presence of a metal ground plane at critical bonding conditions corresponding to FIGS. 1B and 1C, respectively. A graph of the calculated absorption spectrum of the improved stack.
Figure 2 confirms the resonant frequency dependence of the 3-layer-stack on the aperture parameter according to the embodiment of the present application, and Figure 2a shows the inverse structure cross aperture in the metal plane including the dielectric layer (thickness: 50 μm). A schematic diagram of a coupled patch antenna array, FIG. 2b is visualized at the junction of the patch/dielectric layer (top row) and the dielectric layer/inverse structure cross junction (bottom row), for various lengths of crosses (l); Resonant surface charge density distribution and surface current vector are shown (the electric field is incident and polarized perpendicularly along the x-axis). Figure 2c shows the resonant magnetic field intensity distributions and corresponding magnetic field vectors visualized in the central plane parallel to the electric field (top row) and perpendicular (bottom row) for various cross lengths, and Figure 2d shows the three ranges of An LC circuit model based on the distinct electric and magnetic field characteristics of l is shown. Fig. 2e is a graph showing the reflection spectrum as a function of simulated l in the finite element method superimposed with the resonant frequency predicted by the LC circuit model, and Fig. 2f is a graph showing the simulated transmission spectrum as a function of l.
3 is a diagram related to the fabrication of an ultra-thin three-layer stack using a metal ground plane and measurement of its properties. 3A is a schematic diagram of an ultra-thin full absorber comprising a three-layer metamaterial stack array on a metal ground plane, and FIG. 3B is a photograph of a return loss measurement configuration, wherein two horn antennas used are used to generate and receive electromagnetic signals; It is directed towards a stack that is firmly attached to the Cu wall. 3c and 3d show simulated absorption spectra and configuration for an insulating stack (patch/dielectric/inverse structure cross), a stack on a metal ground plane, and a stack without apertures, respectively, for a selected pore spacing; Ratio of simulated operating wavelength to total thickness as a function of dielectric layer thickness.
4 is a graph relating to the characteristics of critical coupling condition and angular dependence. Figure 4a is a simulated spectrum of the total reflection phase transition as a function of the loss tangent in the dielectric layer (the dashed circles indicate the loss tangent values for critical coupling and resonant frequencies. The thickness t _d of the dielectric layer is 25 μm; , void thickness ta _is 100 μm), FIG. 4b is the corresponding absorption spectrum of FIG. 4a for different loss tangent values, and FIG. 4c is the simulated return loss in absorber configurations with dielectric layer thicknesses of 25 μm, 50 μm and 125 μm. Spectral and tangent loss values to achieve critical coupling are shown (inset shows the corresponding absorption spectra for the three configurations). Figure 4d is the measured return loss spectrum for absorber configurations with dielectric thicknesses of 25 μm, 50 μm and 125 μm, and Figure 4e is the calculated absorption as a function of angle of incidence for absorbers with t _d = 25 μm and ta ₌ 100 μm. spectrum, and Fig. 4f is at different angles of incidence The measured return loss for the t _d = 25 μm configuration is shown.

Hereinafter, with reference to the accompanying drawings, embodiments and examples of the present invention will be described in detail so that those of ordinary skill in the art to which the present invention pertains can easily carry out. However, the present application may be embodied in several different forms and is not limited to the embodiments and examples described herein. And in order to clearly explain the present invention in the drawings, parts irrelevant to the description are omitted, and similar reference numerals are attached to similar parts throughout the specification.

Throughout this specification, when a part is "connected" with another part, this includes not only the case where it is "directly connected" but also the case where it is "electrically connected" with another element interposed therebetween. do.

Throughout this specification, when a member is said to be located “on” another member, this includes not only a case in which a member is in contact with another member but also a case in which another member is present between the two members.

Throughout this specification, when a part "includes" a certain element, it means that other elements may be further included, rather than excluding other elements, unless otherwise stated.

As used herein, the terms "about," "substantially," and the like are used in a sense at or close to the numerical value when the manufacturing and material tolerances inherent in the stated meaning are presented, and to aid in the understanding of the present application. It is used to prevent an unconscionable infringer from using the mentioned disclosure in an unreasonable way.

As used throughout this specification, the term “step of doing” or “step of” does not mean “step for”.

Throughout this specification, the term "combination(s) of these" included in the expression of the Markush form means one or more mixtures or combinations selected from the group consisting of the components described in the expression of the Markush form, It means to include one or more selected from the group consisting of the above components.

Throughout this specification, reference to “A and/or B” means “A or B, or A and B”.

Throughout this specification, "electromagnetic wave" means an electromagnetic wave having a frequency of about 1 MHz to about 400 THz.

Throughout this specification, the term "inverse structure cross" means that the cross section is formed as an aperture in a plane.

Hereinafter, embodiments of the present application have been described in detail, but the present application may not be limited thereto.

A first aspect of the present disclosure provides an ultra-thin electromagnetic wave absorber comprising a stack and a reflector, the stack comprising a metal layer, a dielectric layer and an inverse cross structure.

In one embodiment of the present application, an air space or a dielectric spacer may exist between the stack and the reflector, but may not be limited thereto.

In one embodiment of the present application, the reflector may be a metal plane, but may not be limited thereto.

In one embodiment of the present application, the metal layer and the inverse structure cross structure may include the same or different metals. The metal layer and the inverse structure cross structure are, each independently, copper, silver, gold, aluminum, molybdenum, zinc, lithium, nickel, brass, steel, platinum, palladium, tin, bronze, lead , titanium, iron, chromium, cobalt, iron alloys such as FeCrAl, and stainless steel, may be selected from the group consisting of, alloys thereof, and combinations thereof.

In one embodiment of the present application, the metal layer may have a rod, patch, rectangle, or square shape, but may not be limited thereto. In one embodiment of the present application, the metal layer may be a metal patch. In one embodiment of the present application, the metal layer may be a square-shaped patch.

In one embodiment of the present application, the inverted cross structure is a Greek cross, and may have a quadruple symmetric structure, but may not be limited thereto.

In one embodiment of the present application, the length of the inverted cross structure may be equal to or smaller than the length of the metal layer. In this case, the inductance is seen as two inductances with the magnetic flux divided, which can be thought of as two solenoid inductors with dipole magnetic fields in the same direction arranged in parallel. In one embodiment of the present application, the length of the inverted cross structure may be greater than the length of the metal layer. In this case, the magnetic flux area and/or inductance significantly increases, and the resonance frequency is significantly reduced compared to the case without the inverse structure cross structure. Here, the length of each of the arm portion and the metal layer of the inverse structure cross structure is based on the longest edge.

In one embodiment of the present application, the thickness of the ultra-thin electromagnetic wave absorber may be about λ ₀ /500 to about λ ₀ /2500, but may not be limited thereto. For example, the thickness of the ultra-thin electromagnetic wave absorber is about λ ₀ /500 to about λ ₀ /2500, about λ ₀ /500 to about λ ₀ /2000, about λ ₀ /500 to about λ ₀ /1500, about λ ₀ /1000 to about λ ₀ /2500, about λ ₀ /1000 to about λ ₀ /2000, about λ ₀ /1000 to about λ ₀ /1500, about λ ₀ /1500 to about λ ₀ /2500, about λ ₀ /1500 to about λ ₀ /2000, or from about λ ₀ /2000 to about λ ₀ /2500. In one embodiment of the present application, the thickness of the ultra-thin electromagnetic wave absorber may be about λ ₀ /500 to about λ ₀ /1500.

In one embodiment of the present application, the ultra-thin electromagnetic wave absorber may have a resonance frequency and/or absorption constant regardless of the incident angle of the electromagnetic wave, but may not be limited thereto.

In one embodiment of the present application, the thickness of the metal layer, the inverse structure cross structure and the reflector, each independently, may be about 5 μm to about 20 μm, but may not be limited thereto. For example, the thickness of the metal layer, the inverted cross structure, and the reflector may each independently be about 5 μm to about 20 μm, about 5 μm to about 15 μm, about 5 μm to about 10 μm, about 10 μm to about 20 μm, or about 15 μm to about 20 μm, but may not be limited thereto.

In one embodiment of the present application, the thickness of the dielectric layer may be about 10 μm to about 250 μm, but may not be limited thereto. For example, the thickness of the dielectric layer is from about 10 μm to about 250 μm, from about 10 μm to about 200 μm, from about 10 μm to about 150 μm, from about 10 μm to about 100 μm, from about 10 μm to about 90 μm, about 10 μm to about 80 μm, about 10 μm to about 70 μm, about 10 μm to about 60 μm, about 10 μm to about 50 μm, about 10 μm to about 40 μm, about 10 μm to about 30 μm, about 10 μm to about 20 μm, about 20 μm to about 250 μm, about 20 μm to about 200 μm, about 20 μm to about 150 μm, about 20 μm to about 100 μm, about 20 μm to about 90 μm, about 20 μm to about 80 μm, about 20 μm to about 70 μm, about 20 μm to about 60 μm, about 20 μm to about 50 μm, about 20 μm to about 40 μm, about 20 μm to about 30 μm, about 30 μm to about 250 μm, about 30 μm to about 200 μm, about 30 μm to about 150 μm, about 30 μm to about 100 μm, about 30 μm to about 90 μm, about 30 μm to about 80 μm, about 30 μm to about 70 μm, about 30 μm to about 60 μm, about 30 μm to about 50 μm, about 30 μm to about 40 μm, about 40 μm to about 250 μm, about 40 μm to about 200 μm, about 40 μm to about 150 μm, about 40 μm to about 100 μm, about 40 μm to about 90 μm, about 40 μm to about 80 μm, about 40 μm to about 70 μm, about 40 μm to about 60 μm, about 40 μm to about 50 μm, about 50 μm to about 250 μm, about 50 μm to about 200 μm, about 50 μm to about 150 μm, about 50 μm to about 100 μm, about 50 μm to about 90 μm, about 50 μm to about 80 μm, about 50 μm to about 70 μm, about 50 μm to about 60 μm, about 60 μm to about 250 μm, about 60 μm to about 200 μm, about 60 μm to about 150 μm, about 60 μm to about 100 μm, about 60 μm to about 90 μm, about 60 μm to about 80 μm, about 60 μm to about 70 μm, about 70 μm to about 250 μm, about 70 μm to about 200 μm, about 70 μm to about 150 μm, about 70 μm to about 100 μm, about 70 μm to about 90 μm, about 70 μm to about 80 μm, about 80 μm to about 250 μm, about 80 μm to about 200 μm, about 80 μm to about 150 μm, about 80 μm to about 100 μm, about 80 μm to about 90 μm, about 90 μm to about 250 μm, about 90 μm to about 200 μm, about 90 μm to about 150 μm, about 90 μm to about 100 μm, about 100 μm to about 250 μm, about 100 μm to about 200 μm, about 100 μm to about 150 μm, about 150 μm to about 250 μm, about 150 μm to about 200 μm, or about 150 μm to about 250 μm, but may not be limited thereto . In one embodiment of the present application, the thickness of the dielectric layer may be about 15 μm to about 200 μm.

In one embodiment of the present application, the resonance frequency may vary depending on the thickness of the dielectric layer, but may not be limited thereto. In one embodiment of the present application, as the thickness of the dielectric layer increases, the resonance frequency may increase. For example, using dielectric layers each having a thickness of about 25 μm, about 50 μm, and about 125 μm, the resonant frequencies may be about 1.5 GHz, about 2.1 GHz, and about 3 GHz, respectively.

In one embodiment of the present application, the thickness of the gap or the dielectric spacer may be about 50 μm to about 500 μm, but may not be limited thereto. For example, the thickness of the pores is about 50 μm to about 500 μm, about 50 μm to about 400 μm, about 50 μm to about 300 μm, about 50 μm to about 200 μm, about 50 μm to about 100 μm, about 100 μm to about 500 μm, about 100 μm to about 400 μm, about 100 μm to about 300 μm, about 100 μm to about 200 μm, about 200 μm to about 500 μm, about 200 μm to about 400 μm, about 200 μm to about 300 μm, about 300 μm to about 500 μm, about 300 μm to about 400 μm, or about 400 μm to about 500 μm, but may not be limited thereto.

In one embodiment of the present application, in the ultra-thin electromagnetic wave absorber, one or more units including the stack and the reflector may be arranged at a period of about 6 mm to about 50 mm, but may not be limited thereto. For example, in the ultra-thin electromagnetic wave absorber, the unit including the stack and the reflector is about 6 mm to about 50 mm, about 6 mm to about 40 mm, about 6 mm to about 30 mm, about 6 mm to about 20 mm , about 10 mm to about 50 mm, about 10 mm to about 40 mm, about 10 mm to about 30 mm, about 10 mm to about 20 mm, about 15 mm to about 50 mm, about 15 mm to about 40 mm, about One or more may be arranged in a period of 15 mm to about 30 mm, or about 15 mm to about 20 mm, but may not be limited thereto. In one embodiment of the present application, in the ultra-thin electromagnetic wave absorber, one or more units including the stack and the reflector may be arranged in a period of about 15 mm to about 20 mm.

In one embodiment of the present application, the units may be arranged in an array form.

In one embodiment of the present application, when the specification of the metal layer, the specification of the inverse structure cross structure, and the period are collectively increased or decreased by the same multiple, the resonance frequency may be increased or decreased by the same multiple there is. That is, even if the resonant frequency is enlarged or reduced according to the specification of the ultra-thin electromagnetic wave absorber according to the present application, hunting may not occur and it may operate properly.

A second aspect of the present application provides a device including the ultra-thin electromagnetic wave absorber according to the first aspect.

Although a detailed description of overlapping parts with the first aspect of the present application is omitted, the contents described for the first aspect of the present application may be equally applied even if the description thereof is omitted in the second aspect of the present application.

In one embodiment of the present application, the device may be used without limitation not only in the current technical field related to electromagnetic wave absorption, but also in the future technical field that can be utilized in accordance with the present application.

In one embodiment of the present application, the device may be used in an electronic device, stealth, sensor, or communication device, but may not be limited thereto.

Hereinafter, the present application will be described in more detail using examples, but the following examples are only illustrative to help the understanding of the present application, and the content of the present application is not limited to the following examples.

[Example]

에 따라 공명 주파수를 정의한다. 상기 금속 막대, 유전체 및 금속 평면은 종래의 2-면 커패시터 (capacitor)를 구성하기 때문에, 금속 막대를 확대하여 C를 증가시킴으로써 공명 주파수를 최소화할 수 있다.1A-1C show a series of simplified schemes of key steps to achieve full absorption at extreme sub-wavelength thicknesses. First, as shown in Figure 1a, considering the unit cell photograph of a metal strip on a metal plane separated by a thin dielectric layer, it was confirmed what structural modifications were needed to reduce the resonance frequency by increasing the wavelength-to-thickness ratio. . The long length of the metal rod, the short length of the metal rod and the thickness of the dielectric layer in this embodiment are 40%, 13% and 0.17% of the period, respectively. At the fundamental resonant frequency, when an electromagnetic wave is incident at right angles to an absorber with an electric field polarized along a metal rod, an electric current is induced in the metal plane and the rod antiparallel to each other. Depending on the thickness of the dielectric layer and the amount of loss, the three-layer system can partially or completely block the incident wave because transmission is blocked by the metal plane and reflection is partially or completely canceled out by the electromagnetic wave generated from the induced current. can be completely absorbed. Naturally, this current sets up a dipole magnetic field in the dielectric that is normal to the plane containing the current, and produces an associated inductance L proportional to the magnetic flux, indicated by the pink area in Figure 1a. . Likewise, the dipole electric field between the metal rod and the metal plane sets up a capacitance, C, proportional to the area of the patch antenna. L and C are

Define the resonant frequency according to Since the metal rod, the dielectric, and the metal plane constitute a conventional two-sided capacitor, the resonant frequency can be minimized by enlarging the metal rod to increase C.

However, the aim of the present invention here is to reduce the resonant frequency of the absorber without changing the rod size. The present inventors hypothesized that a rod-shaped aperture of the same shape and size as that of the metal rod was inserted into the lower metal plane perpendicular to the long axis of the metal rod. When the electromagnetic wave at the resonant frequency is incident perpendicularly to the arrangement, the current in the bottom metal plane should split and flow around both ends of the inverted structure rod, as shown in Fig. 1b. In this process, an antiparallel dipole magnetic field perpendicular to the plane containing the current is created at the tip of the inverted rod, whereas the longitudinal dipole magnetic field is no longer sustained as the original current path is broken. Thus, the magnetic flux proportional to the area over which the magnetic field is maintained is at least ten times greater than that of a conventional patch antenna arrangement, which is the conventional configuration defined by rod dimensions, whereas the configuration herein is sub-wavelength dielectric thickness. because it is defined. This is interpreted to be accompanied by an increase in inductance that is much larger than a decrease in capacitance caused by inserting the aperture. Also, since the two dipole magnetic fields are antiparallel to each other, the mutual inductance further enhances the total inductance. This is a study modeling complex coupled nanophotonic elements as a simple arrangement of dipoles, evidently that a pair of transverse antiparallel magnetic dipoles has a lower energy than unpaired magnetic dipoles. consistent with the insights obtained from

The change in resonant frequency due to aperture insertion is shown in Figure 1d, where the reflection is at a scale-invariant frequency (ie, the ratio of period to wavelength) [(scale-invariant frequency (ie, the ratio of period to wavelength)] Plotted, highlighting the applicability of the design at various spatial scales: Resonance, characterized by a sharp reduction in reflection, is significantly more than 6 times in frequency due to the introduction of an inverted structure rod without any change in the patch antenna and dielectric layer. appeared to decrease.

As a result of forming the inverted rod on the metal plane, the transmission is no longer zero because electromagnetic waves can leak through the aperture. This prevents the three-layer stack from achieving full absorption. A solution to the above problem is to place a metal ground plane with sufficient spacing under the stack such that the near field coupling between the stack and the metal plane is negligible, as shown in Fig. 1c. The metallic ground plane constructs multiple reflected EM waves in the spacer layer that completely blocks transmission and the overlap interferes with and cancels total reflection, provided that critical coupling is met. This condition can be realized by incorporating an appropriate dielectric loss in the stack for a given dielectric layer thickness, which matches the dielectric parameters with and without a metal ground plane separated from the stack by 6 times the dielectric layer thickness in a concentrated frequency range. 1e, which shows the absorption spectrum of the stack fabricated by Without the metal ground plane, the stack does not absorb well due to power leakage through the aperture. However, with the grounding, absorption reaches unity. It is clear that the spacer (void) thickness increases the total thickness of the setup, but due to the extreme sub-wavelength thickness of the stack, the ratio of wavelength to total thickness is still 1.4 times greater than that of the original absorber in Figure 1a. Big.

The absorber described above is polarization dependent. Therefore, to ensure polarization independence using the above concept, the design was extended to a four-fold symmetric patch and aperture configuration. The rods were replaced with square patches, and the apertures were modified with inverted crosses elongated than the patch dimensions. Figure 2a shows a schematic diagram of a stack array represented by geometric parameters. In order to clearly elucidate the role of the aperture on the resonant frequency in the modified configuration, we observed the induced charge, current and magnetic field in the relevant plane while gradually increasing the length of the inverse cross-arm from 0 mm to 14 mm. did Simulations were performed using a finite element method solver (COMSOL, Inc), and the dimensions used were p=15 mm, w=5.8 mm, d=2 mm, l=14 mm, t _m =12 μm and t _d =50 μm. For the metal, a copper model was used. The index of the dielectric layer was initially set to a real number (n = 1.82) in order to keep the analysis focused on the change in the resonance frequency. The dielectric loss derived from the refractive index was then considered in the study of complete absorption. Unless otherwise specified, electromagnetic waves are incident perpendicularly to the absorber and are polarized along the x-direction.

First, as shown in Fig. 2b, we highlight the main characteristics of the induced surface charge density and current of the patch (top) and aperture (bottom) for various l (length of inverse cross-arms). The induced surface charge densities in the top and bottom planes are mirror images of each other. This clearly shows the coupling properties of the top and bottom planes, and in addition, it means that changing the current and charge dynamics in one plane affects the dynamics in the other plane. This is a powerful option when you want to control the current and charge patterns of a three-layer stack without changing the physical dimensions of the patch. When an aperture of l=2 mm (ie, a square aperture) is introduced in the metal plane, this capability can be observed more clearly (Fig. 2b). Although there is no physical aperture in the patch, the current and charge distribution of the patch behaves as if the aperture were present. Likewise, for all values of l, the induced charges in both planes are spatially confined to the outer and inner boundaries of the patch and inverse structure crosses, respectively. This charge represents a path-forbidden current flow outside the patch and inside the apertures in each plane.

In order to explain the change of the resonance frequency according to the increase of L, a simple circuit model was developed. From the plots of current, charge and associated magnetic field patterns, illustrated in FIGS. 2B and 2C , three distinct trends can be identified, which warrant three circuit models. The first model corresponds to the case without aperture (l=0 mm), and represents a parallel LC circuit as depicted in Fig. 2d, where the possible outputs are mediated by the exchange of magnetic and electrical energy. only reflection The magnetic field can be visualized in a central plane parallel and perpendicular to the patch surface, as shown in the top and bottom panels of Fig. 2c, respectively. As expected, a nearly uniform magnetic field is created within the dielectric layer along the y-direction flowing perpendicular to the induced current. Likewise, a strong electric field of alternating polarity is generated across the patch width along the z-direction between the patch and the metal plane. According to this observation, the indentance L and the capacitance C can be defined from two parallel square plates of width w, as in the following equation.

(1)

(One)

(2)

(2)

where a is the adjustment factor that remains fixed for the non-uniform magnetic field and all l. ε ₀ and μ ₀ are the absolute permittivity and the magnetic permeability in vacuum, respectively.

When an aperture is introduced, the charges, currents, and magnetic fields in the top and bottom planes are rearranged due to exclusion of the charges, currents, and magnetic fields inside the aperture. The resulting behavior remains similar even for l < w, where the inverse cross arm is shorter than the patch size. It is clear that C decreases as the area supporting the induced charge in both planes decreases due to the enlarged aperture. On the other hand, L, as shown in Fig. 2c, increases by dividing the original magnetic flux into two regions separated by an inverse structure cross. The split flux exhibits two inductances, which can be approximated by two solenoid-type inductors arranged in parallel with a symmetric longitudinal dipole magnetic field, as shown in the middle panel of Fig. 2d. Representations of circuit elements are detailed in the supporting information. An important characteristic shown in Fig. 2c is that the magnetic field is strengthened as l increases up to w at the increased surface current density flowing through the narrowing passage, as shown in Fig. 2b. Our model captures this phenomenon, since a larger l corresponds to a reduced solenoid length that enhances the magnetic field of each inductor. In addition, the mutual inductance M, which deals with the interaction between the two inductors, must also be considered. It can be readily seen that the interaction of two symmetric longitudinal magnetic dipoles results in a more energetically stable state than the interaction of a lone magnetic dipole. Thus, despite the parallel configuration of the two inductors reducing the resulting inductance in half, the contribution of M ultimately enhances the total L, effectively reducing the resonant frequency.

For l > w, where the arms of the inverted cross are longer than the patch width, the electromagnetic properties undergo a dramatic transition characterized by several key changes. First, since the region capable of holding the induced charge does not change, C remains stationary (see Fig. 2b). On the other hand, L increases in a completely different way than in the case of l < w. The two dipole magnetic field vectors are switched in the z-direction within the two open cross arms outside the patch in alignment along the y-direction of the dielectric layer. As shown in Figure 2b, this modified magnetic field is generated because the current path is relocated in the aperture plane so that the current flows around the two open cross arms. As a result, compared to the case of l=0 mm, the area of the magnetic flux is enlarged 57 times when l=14 mm, which is a dimension including the sub-wavelength dielectric layer thickness when l=14 mm, whereas when l=0 mm This is because it has the same size as the patch. As a first approximation, L was obtained for each inductor, assuming that the currents were generated in four long wires connected in a rectangular loop, defined as the boundaries of the open cross arms and whose wire diameter was equal to the metal thickness t _m . Despite its simplicity, the model retains its main feature, which is the apex of the magnetic field at the edge of the open cross arm.

Referring to Figure 2c, the transverse magnetic fields of the two open cross arms are antiparallel to each other due to the mirror symmetry of the current flow about the x-axis. The above scenario represents the metal rod model shown in Fig. 1b. Therefore, the resonance of a three-layer stack with l > w can be expected to be lower in energy than without aperture (l=0 mm). For l < w, the two coupled magnetic dipoles can be modeled through M shown in the supporting information. As l approaches 14 mm, the mutual inductance M' of the adjacent cells must also be taken into account, since the opposite two ends of the adjacent open cross arms are closer together, which improves the interaction. The antiparallel configuration of the transverse magnetic dipoles of M and M' certainly contributes to a further increase in the total L, but the contribution is weak as the magnitude is much smaller than the individual inductances of the apertures. Indeed, the relative contributions of M and M' to the total L calculated in our LC circuit model are only 1.5% and 8.5%, respectively, at l=14 mm, whereas M for l=2 mm is 64.3%. This indicates that the magnetic dipole coupling for l > w is much weaker than for l < w. The difference in the contribution of M between the cases l < w and l > w can be attributed to the unique coupling characteristics of the two inductors. It is known that longitudinally coupled dipoles (l < w) result in greater energy degradation than transversely coupled dipoles (l > w) when separated by the same distance. In the present case, the separation distance between the magnetic dipoles of M is much larger at l > w than at l < w, which means that the magnetic dipole coupling is relatively weak at l < w.

Ultimately, the total L calculated from the circuit model at l=14 mm increased by 78 times compared to the case without the aperture, reducing the resonant frequency by 7.5 times while keeping the patch and dielectric layer intact. In Fig. 2e, the resonant frequencies obtained from the three LC circuit models of the present application are compared with the simulated reflection spectra with the limiting element method at several l. Despite using simple assumptions to build the circuit model, good agreement was found between the predictive model and the simulation. Here, the resonant frequencies at l=0 and 14 mm predicted by the model were 14.2 GHz and 2.5 GHz, respectively, and the simulated results were 14.2 GHz and 1.9 GHz. The L and C models that more accurately handle the field (magnetic field) properties are expected to improve the agreement. The simple model construction herein is that it can accurately account for the distinct resonant frequency dependence on l, characterized by a sharp frequency change at l<w and a relatively smaller frequency change at l>w. According to the circuit model above, these two distinct trends can be explained by understanding the dominant mechanism behind resonant shift, magnetic dipole coupling for l < w and aperture size dependent flux change for l > w. This insight provides qualitative guidance for tuning resonance to l without altering the patch and dielectric layer.

Resonant redshift due to aperture insertion causes power dissipation through transmission. To observe this more clearly, the transmission through the stack as a function of l was plotted in FIG. 2f . The plot shows that the maximum transmittance increases with l, a result of the apparent enlarged aperture insertion. Ultimately at l=14 mm, the transmittance at resonance reaches unity and the reflectance becomes zero. That is, the stack becomes a bandpass filter. This conversion prevents the stack from functioning as a full absorber, even when losses are incorporated in the dielectric, since the radiative loss rate cannot match the absorption loss rate.

In order to use the passband filter as a perfect absorber, an adjustment must be made to restore the balance between the radiative and absorption loss factors. As a first step, the transmission of electromagnetic waves through the aperture must be prevented. This scenario can be achieved by suspending the stack on a metal ground plane, as shown in Figure 3a. As in the metal rod model described above, metal grounding enables non-transmission (zero transmission), and selecting the appropriate dielectric parameters configures the system for non-reflection (zero reflection).

This capability was demonstrated by fabricating a three-layer stack with l = 14 mm and the geometric parameters used in the simulations herein above. The stack was attached to the Cu wall, and two horns were used to excite and sense electromagnetic waves of 0.5 GHz to 6 GHz that were almost normally incident, and absorption measurements were performed. The distance between the horn and the Cu wall was 1 m. The signal from the transmitting horn to the receiving horn was measured with an agilent Network Analyzer (E8364B) in an anechoic chamber ( FIG. 3B ).

The stack placement in front of the Cu wall represents our approach in FIG. 3A to counteract transmission. When the stack was attached to the Cu wall, the sides of the stack were secured to the wall using adhesive leaving the central region of the sheet slightly floating in the air. The air gap between the sheet and the Cu wall was measured to be 100 μm to 150 μm. The exact gap spacing varies from point to point, but since this difference constitutes a negligible portion of the target wavelength, it does not affect the resonant frequency location or absorption characteristics. In fact, the phase delays due to the two spacings of 100 μm and 150 μm (i.e., β=nk _d ) are only 0.0063 and 0.0094, which are too small to significantly affect the resonance and absorption properties. In order to explain this more clearly, as shown in Fig. 3c, absorption spectra for selected pore spacings of up to 500 μm were simulated. As expected, the resonant frequency and absorption did not change significantly over a wide part of the considered spacer thickness range from 100 μm to 500 μm. However, the large deviation, characterized by increased blueshift and decreased absorption with decreasing spacer thickness, is 100 μm, due to enhanced near-field interactions between stack and plane accompanied by mismatch between absorption and radiation loss rates. becomes clear below.

Even if air gaps are present, the overall construction can be several orders of magnitude thinner than a conventional three-layer patch antenna absorber for the same resonant frequency. 3D illustrates this concept, wherein the ratio of the operating wavelength to the total thickness of the conventional three-layer absorber and the void-included construction of the present application is plotted for constituent dielectric layer thicknesses from 15 μm to 200 μm. Here, the total thickness includes the thickness of dielectric layers, voids, and all metal planes, where it is assumed that the metal ground plane shares the same thickness (t _m ) with the other metal planes. For reference, the results for an insulating stack of l=14 mm acting as a passband filter are also presented. The above results represent the upper limit of the wavelength to total thickness ratio for the setup herein, with values approaching 10 ⁴ for the thinnest t _d of 15 μm considered. In addition, the plot shows a greater rate of increase in the wavelength to total thickness ratio as a function of the shrinkage t _d of the insulating stack than the conventional three-layer absorber. The above effect occurs because the resonance frequency of the former depends on t _d whereas the resonance frequency of the latter shows no t _d dependence. For l=0 mm, the product of the corresponding capacitance and inductance defined from equations (1) and (2) cancels the t _d dependence, whereas for l=14 mm the product of the capacitance and inductance is independent of t _d . Therefore, it can be confirmed that the inverse structure t _d dependency performed by the capacitance is maintained. Thus, as t _d decreases and the operating wavelength increases, the ratio of extreme wavelength to total thickness can increase by a factor of 100 or more for a passband filter.

Because of this extreme ratio, adding a 100 μm thick air gap and a metal ground plane to the stack becomes several times thinner than a conventional patch antenna for the same operating wavelength in the range where the total thickness of the set up is considered. Sub-wavelengths can be kept deep. For example, a setup comprising voids can be obtained at selected t _d values of 25 μm, 50 μm, and 125 μm, compared to the wavelength-to-thickness ratios 434, 286 and 143 of conventional patch antenna absorbers, 1246, 798, and an excellent ratio of 390. This ratio can be further improved if the voids are replaced with a dielectric that can be an adhesive that adheres to the conductive surface.

The absorption properties of the set up herein can be analyzed using wave interference theory, which provides an intuitive explanation of the conditions required to achieve complete absorption (24-26). According to the above theory, the reflection of our setup can be explained as the resulting wave interference between the reflection outside the stack and the superposition of multiple reflections generated within the stack and the metal ground plane. If there is no strong short-range coupling between the stack and the metal ground plane, the total reflection coefficient can be expressed as Equation 3 below.

(3)

(3)

은 공기-스택 계면에서의 복합 반사 및 투과 (transmission) 계수이다. 상기 계수들은 시뮬레이션으로부터 수득한 S 매트릭스 (matrix)로부터 수득하였다. 여기서, 상기 스택은 극한 서브 파장 두께로 인해 균일한 평면으로 처리된다.In Equation 3 above, the first term describes the reflection of the stack, and the second term captures the effect of multiple reflections in the void region set by the electromagnetic wave bouncing between the stack and the metal ground plane. Here, β is a propagation phase delay,

is the complex reflection and transmission coefficients at the air-stack interface. The coefficients were obtained from the S matrix obtained from the simulation. Here, the stack is treated as a uniform plane due to its extreme sub-wavelength thickness.

Using Equation 3 above, the reflection coefficient and transmission coefficient were adjusted through the loss and thickness of the dielectric layer to obtain conditions for complete absorption at a fixed pore thickness. In conventional metamaterial perfect absorbers, these two parameters often control the balance between radiation loss and absorption loss to switch the absorber into different operating modes. In particular, if the radiative loss rate is greater than, less than, or equal to the absorption loss rate, the system may be underdamped, overdamped, or completely absorbed, respectively. The condition in which the two loss rates are equal is also called critical coupling, and can be easily identified by analyzing the phase shift of total reflection near the resonance frequency. The reflection phase shift can cover the entire 2π range over the resonant frequency in the under-attenuated state, whereas it is limited to a range less than π in the over-attenuated state. These two different regions are shown in Figure 4a, which shows the calculated reflection phase shift versus loss tangent (ie, dielectric loss) for a given frequency range. Here, the thinnest experimental setup was modeled by setting the dielectric layer and pore thickness to 25 μm and 100 μm, respectively. The complex reflection and transmission coefficients of the stack for various loss tangent values were searched for in the simulation, and then used in equation (3) to obtain the phase shift of the total reflection in the setup. A critical coupling condition occurs when there is a phase transition between the under-attenuation region and the over-attenuation region located at a loss tangent value of 0.0253 in Fig. 4a. Absorption spectra at various loss tangent values shown in Figure 4b confirm that the setup exhibits 100% absorption at the specified loss tangent values, while at other loss tangent values of 0 and 0.053, attenuated and over-attenuated, respectively. did.

The above analysis was repeated for the other two stacks made with polyimide film thicknesses of 50 μm and 125 μm to determine the loss tangent values required to reach critical bonding. Full absorption was shown to occur at loss tangent values of 0.0386 and 0.0662, respectively, indicating that thicker dielectric layers require greater losses to reach critical coupling. Figure 4c shows the resulting return loss spectra of the three stacks, 100 μm away from the metal ground under critical coupling conditions. The resonant frequencies for the three stacks were 1.5 GHz, 2.1 GHz and 3.0 GHz, in order of increasing dielectric layer thickness, showing a blueshift of the thicker dielectric layer due to the reduced capacitance. The resonant frequency of the thinnest stack was reduced by about a factor of 10 compared to the stack without the aperture. A return loss of less than about 35 dB was calculated, which corresponds to about 100% absorption for all three setups as shown in the inset.

The actual return loss measurement is shown in Fig. 4d. The resonant frequencies of 1.49 GHz, 2.01 GHz, and 3.06 GHz for the three stacks were in good agreement with the prediction of the wave interference theory, confirming that the stack and the metal ground were not strongly coupled. On the other hand, the measured return loss value was different from the calculated value. The return losses of -20.89 dB, -24.1 dB and -27.34 dB, corresponding to near-complete absorption of 99.19%, 99.61% and 99.82%, were obtained from the configuration using 25 μm, 50 μm and 125 μm thick dielectric layers, respectively. It was recorded. Since the resonant frequency agrees well with the calculated results, indicating that the experimental dimensions are close to the target parameters, the consistency of absorption between the three setups shows that each of the three dielectric components exhibits slightly different complex refractive indices. This suggests that it arises from the different process conditions used to make the three polyimide films. As shown in Fig. 4b, it can be inferred that the effective loss tangent value of each of the three films should be between 0.0253 and 0.0662, which corresponds to a composite refractive index between n=1.82 + i0.02 and n=1.82 + i0.06. is a reasonable value for polymers in the microwave range.

The total thickness of the three setups is the sum of the measured thicknesses of all components including metal, dielectric and void layers. For the metal ground, which is the Cu wall here, a thickness t _m = 12 μm was used. Using 100 μm thick spacers, the resulting wavelength to total thickness ratios were 1250, 802 and 375, respectively, for the setup with 25 μm, 50 μm and 125 μm thick dielectric layers. Even with 150 μm thick pores, a 25 μm thick dielectric layer can exhibit a ratio of 947. Here, 1250 is the largest ratio reported to date, but it is possible to further increase this ratio by using a thinner Cu layer. This is because Cu acts as a perfect conductor in the microwave range, replacing the air spacer with a dielectric spacer, reducing the dielectric layer thickness and optimizing the aperture design of the stack, as shown in Fig. 3d.

Indeed, this setup can be used as a stand-alone absorber sheet by employing a dielectric spacer attached to a thin Cu film instead of the currently used void and Cu wall configuration.

As shown in Fig. 4e, the incident angle dependence of the l=14 mm stack with a dielectric thickness of 25 μm and a spacer thickness of 100 μm under the critical coupling was investigated by calculating the absorption spectrum in the range of incidence angles of up to 70 degrees. The plot shows that the configuration maintains the resonant frequency and full absorption in the angular range considered, indicating that the absorber should be independent of the angle of incidence up to high angles. To confirm the above function, specular reflection loss was measured from the setup at various angles of incidence. Measurements were performed in an electromagnetic shielding environment, with a transmitting horn positioned at 10 degrees, 30 degrees, 50 degrees and 70 degrees from normal incidence angles, and a detecting horn opposite the receiving end. The distance from the horn to the setup was fixed at 5 m. Figure 4f shows the result that the resonant frequency remains almost constant up to 70 degrees, as predicted by the calculations. However, contrary to the calculated results, it was found that the resonance intensity decreased at a larger angle with an increase in noise and stray peaks in the spectrum. This effect is presumably due to the divergence of the microwave beam propagating between the two horns, defocusing the beam in such a way that an external signal outside the target is picked up through the beam path. With improved electromagnetic wave collimation, a larger target area and a better shielding environment allow the return loss to reach the predicted value at high angles.

In summary, we demonstrated a metamaterial perfect absorber with an ultimate sub-wavelength thickness that is up to 10 ³ times smaller than the operating wavelength. The absorber consists of a three-layer stack consisting of a patch antenna over metal ground, a dielectric layer and a metal aperture array. The resonant frequency of the stack can be significantly reduced by inserting a cross-shaped aperture in the metal plane located below the patch antenna that redefines the induced current path, without modifying the dielectric layer and patch. The simple modification is to increase the inductance of the stack by a factor of 10 to ¹⁰ compared to the inductance magnitude of a conventional patch antenna absorber, and increase the single dipolar magnetic field to an inverse cross structure for the relative patch size. It further transforms into a coupled pair of dipole magnetic fields with coupling properties determined by the length of the arm. Here, we demonstrate that this effect results in a significant reduction in the resonant frequency, rendering the stack deep in thickness to sub-wavelengths. To counteract transmission through the aperture, a metal ground is placed below the stack, and the phase shift of total reflection versus resonant frequency is analyzed to target the exact loss tangent and thickness of the constituent dielectric layers identified. , and the reflection was set to 0. Applying this concept, the absorber of the present invention prepared with different dielectric thicknesses showed an absorption efficiency of 99% or more, where the thinnest absorber sheet had a wavelength to total thickness ratio of 1250, which was the thinnest so far. It is a complete electromagnetic wave absorber. In addition, the above results suggest that, besides structural and material changes, there is room for further thickness reduction with appropriate modifications to the patch and aperture design. The guidance discussed so far makes it possible to realize unprecedented thin, flexible and lightweight electromagnetic wave absorbers for use in complex electronic and communications systems.

The foregoing description of the present application is for illustration, and those of ordinary skill in the art to which the present application pertains will understand that it can be easily modified into other specific forms without changing the technical spirit or essential features of the present application. Therefore, it should be understood that the embodiments described above are illustrative in all respects and not restrictive. For example, each component described as a single type may be implemented in a distributed manner, and likewise components described as distributed may also be implemented in a combined form.

The scope of the present application is indicated by the following claims rather than the above detailed description, and all changes or modifications derived from the meaning and scope of the claims, and their equivalents should be construed as being included in the scope of the present application. .

Claims (19)

An ultra-thin electromagnetic wave absorber comprising a stack and a reflector, comprising:
wherein the stack comprises a metal layer, a dielectric layer and an inverted cross structure.
The method of claim 1,
An ultra-thin electromagnetic wave absorber, wherein a void or dielectric spacer exists between the stack and the reflector.
The method of claim 1,
The reflector will be a metal plane, ultra-thin electromagnetic wave absorber.
The method of claim 1,
The metal layer and the inverse structure cross structure will include the same or different metal, ultra-thin electromagnetic wave absorber.
The method of claim 1,
The metal layer has the shape of a rod, a patch, a rectangle, or a square, an ultra-thin electromagnetic wave absorber.
The method of claim 1,
The length of the reverse structure cross structure is equal to or less than the length of the metal layer, the ultra-thin electromagnetic wave absorber.
The method of claim 1,
The length of the reverse structure cross structure is greater than the length of the metal layer, the ultra-thin electromagnetic wave absorber.
The method of claim 1,
The thickness of the ultra-thin electromagnetic wave absorber is λ ₀ /500 to λ ₀ /2500, the ultra-thin electromagnetic wave absorber.
The method of claim 1,
The thickness of the ultra-thin electromagnetic wave absorber is λ ₀ /500 to λ ₀ /1500, the ultra-thin electromagnetic wave absorber.
The method of claim 1,
In the ultra-thin electromagnetic wave absorber, the resonant frequency and/or absorbance are constantly maintained regardless of the incident angle of the electromagnetic wave.
The method of claim 1,
The thickness of the metal layer, the inverse structure cross structure and the reflector, each independently, is 5 μm to 20 μm, the ultra-thin electromagnetic wave absorber.
The method of claim 1,
The thickness of the dielectric layer will be 10 μm to 250 μm, an ultra-thin electromagnetic wave absorber.
The method of claim 1,
The resonant frequency will vary depending on the thickness of the dielectric layer, the ultra-thin electromagnetic wave absorber.
3. The method of claim 2,
The thickness of the gap or the dielectric spacer will be 50 μm to 500 μm, an ultra-thin electromagnetic wave absorber.
The method of claim 1,
In the ultra-thin electromagnetic wave absorber, one or more units including the stack and the reflector are arranged with a period of 6 mm to 50 mm.
16. The method of claim 15,
The unit is to be arranged in an array form, an ultra-thin electromagnetic wave absorber.
16. The method of claim 15,
When the specification of the metal layer, the specification of the inverse structure cross structure, and the period are collectively increased or decreased by the same multiple, the resonance frequency increases or decreases by the same multiple, the ultra-thin electromagnetic wave absorber.
18. A device comprising an ultra-thin electromagnetic wave absorber according to any one of claims 1 to 17.
19. The method of claim 18,
The device of claim 1, wherein the device is used in an electronic, stealth, sensor, or communications appliance.

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