KR20210083146A - Method and Device for Controlling Inverter of Interior Permanent Magnet Synchronous Motor - Google Patents

Abstract

Disclosed are a device for controlling an inverter of an embedded permanent magnet synchronous motor, capable of ameliorating a long over-modulation time and low voltage utilization rate of weak magnetic flux control that adjusts a d-axis reference current in a negative direction, and an operation method thereof. According to one aspect of the present invention, the device for controlling an inverter, in a device for controlling an inverter of a synchronous motor, includes: a current controller which generates a reference voltage according to a dq-axis reference current; a modulator which generates an over-modulation signal and modulating the reference voltage over a hexagon of a space voltage vector diagram when the speed of the synchronous motor reaches a first reference speed; a d-axis current compensator which adds a compensating reference current to a d-axis reference current of the dq-axis reference current according to the reference voltage and the over-modulation signal; and a PWM signal generator which generates a pulse-width modulated signal based on the reference voltage modulated by the modulator and a carrier wave to apply the pulse-width modulated signal to an inverter.

Description

Inverter control device for a buried permanent magnet synchronous motor and its operating method TECHNICAL FIELD

Embodiments of the present invention relate to an inverter control device of an IPMSM that adjusts a voltage command, a current command, and a PWM signal to minimize leakage magnetic flux due to a permanent magnet, and an operating method thereof.

The content described in this section merely provides background information on the present invention and does not constitute the prior art.

An interior permanent magnet synchronous motor (IPMSM) is a synchronous motor in which a permanent magnet is inserted into the core of a rotor, and has excellent high-speed durability and high-speed operation. In particular, IPMSM is advantageous for maintenance due to its fully enclosed structure and is mainly used as a motor for railway vehicles and electric vehicles because it can increase output compared to the same capacity due to high torque per unit volume. In Europe and Japan, IPMSM is implemented and used for rail vehicles. In addition, research is underway to apply IPMSM to xEVs (EVs, HEVs, PHEVs, etc.) for electric vehicles.

An inverter is attached to the embedded permanent magnet synchronous motor, and the inverter control device generates a current command and a voltage command from an external torque command, and controls the inverter switch based on the voltage command to control the speed and torque of the synchronous motor. do.

The inverter control device controls the pulse-width modulation when controlling the switching of the inverter, and the PWM control area is divided into a linear area, an overmodulation area and a 6-step area according to the voltage command and the carrier wave. The linear region is a region in which the output voltage of the inverter can follow the voltage command linearly when the synchronous motor rotates at a low speed. The overmodulation region is a region where it is difficult for the output voltage of the inverter to follow the voltage command linearly, and is a region where overmodulation with respect to the voltage command is performed. The 6-step region is a region in which the output voltage of the inverter is output as a square wave, and is an region in which the inverter control device can stably and efficiently operate the synchronous motor at high speed.

The inverter control device modulates the voltage reference so that the output voltage of the inverter follows the voltage reference as linearly as possible in the overmodulation region. When the inverter control device controls the inverter with space vector pulse-width modulation (SVPWM), minimum distance overmodulation and in-phase overmodulation are used as overmodulation techniques to increase linearity. Minimum distance overmodulation is a method of modulating the voltage command up to the point where the voltage command is perpendicular to the hexagon by modulating both the phase and magnitude of the voltage command in the spatial voltage vector diagram. On the other hand, in-phase overmodulation is a method of modulating the voltage reference over a hexagon by fixing the phase of the voltage reference and adjusting the magnitude.

However, dynamic overmodulation methods such as in-phase overmodulation and minimum distance overmodulation have limitations in improving the linearity of PWM control. Also, the two methods cannot switch to the 6-step region without high-gain modulation.

On the other hand, the torque of the synchronous motor is limited due to the current limitation of the synchronous motor and the inverter, and the inverter control device controls the current and torque of the synchronous motor by controlling the voltage within the limiting range. At this time, the speed of the synchronous motor is also limited, and the inverter control device performs flux weakness control to rotate the synchronous motor at a higher speed.

The weak flux control is a control method that allows the synchronous motor to operate at low torque and high speed by adjusting the d-axis current command among the dq-axis current commands in the negative direction to attenuate the gap-linkage flux of the synchronous motor.

The weak magnetic flux control that adjusts the d-axis current command in the negative direction aims to enable the inverter output voltage to follow the voltage command linearly in the overmodulation region, and does not consider the overmodulation execution time. That is, there is a problem in that the voltage utilization ratio cannot be maximized because the conventional magnetic flux control performs overmodulation for a long time.

Meanwhile, the conventional PWM control operates in an asynchronous mode in which a PWM signal is generated while the frequency of a carrier wave is fixed, and a synchronous mode in which a carrier wave and a voltage command are synchronized. Specifically, the inverter control device operates in an asynchronous mode in which the carrier wave and the fundamental wave are not synchronized when the carrier frequency and the fundamental frequency ratio, ie, the pulse number, is low, such as in low-speed operation. On the other hand, when the modulation number is high, such as in high-speed operation, the carrier wave and the fundamental wave operate in a synchronized mode.

However, although the conventional PWM control can solve the imbalance of the inverter output current in the overmodulation region and the 6-step region through the synchronous mode, there is a problem in that it is difficult to reduce the switching loss and heat generated by numerous switching operations. .

Finally, the current control unit included in the inverter control device is a component that generates a voltage command according to the dq-axis current command. As the speed of the synchronous motor increases, the frequency of the PWM control signal increases, but in general, the current controller samples the dq-axis current reference at a constant period and generates a voltage reference. This lowers the control precision of the inverter control device and lowers the quick responsiveness of the voltage and current of the synchronous motor to follow the voltage command and the current command.

Another aspect of the present invention is to provide an inverter control device and an operating method capable of improving the long overmodulation time and low voltage utilization rate of weak magnetic flux control that adjusts the d-axis current command in the negative direction. have.

Embodiments of the present invention perform an overmodulation technique having higher linearity than a minimum distance overmodulation, but provide an inverter control device and an operating method capable of starting voltage command overmodulation even within a linear region according to the speed of a synchronous motor. It has a main purpose.

Another embodiment of the present invention is an inverter control device capable of reducing switching loss and heat generation by reducing the switching frequency of the inverter as well as resolving the imbalance of the inverter output current, but operating in a synchronous mode for synchronizing a carrier wave to a fundamental wave, and An object of the present invention is to provide a method of operation.

According to one aspect of the present invention, there is provided an inverter control device for a synchronous motor, comprising: a current controller configured to generate a voltage command according to a dq-axis current command; a modulator for generating an overmodulation signal and modulating the voltage command over a hexagon of a space voltage vector diagram when the speed of the synchronous motor reaches a first reference speed; a d-axis current compensator for adding a compensating current command to a d-axis current command among the dq-axis current commands according to the voltage command and the over-modulation signal; and a PWM signal generator for generating a pulse width modulated signal based on the voltage command and the carrier modulated by the modulator and applying it to the inverter.

According to another aspect of this embodiment, there is provided a method of operating an inverter control apparatus for a synchronous motor, the method comprising: obtaining a dq-axis current command; adding a compensation current command to the d-axis current command among the dq-axis current commands when the speed of the synchronous motor reaches the first reference speed; generating a voltage reference according to the compensated dq-axis current reference; modulating the voltage reference over a hexagon of a space voltage vector diagram; and generating a pulse width modulated signal based on the modulated voltage command and the carrier wave and applying the generated pulse width modulated signal to the inverter.

As described above, according to an embodiment of the present invention, the overmodulation time can be shortened by adding the compensating current command to the d-axis current command, and the voltage utilization rate of the inverter can be maximized.

According to another embodiment of the present invention, even if the voltage command is located within the linear region rather than the limit of the linear region, overmodulation can be performed according to the speed reference of the synchronous motor, and superior linearity can be obtained compared to the static overmodulation technique. .

According to another embodiment of the present invention, current imbalance, switching loss and heat generation can be reduced through a synchronization method of synchronizing a carrier wave to a voltage command in PWM control, and control precision by updating the sampling period of the current controller according to carrier frequency update It is possible to increase the responsiveness of the eddy current control unit.

1 is a configuration diagram illustrating components of an inverter control system according to an embodiment of the present invention.
2 is a configuration diagram illustrating components of an inverter control device according to an embodiment of the present invention.
3A and 3B are diagrams for comparing an over-modulation technique according to an embodiment of the present invention and a conventional over-modulation technique.
4 is a diagram illustrating a compensation current command according to an embodiment of the present invention.
5 is a diagram illustrating a voltage command overmodulation technique according to an embodiment of the present invention.
6 is a diagram illustrating a carrier frequency update process in a synchronous mode according to an embodiment of the present invention.
FIG. 7 is a diagram exemplified to explain control precision and quick response of a current controller according to an operation frequency update.
8 is a flowchart illustrating a process in which the inverter control device adjusts overmodulation and compensation current commands according to an embodiment of the present invention.
9 is a flowchart illustrating an operation process of an inverter control device according to an embodiment of the present invention.

Hereinafter, some embodiments of the present invention will be described in detail with reference to exemplary drawings. In adding reference numerals to the components of each drawing, it should be noted that the same components are given the same reference numerals as much as possible even though they are indicated on different drawings. In addition, in describing the present invention, if it is determined that a detailed description of a related known configuration or function may obscure the gist of the present invention, the detailed description thereof will be omitted.

In addition, in describing the components of the present invention, terms such as first, second, A, B, (a), (b), etc. may be used. These terms are only for distinguishing the component from other components, and the essence, order, or order of the component is not limited by the term. Throughout the specification, when a part 'includes' or 'includes' a certain element, it means that other elements may be further included, rather than excluding other elements, unless otherwise stated. . In addition, terms such as '~ unit' and 'module' described in the specification mean a unit that processes at least one function or operation, which may be implemented as hardware or software or a combination of hardware and software.

1 is a configuration diagram illustrating components of an inverter control system according to an embodiment of the present invention.

1, the mask cone 100, the torque current map 102, the torque / magnetic flux axis current calculator 104, the Q-axis inductance map 106, the current phase angle map 108, the speed calculator 110, A coordinate converter 112 , a position sensor 114 , an inverter control unit 116 , an inverter 118 and a synchronous motor 120 are shown.

The inverter control device (not shown) according to an embodiment of the present invention may mean the inverter control unit 116, and additionally a torque current map 102, a torque/magnetic flux axis current calculator 104, and a Q-axis inductance map ( 106 ), a current phase angle map 108 and a velocity calculator 110 , and a coordinate converter 112 may be further included.

The mascon 100 is a component that generates an acceleration signal or a braking signal by a user and generates a torque command accordingly. In this case, the mascon 100 may generate a forward reversing signal, a reverse reversing signal, or a reversing signal as an acceleration signal. The mask cone 100 is located in the engine room in the railway vehicle.

The torque current map 102 is a look-up table in which the magnitude of the phase current for generating torque in the synchronous motor 120 is recorded, and the magnitude of the phase current of the synchronous motor 120 according to the torque command of the mascon 100 is component that determines According to an embodiment of the present invention, when the synchronous motor 120 is a motor of a railway vehicle, the torque current map 102 determines the magnitude of the phase current using information on the applied load of the railway vehicle.

The torque/flux axis current calculator 104 is a component that generates a dq-axis current command using the phase current magnitude obtained from the torque current map 102 and the current phase angle obtained from the current phase angle map 108 . Here, the d-axis current command means a magnetic flux current command, and the q-axis current command means a torque current command. The torque/flux axis current calculator 104 according to an embodiment of the present invention calculates the magnitude of the d-axis current command first, and calculates the magnitude of the q-axis current command within the phase current magnitude range.

The Q-axis inductance map 106 is a look-up table in which the Q-axis inductance change according to the rotation of the synchronous motor 120 is recorded when the synchronous motor 120 is an interior permanent magnet synchronous motor (IPMSM). . Since the magnetic flux according to the rotation angle is irregular by the permanent magnet inside the IPMSM, the Q-axis inductance used to generate the dq-axis current command fluctuates frequently. In detail, since the Q-axis inductance varies greatly depending on the speed of the motor and the magnitude of the current applied to the motor, it is necessary to adjust the Q-axis inductance frequently. The Q-axis inductance map 106 receives current phase angle information from the current phase angle map 108 and provides an appropriate Q-axis inductance to the inverter controller 116 .

The current phase angle map 108 receives the magnitude of the phase current from the torque current map 102 and receives the speed information of the synchronous motor 120 from the speed calculator 110 to determine the current phase angle for generating the maximum torque under a given condition. Recorded lookup table.

The speed calculator 110 is a component that calculates the speed of the synchronous motor 120 by receiving position information from the position sensor 114 attached to the synchronous motor 120 .

The coordinate converter 112 is a component that converts the coordinate system of the three-phase output current of the inverter 118 into a dq-axis Cartesian coordinate system. That is, the three-phase output current represented by the U-phase, V-phase, and W-phase is converted into a d-axis current and a q-axis current by the coordinate converter 112 .

The position sensor 114 is a component that is attached to the synchronous motor 120 and measures the rotational position of the synchronous motor 120 . The position information measured by the position sensor 114 is used to calculate the speed of the synchronous motor 120 .

The inverter control unit 116 generates a voltage command for the dq-axis current command based on the Q-axis inductance information, the position information and speed information of the synchronous motor 120 , and the three-phase output current of the inverter 118 , and the voltage command It is a component that controls the switch of the inverter 118 by generating a PWM signal for The inverter control unit 116 will be described in detail with reference to FIG. 2 .

The inverter 118 is a component that flows a current to the synchronous motor 120 according to the PWM signal by the inverter control unit 116 . The inverter 118 may be implemented by including at least one of a DC voltage power supply, a capacitor, a plurality of switching elements, and a plurality of diodes.

The synchronous motor 120 is a motor that rotates according to the output current of the inverter 118 , and the synchronous motor 120 according to an embodiment of the present invention is an embedded permanent magnet synchronous motor.

2 is a configuration diagram illustrating components of an inverter control device according to an embodiment of the present invention.

Referring to FIG. 2 , an inverter control device (not shown) according to an embodiment of the present invention includes a d-axis current compensator 200 , a current controller 210 , a modulator 220 , a phase detector 230 , and a frequency It may include an update unit 240 and a PWM signal generator 250 .

The inverter control device receives the dq-axis current command, inductance information, three-phase output current of the inverter 118, position information and speed information of the synchronous motor 120 from the outside.

The d-axis current compensator 200 receives a voltage command from the current controller 210, receives an over-modulation signal from the modulator 220, and generates a compensating current command and adds it to the d-axis current command. All. That is, the d-axis current command among the dq-axis current commands is adjusted by the d-axis current compensator 200 .

Specifically, the d-axis current compensator 200 maintains a difference between the magnitude of the back EMF voltage generated by the characteristics of the IPMSM of the synchronous motor 120 and the magnitude of the output voltage of the inverter 118 in order to maintain the d-axis current. Compensate the reference as much as the compensating current reference. The compensating current command may be added to the dq-axis current command by the torque/flux axis current calculator 104 in addition to the d-axis current compensator 200 .

The d-axis current compensator 200 according to an embodiment of the present invention generates a compensation current command having a positive magnitude and adds it to the d-axis current command. The compensating current command according to another embodiment of the present invention decreases after increasing to a preset value from when the d-axis current compensator 200 receives the overmodulation signal. When the compensating current command increases, it has a value greater than 0, and after reaching a preset value, it gradually decreases to a value less than 0. At this time, the compensating current command has a shape of a convex peak and approaches an asymptote having a negative value as time goes by.

The current controller 210 is a component that generates a voltage command according to the dq-axis current command. Specifically, the current control unit 210 is a proportional integral (PI) for the output current of the inverter 118 and the dq-axis current command based on the Q-axis inductance information, the position information and the speed information of the synchronous motor 120 ) control to generate a voltage command. Also, the current controller 210 may perform current control using various techniques including PI, PID, IP, PI-IP, and Puzzy. The output current of the inverter 118 may follow the dq-axis current command by the voltage command generated by the current controller 210 .

Meanwhile, the current controller 210 samples the dq-axis current command according to the operating frequency and generates a voltage command. According to an embodiment of the present invention, the operating frequency of the current controller 210 is updated according to the carrier frequency update by the frequency updater 240 . At this time, the current control unit 210 samples the dq-axis current command according to the updated operating frequency and generates a voltage command. For example, as the speed of the synchronous motor 120 increases, the carrier frequency also increases, and when the carrier frequency increases, the operating frequency of the current controller 210 also increases. For this reason, the responsiveness of the output current of the inverter 118 to the dq-axis current command is improved, and the inverter control device can precisely control the inverter 118 .

When the speed of the synchronous motor 120 reaches the first reference speed, the modulator 220 is a component that generates an overmodulation signal and modulates the voltage command over the hexagon of the space voltage vector diagram. Here, the first reference speed may be set by the user. The space voltage vector diagram is a vector diagram expressing the voltage command in the dq-axis coordinate system. The maximum output voltage of the inverter is expressed as a hexagon, and the inscribed circle of the hexagon means a linear modulation region.

The modulator 220 does not overmodulate until the speed of the synchronous motor 120 reaches the first reference speed. On the other hand, when the speed of the synchronous motor 120 reaches the first reference speed, the modulator 220 starts overmodulation according to the speed of the synchronous motor 120 even if the voltage command is within the linear region. The modulator 220 finally places the voltage command at the hexagonal vertex of the spatial voltage vector diagram to operate the inverter 118 in the 6-step mode. In the above process, the modulator 220 may use position information and speed information of the synchronous motor 120 . After the inverter 118 enters the 6-step region, the modulator 220 maintains the inverter 118 in the 6-step mode until the speed of the synchronous motor 120 decreases to a specific speed.

A detailed overmodulation method of the modulator 220 will be described with reference to FIG. 5 .

The phase detector 230 is a component that detects the phase of the voltage command generated by the current controller 210 . The phase detector 230 provides the voltage vector phase of the voltage command to the frequency updater 240 .

When the speed of the synchronous motor 120 reaches the second reference speed, the frequency updater 240 updates the frequency of the carrier so that the carrier is synchronized with the voltage command in consideration of the voltage vector phase and transmits it to the PWM signal generator 250 . is a component that Also, the frequency updater 240 updates the operating frequency of the current controller 210 based on the updated frequency of the carrier wave and transmits the update to the current controller 210 . Here, the first reference speed may be set by the user. The PWM signal generator 250 generates a PWM signal using the updated carrier frequency, and the current controller 210 operates according to the updated operating frequency.

The frequency updater 240 does not update the frequency of the carrier until the speed of the synchronous motor 120 reaches the second reference speed. However, the frequency updater 240 according to an embodiment of the present invention may update the operating frequency of the current controller separately from the update of the carrier frequency. The frequency updater 240 according to another embodiment of the present invention may update the operating frequency based on the updated carrier frequency.

The updating process of the operating frequency can be expressed as Equation (1).

는 갱신된 동작 주파수,

은 PI제어에 이용되는 파라미터,

는 인버터 스위칭 주파수,

는 반송파의 한 주기를 의미한다.

은 PI 제어에서 P 게인과 I 게인을 계산하는데 이용될 수 있다. 결과적으로, 반송파 주파수 갱신에 따라 인버터 스위칭 주파수가 증가하면, 전류 제어부의 동작 주파수도 증가한다.in Equation 1

is the updated operating frequency,

is the parameter used for PI control,

is the inverter switching frequency,

denotes one cycle of the carrier wave.

can be used to calculate P gain and I gain in PI control. As a result, when the inverter switching frequency increases according to the carrier frequency update, the operating frequency of the current control unit also increases.

The frequency updater 240 according to an embodiment of the present invention updates the frequency of the carrier so that an integer number of carriers is included in one period of the voltage command. The frequency updater 240 according to another embodiment of the present invention updates the carrier frequency so that the carrier wave is symmetric to the middle value of the voltage command period. The frequency updater 240 according to another embodiment of the present invention may update the carrier frequency so that an integer number of carriers is included in one period of the voltage command and the carriers are symmetrical to the middle value of the voltage command period. The above-described process may be performed at an operating frequency of the current controller 210 , that is, every sampling period.

The PWM signal generator 250 is a component that generates a PWM signal based on a voltage command and a carrier wave and applies it to the inverter 118 . When the speed of the synchronous motor 120 reaches the first reference speed, the PWM signal generator 250 generates a PWM signal based on the voltage command and the carrier modulated by the modulator 220 . Also, when the speed of the synchronous motor 120 reaches the second reference speed, the PWM signal generator 250 generates a PWM signal using the carrier frequency updated by the frequency updater 240 . At this time, when the voltage command is expressed in the dq-axis coordinate system, the PWM signal generator 250 may generate a PWM signal after converting the voltage command into a three-phase voltage command, and a triangular carrier wave may be used as a carrier wave. That is, the PWM signal generator 250 may generate a space vector pulse width modulation (SVPWM) signal and additionally use a dead time. In addition, it serves as a gate driver that turns on/off the switching element of the inverter 118 .

3A is a diagram illustrating a comparison between an inverter control method according to an embodiment of the present invention and a conventional inverter control method.

Referring to FIG. 3A , a graph of the first phase voltage 300 according to the conventional inverter control method and the second phase voltage graph according to the inverter control method according to the present embodiment is shown. The first phase voltage 300 and the second phase voltage 310 both mean the output phase voltage of the inverter, and the voltage located at the boundary line between the linear region and the overmodulation region corresponds to the circumference of the hexagonal inscribed circle in the space voltage vector diagram. . In addition, a voltage at a portion where over-shoot starts in the over-modulation region is a voltage corresponding to the length of one side of the hexagon. That is, it means the distance from the origin to the vertex of the hexagon.

First, they coincide with each other in the linear domain and the 6-step domain. This is because the two inverter control methods are the same in the linear domain and the 6-step domain.

However, in the overmodulation region, the first phase voltage 300 has a gentle slope compared to the second phase voltage 310 , and it takes longer to perform the overmodulation. On the other hand, the second phase voltage 310 has a steeper slope than the first phase voltage 300 and is linear, and the overmodulation execution time is short. By the d-axis current compensation command, the overmodulation method, the synchronization mode through carrier frequency update, and the operating frequency (sampling period) update of the current controller according to an embodiment of the present invention, the inverter moves from a linear region to a 6-step region can be converted quickly and reliably.

3B is a diagram illustrating a comparison between an inverter control method according to an embodiment of the present invention and a conventional inverter control method.

Referring to FIG. 3B , a first output voltage 320 according to a conventional inverter control method and a second output voltage 330 according to an embodiment of the present invention are shown. The first output voltage 320 and the second output voltage 330 mean an inverter output phase voltage. The first output voltage 320 and the second output voltage 330 are switched in the order of a linear modulation region, an overmodulation region, and a 6-step region.

The first output voltage 320 is not naturally switched from the left linear modulation region to the right 6-step region, and is not completely switched to the 6-step region. Even if the first output voltage 320 enters the 6-step region, the first output voltage 320 has a square wave shape including noise.

On the other hand, in the second output voltage 330 , the 6-step region is clearly distinguished from other regions. When the second output voltage 330 enters 6-step, it has an accurate square wave shape. Therefore, the process in which the second output voltage 330 according to an embodiment of the present invention is switched to a 6-step region is more stable than that of the first output voltage 320 according to the conventional inverter control method.

Also, the first output voltage 320 contains a lot of noise in the 6-step region, which means that the switching frequency is high in the 6-step region. That is, there are many inverter switching times within one cycle.

On the other hand, in the second output voltage 330 , the number of inverter switching within one cycle in the 6-step region is much smaller than that of the first output voltage 320 . Accordingly, the second output voltage 330 means less switching loss and less heat than the first output voltage 320 .

4 is a diagram illustrating a compensation current command according to an embodiment of the present invention.

Referring to FIG. 4 , the maximum speed 400 of the synchronous motor, the rotation speed 410 , the compensation current command 420 , the 6-step voltage 430 , the overmodulation start voltage 440 , and the output phase voltage 450 . This is shown. T1 denotes a linear modulation region, T2 denotes an overmodulation region, and T3 denotes a 6-step region.

The rotational speed 410 of the synchronous motor gradually increases over time, and the maximum speed 400 is targeted.

When the speed of the synchronous motor reaches the first reference speed, the compensation current command 420 has a positive magnitude, increases to a preset value, and then decreases to a negative value again. A time point at which the compensation current command 420 decreases between T2 and T3 is a time point at which the inverter control device performs the weak magnetic flux control.

The 6-step voltage 430 means the distance from the origin to the vertex of the hexagon in the space voltage vector diagram, and the inverter outputs a phase voltage like the 6-step voltage 430 in the 6-step mode. The overmodulation start voltage 440 may mean the radius of the hexagonal inscribed circle, and may mean the phase voltage output by the inverter when the synchronous motor reaches the first reference speed according to an embodiment of the present invention. .

The output phase voltage 450 means a voltage for one of the three-phase voltages transferred from the three-phase output voltage of the inverter to the synchronous motor. The output phase voltage 450 is a result of overmodulation from the time when the compensation current command 420 is generated, that is, when the synchronous motor reaches the first reference speed. The output phase voltage 450 may reach the 6-step voltage 430 within a short time by the compensation current command 420 and the overmodulation technique.

5 is a diagram illustrating a voltage command overmodulation technique according to an embodiment of the present invention.

5 shows a hexagon representing the maximum output voltage of the inverter, an inscribed circle representing a linear modulation region, a voltage command, and a modulated voltage command on the space voltage vector diagram.

The first stage is a stage in which the inverter control device starts overmodulating the voltage command according to the speed of the synchronous motor. Before step 1, the modulator outputs the voltage command as it is. The modulator places the voltage command above the hexagon when the speed of the synchronous motor reaches the first reference speed even if the voltage command is within the linear region. That is, when the voltage command is within the inscribed circle of the hexagon, the modulator according to an embodiment of the present invention adjusts the magnitude of the voltage command to position the voltage command above the hexagon. At this time, the phase of the voltage command is not adjusted, only the magnitude is adjusted.

In steps 2 and 3, if the magnitude of the voltage command is greater than the hexagon's inscribed circle radius and smaller than the length of one side of the hexagon, the modulator adjusts the phase of the voltage command to position the voltage command as the nearest intersection among the intersections with the hexagon. it's a step When the voltage reference is on the hexagon, the modulator outputs the voltage reference as it is. At this time, the magnitude of the voltage command is not adjusted, only the phase is adjusted.

In step 4, if the size of the voltage command is greater than the length of one side of the hexagon, the modulator places the voltage command as the nearest vertex among the vertices of the hexagon. In this case, the modulator adjusts both the magnitude and the phase of the voltage command. In 6-step mode after step 4, it is located only at the vertex of the hexagon of the voltage command.

The above-described overmodulation technique allows the inverter output phase voltage to linearly follow the voltage command compared to the conventional in-phase overmodulation or minimum distance overmodulation.

6 is a diagram illustrating a carrier frequency update process in a synchronous mode according to an embodiment of the present invention.

6 shows a voltage command 600 , a carrier wave 610 , and an updated carrier wave 620 . The voltage command 600 refers to one phase of the three-phase voltage command, and the carrier wave is a triangular carrier wave.

In general, as the speed of the synchronous motor increases, the period of the voltage command becomes shorter and the period of the carrier wave also becomes shorter. In addition, when the speed of the synchronous motor is increased, the current phase angle is changed, and the phase angle of the voltage command is also changed every moment. When sampling is performed for closed-loop control of a synchronous motor, if the voltage command 600 and the carrier wave 610 are not synchronized due to a sampling error, an imbalance problem occurs in the inverter output current in the overmodulation region and the 6-step region. .

)는 수학식 2와 같다. 즉, 주파수 갱신부는 반송파(610)의 주기에

를 더함으로써, 주파수를 갱신한다.Accordingly, in one embodiment of the present invention, the frequency update unit updates the frequency of the carrier wave every sampling period of the current controller and provides it to the PWM signal generator to synchronize the voltage command and the carrier wave. The difference between the period of the carrier wave 610 and the period of the updated carrier wave 620 (

) is the same as in Equation 2. That is, the frequency update unit is in the cycle of the carrier wave (610).

By adding , the frequency is updated.

는 주파수가 갱신된 반송파의 주기,

는 인버터 출력 주파수,

는 dq-축 전압 지령의 위상각, N은 기 설정된 변조수, int(x+0.5)는 x를 반올림하는 함수를 의미한다. 변조수는 전압 지령의 한 주기 내 샘플링 횟수, 즉 반송파 개수를 의미한다.In Equation 2, the first term on the right means the period of the carrier before the frequency is updated, and the second term on the right means the compensation period for updating the frequency of the carrier.

is the period of the carrier whose frequency is updated,

is the inverter output frequency,

is the phase angle of the dq-axis voltage command, N is a preset modulation number, and int(x+0.5) is a function that rounds x. The number of modulation means the number of samplings within one cycle of the voltage command, that is, the number of carriers.

The frequency-updated carrier wave may be included in a preset number within one period of the voltage command. In addition, the carrier wave is symmetrical with respect to the middle value of one voltage command period. Mainly, since the period of the carrier wave whose frequency is updated becomes longer, the switching frequency of the inverter is lower than that of the carrier wave before the update.

Therefore, it is possible to solve the problem of unbalance of inverter output current and reduce switching loss and heat generation through carrier frequency update.

FIG. 7 is a diagram exemplified to explain the control precision and quick response of the current controller according to the update of the operating frequency.

7 shows a rotational speed 700 of the synchronous motor, an updated operating frequency 710 , an inverter switching frequency 720 , a constant operating frequency 730 , a first d-axis current 740 , and a second d-axis current 750 , a first q-axis current 760 , and a second q-axis current 760 are shown.

The rotation speed 700 of the synchronous motor and the switching frequency 720 of the inverter gradually increase with time, and the conventional current controller operates at an operating frequency 730 having a constant value. Here, the operation of the current controller refers to the operation of sampling the dq-axis current command and generating the voltage command. The current controller according to an embodiment of the present invention operates with the operating frequency 710 updated by the frequency updater.

Referring to the graph for the d-axis current command, the first d-axis current 740 is a graph in which the output current of the inverter is measured when the current controller operates according to the updated operating frequency 710, and the second d The -axis current 750 is a graph in which the output current of the inverter is measured when operating according to the constant operating frequency 730 . It can be seen that the first d-axis current 740 follows the d-axis current command faster than the second d-axis current 750 .

Referring to the graph for the q-axis current command, the first q-axis current 760 is the inverter output current according to the constant operating frequency 730, and the second q-axis current 770 is the updated operating frequency ( 710) according to the inverter output current. The second q-axis current 770 is closer to the q-axis current command than the first q-axis current 760, and the error from the q-axis current command is smaller.

Accordingly, by updating the frequency of the current control unit, the inverter control device can more precisely control the synchronous motor and increase the quick response to the command.

8 is a flowchart illustrating a process in which the inverter control device adjusts overmodulation and compensation current commands according to an embodiment of the present invention.

The inverter control device obtains a dq-axis current command from the torque/flux axis current calculator (S800).

When the speed of the synchronous motor reaches the first reference speed, the inverter control device adds a compensating current command to the d-axis current command among the dq-axis current commands ( S802 ). The compensating current command according to an embodiment of the present invention may have a positive magnitude, a shape that decreases after increasing to a preset value, or may have both characteristics.

The inverter control device generates a voltage command according to the compensated dq-axis current command (S804). The inverter control device according to an embodiment of the present invention may update an operating frequency indicating a generation frequency of a voltage command according to a carrier frequency update, and may generate a voltage command based on the updated operating frequency.

The inverter control device modulates the voltage command over the hexagon of the space voltage vector diagram (S806). In the inverter control device according to an embodiment of the present invention, when the voltage command is within the inscribed circle of the hexagon, the voltage command is positioned above the hexagon by adjusting the magnitude of the voltage command. In addition, when the magnitude of the voltage command is greater than the radius of the hexagon's inscribed circle and smaller than the length of one side of the hexagon, the inverter control device adjusts the phase of the voltage command to position the voltage command as the nearest intersection among the intersections with the hexagon. . When the magnitude of the voltage command is greater than the length of one side of the hexagon, the inverter control device performs overmodulation by locating the voltage command to the nearest vertex among the vertices of the hexagon.

The inverter control device generates a pulse width modulated signal based on the modulated voltage command and the carrier wave and applies it to the inverter (S808). The inverter control apparatus according to an embodiment of the present invention may update a frequency of a carrier and generate a PWM signal using the updated carrier. Specifically, the inverter control device detects the voltage vector phase of the voltage command, and when the speed of the synchronous motor reaches the second reference speed, the frequency of the carrier wave can be updated in consideration of the voltage vector phase so that the carrier wave is synchronized with the voltage command. have. In addition, the inverter control device may update the frequency of the carrier so that an integer number of carriers are included in one period of the voltage command and the carriers are symmetrical with respect to the middle value of the voltage command period.

The inverter operates according to the PWM signal of the inverter control device, and the synchronous motor operates according to the output of the inverter.

9 is a flowchart illustrating an operation process of an inverter control device according to an embodiment of the present invention.

Hereinafter, it will be described that the inverter control device includes a torque current map, a torque/flux axis current calculator, a Q-axis inductance map, a current phase angle map, a speed calculator, a coordinate converter, and an inverter controller.

The inverter control device receives the torque command from the mascon and receives the applied load information from the external device to generate the dq-axis current command (S900). Here, the load information may be a load value of the railroad vehicle.

The inverter control device generates a voltage command by performing PI control on the dq-axis current command (S902). Specifically, the inverter control device performs PI control so that the three-phase output current of the inverter follows the dq-axis current command. However, PI control is only one embodiment, and the inverter control device may perform at least one of PID, IP, PI-IP, and puzzy control. In addition, the q-axis inductance and the d-axis inductance received from the Q-axis inductance map may be used. In addition, the inverter control device may use the SVPWM control method.

The inverter control device generates a PWM signal from the voltage command to rotate the synchronous motor, and determines whether the speed of the synchronous motor reaches a first reference speed (S906). The first reference speed may be set by a user. According to an embodiment of the present invention, it may be further determined whether the period of the carrier has reached a preset period.

If the speed of the synchronous motor does not reach the first reference speed, the inverter control device controls the inverter within the linear modulation region (S908). According to an embodiment of the present invention, even if the speed of the synchronous motor reaches the first reference speed, if the period of the carrier wave does not reach the preset period, the inverter control apparatus may operate in the linear modulation region. In this case, the inverter control device may use the SVPWM control and perform the weak magnetic flux control.

On the other hand, if the speed of the synchronous motor reaches the first reference speed, the inverter control device detects the phase of the voltage command, synchronizes the carrier to the voltage command, and updates the operating frequency according to the carrier frequency (S910). Here, the operating frequency means the frequency at which the inverter control device samples the dq-axis current command and generates the voltage command. The operation frequency update process according to another embodiment of the present invention may be performed only when the synchronous motor speed reaches the first reference speed and the carrier wave period reaches a preset period.

Thereafter, the inverter control device determines whether the speed of the synchronous motor reaches a second reference speed (S912). Here, the second reference speed is higher than the first reference speed. If the synchronous motor speed does not reach the second reference speed, the inverter control device operates within the linear modulation region. The inverter control apparatus according to an embodiment of the present invention may further determine whether a synchronous closed-loop control error between the carrier wave and the voltage command in the synchronous mode falls within a predetermined range.

When the synchronous motor speed reaches the second reference speed, the inverter control device generates a d-axis compensation current command and adds it to the d-axis current command (S914). At this time, the d-axis current command increases in a positive direction and decreases after reaching a preset value. The inverter control apparatus according to another embodiment of the present invention may generate the compensating current command only when the synchronous motor speed reaches the second reference speed and the synchronous closed-loop control error falls within a predetermined range.

The inverter control device over-modulates the voltage command while generating the compensating current command (S916).

The inverter control device controls the inverter in the 6-step mode after performing overmodulation for a short time (S918).

Although it is described that steps S800 to S918 are sequentially executed in FIGS. 8 and 9 , this is merely illustrative of the technical idea of an embodiment of the present invention. In other words, those of ordinary skill in the art to which an embodiment of the present invention pertain change the order described in FIGS. 8 and 9 within the range that does not deviate from the essential characteristics of an embodiment of the present invention, or process S800 to process S800. Since it will be possible to apply various modifications and variations to parallel execution of one or more processes in S918, FIGS. 8 and 9 are not limited to a time-series order.

Meanwhile, the processes shown in FIGS. 8 and 9 can be implemented as computer-readable codes on a computer-readable recording medium. The computer-readable recording medium includes all kinds of recording devices in which data readable by a computer system is stored. That is, such a computer-readable recording medium may be a non-transitory medium such as ROM, RAM, CD-ROM, magnetic tape, floppy disk, optical data storage device, etc., and also carrier wave (for example, , transmission over the Internet) and may further include a transitory medium such as a data transmission medium. In addition, the computer-readable recording medium is distributed in a network-connected computer system so that the computer-readable code can be stored and executed in a distributed manner.

The above description is merely illustrative of the technical idea of this embodiment, and various modifications and variations will be possible by those skilled in the art to which this embodiment belongs without departing from the essential characteristics of the present embodiment. Accordingly, the present embodiments are intended to explain rather than limit the technical spirit of the present embodiment, and the scope of the technical spirit of the present embodiment is not limited by these embodiments. The protection scope of the present embodiment should be interpreted by the following claims, and all technical ideas within the equivalent range should be construed as being included in the scope of the present embodiment.

200: d-axis current compensator 210: current control
220: modulator 230: phase detector
240: frequency update unit 250: PWM signal generator

Claims (19)

An inverter control device for a synchronous motor comprising:
a current control unit for generating a reference voltage according to a dq-axis current reference;
a modulator for generating an overmodulation signal and modulating the voltage command over a hexagon of a space voltage vector diagram when the speed of the synchronous motor reaches a first reference speed;
a d-axis current compensator for adding a compensating current command to a d-axis current command among the dq-axis current commands according to the voltage command and the over-modulation signal; and
a PWM signal generator for generating a pulse-width modulation (PWM) signal based on the voltage command and the carrier modulated by the modulator and applying it to the inverter;
Inverter control device comprising a. According to claim 1,
The inverter control device, characterized in that the compensation current command has a positive magnitude. According to claim 1,
The inverter control device, characterized in that the compensation current command increases to a preset value and then decreases. According to claim 1,
The modulator,
If the voltage command is within the inscribed circle of the hexagon, the inverter control device for positioning the voltage command above the hexagon by adjusting the magnitude of the voltage command. According to claim 1,
The modulator,
When the magnitude of the voltage command is greater than the radius of the inscribed circle of the hexagon and smaller than the length of one side of the hexagon, by adjusting the phase of the voltage command, inverter control to position the voltage command as the nearest intersection among the intersections with the hexagon Device. According to claim 1,
The modulator,
If the magnitude of the voltage command is greater than the length of one side of the hexagon, the inverter control device for positioning the voltage command as a nearest vertex among the vertices of the hexagon. According to claim 1,
a phase detector for detecting a voltage vector phase of the voltage command; and
a frequency updater configured to update the frequency of the carrier so that the carrier is synchronized with the voltage command in consideration of the voltage vector phase when the speed of the synchronous motor reaches a second reference speed;
Inverter control device further comprising a. 8. The method of claim 7,
The frequency update unit,
The inverter control device for updating the frequency of the carrier so that an integer number of the carriers are included in one period of the voltage command and the carriers are symmetrical with respect to a median value of the voltage command period. 8. The method of claim 7,
The frequency update unit updates the operating frequency of the current control unit based on the updated frequency of the carrier,
The current control unit generates the voltage command based on the updated operating frequency. A method of operating an inverter control device for a synchronous motor, the method comprising:
obtaining a dq-axis current reference;
adding a compensation current command to the d-axis current command among the dq-axis current commands when the speed of the synchronous motor reaches the first reference speed;
generating a voltage reference according to the compensated dq-axis current reference;
modulating the voltage reference over a hexagon of a space voltage vector diagram; and
generating a pulse width modulated signal based on the modulated voltage command and the carrier and applying it to the inverter;
A method of operating an inverter control device comprising a. 11. The method of claim 10,
The method of operating an inverter control device, characterized in that the compensation current command has a positive magnitude. 11. The method of claim 10,
The method of operating an inverter control device, characterized in that the compensation current command increases to a preset value and then decreases. 11. The method of claim 10,
The modulation process is
and positioning the voltage command above the hexagon by adjusting the magnitude of the voltage command when the voltage command is within the inscribed circle of the hexagon. 11. The method of claim 10,
The modulation process is
When the magnitude of the voltage command is greater than the radius of the inscribed circle of the hexagon and smaller than the length of one side of the hexagon, by adjusting the phase of the voltage command, the process of positioning the voltage command as the nearest intersection among the intersections with the hexagon A method of operating an inverter control device comprising a. 11. The method of claim 10,
The modulation process is
and when the magnitude of the voltage command is greater than the length of one side of the hexagon, positioning the voltage command as a nearest vertex among the vertices of the hexagon. 11. The method of claim 10,
detecting a voltage vector phase of the voltage command; and
updating the frequency of the carrier wave so that the carrier wave is synchronized with the voltage command in consideration of the voltage vector phase when the speed of the synchronous motor reaches a second reference speed;
The method of operation of the inverter control device further comprising a. 17. The method of claim 16,
The process of updating the frequency is
and updating the frequency of the carrier so that an integer number of the carriers are included in one period of the voltage command and the carrier waves are symmetrical with respect to an intermediate value of the voltage command period. 17. The method of claim 16,
Further comprising the step of updating an operating frequency indicating the frequency of generation of the voltage command according to the updated frequency of the carrier wave,
The generating of the voltage command is a process of generating the voltage command based on an updated operating frequency. A computer-readable recording medium in which a program for executing the method of any one of claims 10 to 19 in a computer is recorded.

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