KR20200123600A - Phase shifter and phase shifting method - Google Patents

Abstract

A phase shifter according to the present invention may comprise: a frequency locked loop (FLL) sampling frequencies of an input signal, generating control codes according to the sampled frequencies, and dynamically converting the generated control codes to make the sampled frequencies same with free oscillation frequencies of a quadrature oscillator; a quadrature oscillator generating an output signal having frequencies synchronized with the frequencies of the input signal based on the free oscillation frequencies adjusted according to the control codes; and a phase controller controlling a phase of the output signal. The phase shifter for an FMCW radar device according to the present invention may have an extended operating frequency bandwidth.

Description

Phase shifter and phase shift method {PHASE SHIFTER AND PHASE SHIFTING METHOD}

The present invention relates to a phase shifter and a phase shift method, and more particularly, to a phase shifter and a phase shift method for an FMCW radar device.

A frequency modulation continuous wave radar (FMCW radar) device uses a frequency-modulated modulated wave signal. The FMCW radar device can measure the distance between the target and the radar device by measuring the beat frequency by mixing a part of the reflected wave signal reflected from the target and the transmission frequency. The FMCW radar device can be used for water gauges in aircraft altimeter tanks and tide gauges. Recently, the FMCW radar device is expanding its application range, and it is being developed not only to measure distance but also to detect direction.

The FMCW radar device performs measurement operations using beamforming, and uses a phase shifter for beamforming. In order to diversify the use of the FMCW laser device and improve measurement capability, the operating frequency bandwidth of the phase shifter has a great influence. In accordance with recent developments, various methods for overcoming the limit of the operating frequency bandwidth of the phase shifter used in the FMCW radar device are being studied.

The present invention is to solve the above-described technical problem, the present invention can provide a phase shifter and a phase shift method for an FMCW radar device.

The phase shifter according to the present invention samples the frequencies of the input signal, generates control codes according to the sampled frequencies, and dynamically converts the generated control codes to obtain sampled frequencies and a quadrature oscillator ( A frequency locked loop (FLL) in which the free oscillation frequencies of the quadrature oscillator are the same, and an output signal having frequencies synchronized with the frequencies of the input signal based on the free oscillation frequencies adjusted according to the control codes. It may include a quadrature oscillator to generate, and a phase controller that controls the phase of the output signal.

The phase shifting method according to the present invention includes the steps of sampling frequencies of an input signal received by a phase shifter by a frequency locked loop, and adjusting free oscillation frequencies of the quadrature oscillator according to the sampled frequencies by a frequency locked loop Generating a plurality of control codes to perform and dynamically converting the generated control codes, converting a phase of an input signal by a phase converter, a plurality of dynamically converted controls by a quadrature oscillator Generating an output signal having frequencies synchronized with the frequencies of the converted input signal based on the free oscillation frequencies adjusted according to the free oscillation frequencies adjusted according to the codes, the phase of the output signal by a phase controller It may include controlling and outputting an output signal by a quadrature oscillator.

The phase shifter according to the present invention may enable the FMCW radar device to have a high resolution direction detection function.

The phase shifter for the FMCW radar device according to the present invention may have an extended operating frequency bandwidth.

1 is a block diagram illustrating a phase shifter according to an embodiment of the present invention.
FIG. 2 is a circuit diagram illustrating an exemplary quadrature oscillator of FIG. 1.
3 is a phase diagram for illustratively explaining the operation of the circuit diagram of FIG. 2.
4 are graphs for explaining a common synchronization range of frequencies set according to the phase shifter of FIG. 1.
5 is a circuit diagram illustrating an exemplary configuration of a capacitor included in the circuit diagram of FIG. 2.
6A to 6C are graphs exemplarily showing a synchronization process of the free oscillation frequency of the quadrature oscillator set according to the circuit diagram of FIG. 5.
7 is a block diagram illustrating a phase shifter according to another embodiment of the present invention.
FIG. 8 is a circuit diagram showing the phase shifter of FIG. 7 in more detail.
9 is a flowchart illustrating an exemplary phase shifting method for beamforming of a radar device according to an embodiment of the present invention.

In the following, embodiments of the present invention will be described clearly and in detail to the extent that those skilled in the art can easily implement the present invention.

1 is a block diagram illustrating a phase shifter according to an embodiment of the present invention. The phase shifter 100 may be used for beamforming of a radar device (not shown). The radar device may be a frequency modulation continuous wave radar (FMCW radar). The phase shifter 100 may receive an input signal Sin from a radar device. The input signal Sin input to the phase shifter 100 may be a modulated wave signal frequency modulated by a radar device. The phase shifter 100 may output a final output signal Sout having a phase different from that of the input signal Sin by phase shifting, but having the same frequency as the input signal Sin. Referring to FIG. 1, the phase shifter 100 may include a phase converter 110, a quadrature oscillator 120, and a phase selector 130.

The phase converter 110 may receive an input signal Sin input to the phase shifter 100. The phase converter 110 may convert the phase of the input signal Sin. The phase converter 110 may convert the phase of the input signal Sin in units of 90 degrees. For example, the phase converter 110 generates output phases Si and Sq that differ by 90 degrees from the phase of the input signal Sin. The phase converter 110 may also be referred to as a 90 degree phase shifter.

The quadrature oscillator 120 may receive the first and second converted input signals Si and Sq from the phase converter 110. When the frequencies of the first and second converted input signals (Si, Sq) are included in the locking range of the quadrature oscillator 120, the quadrature oscillator 120 is converted to the first and second First and second output signals Ti' and Tq' having a frequency locked to the input signals Si and Sq may be output. The quadrature oscillator 120 may be an injection locked oscillator. When the input signal Sin is not input, the quadrature oscillator 120 may output an output signal having a free oscillation frequency. The quadrature oscillator 120 may lock the free oscillation frequency with the frequencies of the first and second converted input signals Si and Sq. The quadrature oscillator 120 may have a differential structure. The quadrature oscillator 120 may include first and second mixers 121 and 122, and an in-phase oscillator 123 and a quadrature-phase oscillator 124.

The first and second mixers 121 and 122 may mix at least two signals input to each. The first and second mixers 121 and 122 may receive at least two signals having the same frequency. The first and second mixers 121 and 122 may mix and output at least two received signals.

The in-phase oscillator 123 and the quadrature oscillator 124 may receive signals output from the first and second mixers 121 and 122, respectively. Each of the in-phase oscillator 123 and the quadrature oscillator 124 may be an injection synchronous oscillator. The in-phase oscillator 123 and the quadrature oscillator 124 may be configured to be cross-coupled to each other. The range of the phase change of the first and second output signals Ti' and Tq' of the in-phase oscillator 123 and the quadrature oscillator 124 may be -90 degrees to 90 degrees. Due to the differential structure of the in-phase oscillator 123 and the quadrature oscillator 124 cross-connected to each other, the range of the phase shift of the phase shifter 100 may be 0 to 360 degrees. The in-phase oscillator 123 and the quadrature oscillator 124 are formed into first and second mixers 121 and 122, respectively, by a structure in which the in-phase oscillator 123 and the quadrature oscillator 124 are cross-connected to each other. The first and second phase control signals Ci and Cq may be provided. The in-phase oscillator 123 and the quadrature oscillator 124 may be electrically connected to each other by the first and second phase control signals Ci and Cq, and may be coupled to each other. Phases of the first and second output signals Ti' and Tq' may be determined by the first and second phase control signals Ci and Cq. The magnitudes of the first and second phase control signals Ci and Cq may be variable, and a process in which the first and second phase control signals Ci and Cq are controlled will be described in more detail in FIG. 7. .

The phase selector 130 may receive an output signal from the quadrature oscillator 120. The phase selector 130 includes the first and second output signals Ti' and Tq' and the first and second output signals Ti' and Tq' so that the phases of the first and second output signals Ti' and Tq' include 0 to 360 degrees. One of the phase inversion signals ~Ti' and ~Tq' of each of the first and second output signals Ti' and Tq' may be selected. Here, the phase inversion signals (~Ti', ~Tq') of each of the first and second output signals Ti' and Tq' are in phase with the first and second output signals Ti' and Tq'. These 180-degree differences could mean signals. The phase selector 130 may output a final output signal Sout by selecting one of the first and second output signals Ti' and Tq' positioned in the first quadrant on the phase diagram. In the end, the final output signal Sout has the same frequency as the input signal Sin by frequency locking of the quadrature oscillator 120, but the input signal Sin has a different phase due to phase shift. I can. The phase selector 130 may also be referred to as a quadrant phase selector.

FIG. 2 is a circuit diagram illustrating an exemplary quadrature oscillator of FIG. 1. FIG. 2 will be described with reference to FIG. 1. The circuit 200 may include an oscillator 210 and first to second current sources 220 and 230. The oscillator 210 may be any one of the in-phase oscillator 123 and the quadrature oscillator 124 of FIG. 1. The oscillator 210 may receive the converted input signal Sj from the first current source 220 and may receive the phase control signal Cj from the second current source 230. The converted input signal Sj may be the first or second converted input signal Si and Sq of FIG. 1. The phase control signal Cj may be the first or second phase control signals Ci and Cq of FIG. 1.

The oscillator 210 may include an RLC resonator 211 and a compensation transconductor (Gm). The RLC resonator 211 may include a resistor (R), an inductor (L), and a capacitor (C). The free oscillation frequency and the resonance frequency of the oscillator 210 may be determined by a resistance value of the resistor R, an inductance value of the inductor L, and a capacitance value of the capacitor C. The compensation transconductor Gm may compensate for energy loss due to the resistance R of the RLC resonator 211 by the compensation signal Gj. The compensation transconductor Gm compensates for energy loss due to the resistance R to induce continuous oscillation of the oscillator 210. The resistance R may have a positive resistance value and the compensation transconductor Gm may have a negative resistance.

When each of the signals is expressed as vector signals having a magnitude and a phase, the sum of the converted input signal Sj, the phase control signal Cj, and the compensation signal Gj may be an output signal Tj. The output signal Tj may have a free oscillation frequency determined by the RLC resonator 211. The output signal Tj may be the first or second output signals Ti' and Tq' of FIG. 1.

3 is a phase diagram for illustratively explaining the operation of the circuit diagram of FIG. 2. FIG. 3 will be described with reference to FIG. 2.

The phase diagram of FIG. 3 represents the converted input signal Sj, the phase control signal Cj, the compensation signal Gj, and the output signal Tj of FIG. 2 as vectors. The phase diagram is based on the compensation signal Gj. The vector of the output signal Tj may be a sum of vectors of the converted input signal Sj, the phase adjustment signal Cj, and the compensation signal Gj. The phase A of the output signal Tj may be determined by the magnitudes of the converted input signal Sj, the phase adjustment signal Cj, and the compensation signal Gj. The phase A of the output signal Tj may be a value shifted from the silver input signal. The phase shifter according to the present invention may operate according to the principle of adjusting the phase of the output signal Tj by the magnitude of the phase control signal Cj. In this case, the magnitudes of the converted input signal Sj, the phase control signal Cj, and the compensation signal Gj may be fixed. The following description will be explained in accordance with the above principles.

4 are graphs for explaining a common synchronization range of frequencies set according to the phase shifter of FIG. 1. In FIG. 4, it is assumed that a common synchronization range of frequencies is set according to first to n-th phase shifters (n is a natural number greater than or equal to 2) having the same structure and function as the phase shifter 100 in FIG. 1. .

When a value for shifting the phase of the output signal compared to the input signal of each of the first to nth phase shifters is different, the free oscillation frequencies Fr1 to Frn of each of the first to nth phase shifters may be different. . Even if the synchronization ranges of the first to nth phase shifters are the same, an offset frequency (center frequency) determined according to the free oscillation frequencies Fr1 to Frn of each of the first to nth phase shifters may vary. As a result, as shown in FIG. 4, the common synchronization range of frequencies that the first to nth phase shifters have may be significantly reduced compared to the synchronization ranges of each of the first to nth phase shifters. In order to improve the distance resolution of the FMCW radar device, a signal having a wide bandwidth must be used, but it may be difficult to synchronize the frequency with an input signal having a wide bandwidth with a structure such as the phase shifter 100 of FIG. 1.

5 is a circuit diagram illustrating an exemplary configuration of a capacitor included in the circuit diagram of FIG. 2. 5 will be described with reference to FIGS. 1 and 2.

The circuit diagram of FIG. 5 may be referred to as a capacitor array 300. The capacitor array 300 may be discretely controlled switched capacitors. The capacitor array 300 may include a plurality of unit capacitors Cu0 to Cun-1 and a plurality of switches SW0 to SWn-1. The plurality of unit capacitors Cu0 to Cun-1 may have the same capacitance value, but is not limited thereto.

The capacitor C of FIG. 2 may be replaced with a capacitor array 300. The capacitor array measured between the VC node and the GND node according to the turn-on and turn-off control of the plurality of switches SW0 to SWn-1 respectively connected to the plurality of unit capacitors Cu0 to Cun-1 ( 300) can be adjusted and controlled. The capacitance value of the capacitor array 300 measured between the VC node and the GND node may be a total sum of capacitance values of unit capacitors respectively connected to the turned-on switches. As a result, the free oscillation frequency of the quadrature oscillator 120 including the capacitor array 300 may vary according to the capacitance value of the capacitor array 300 measured between the VC node and the GND node, that is, a plurality of switches ( The free oscillation frequency of the quadrature oscillator 120 may be adjusted according to the control of turn-on and turn-off of SW0 to SWn-1).

The plurality of switches SW0 to SWn-1 respectively connected to the plurality of unit capacitors Cu0 to Cun-1 may be controlled by separate control codes. The control codes may be included in the plurality of control codes CN of FIGS. 6A to 6C and 7 to 9, which will be described later.

6A to 6C are graphs exemplarily showing a synchronization process of the free oscillation frequency of the quadrature oscillator set according to the circuit diagram of FIG. 5 and the FMCW input signal. For example, signals are shown in a period between t0 and t2 in FIGS. 6A through 6C, but signals may be repeated in a period after t2 as well. 6A to 6C will be described based on FIGS. 1 and 5. 6A are graphs of a frequency (Ffmcw) of an input signal and a free oscillation frequency (Fr) of the quadrature oscillator 120 over time. 6B is a graph of the capacitance value Ctank of the capacitor array 300 measured between the VC node and the GND node over time. 6C is a timing diagram showing a plurality of control codes CN over time.

In Fig. 6A, it is assumed that a graph of the frequency (Ffmcw) of the input signal versus time has a sawtooth waveform by frequency modulation of the input signal by a radar device. The start of the input signal is t0, and the frequency (Ffmcw) of the input signal with respect to time has a Pulse Repetition Interval (PRI) period (T), and may have a constant slope. The minimum value of the frequency (Ffmcw) of the input signal may be Fmin and the maximum value may be Fmax.

In FIG. 6A, the free oscillation frequency Fr of the quadrature oscillator 120 may be determined according to the frequency of the sampled input signal. The waveform of the free oscillation frequency Fr may have the form of a step periodic function based on a time interval Tstep used to sample the frequency of the input signal. The modulation period of the free oscillation frequency (Fr) may be the same as the PRI period (T).

Referring to FIG. 6B, the free oscillation frequency Fr may have an outline of FIG. 6A by the capacitor array 300 having a capacitance value Ctank as shown in FIG. 6B. Specifically, when the capacitance value Ctank is Cmax, the free oscillation frequency Fr may be Fmin, and when the capacitance value Ctank is Cmin, the free oscillation frequency Fr may be Fmax. A process of adjusting the free oscillation frequency Fr of the quadrature oscillator 120 as shown in FIG. 6A will be described in detail with reference to FIG. 7.

Referring to FIG. 6C, the control code for controlling the capacitance value Ctank of the capacitor array 300 may be dynamically converted like a plurality of control codes CN including C0 to C4. Accordingly, the capacitance value Ctank of the capacitor array 300 may be dynamically converted as the frequency Ffmcw of the input signal is varied. Here, the dynamic conversion of the control code and the capacitance value Ctank may mean that the control code and the capacitance value Ctank change according to each of a plurality of time intervals. As an example, a plurality of time intervals may have the same unit length, which may be Tstep, but is not necessarily limited to the example.

The plurality of control codes CN and the capacitance values Ctank respectively corresponding to the plurality of time intervals may have different values. For example, during a first time interval in which the capacitance value Ctank is Cmax, the control code may be C0, and during a second time interval in which the capacitance value Ctank is Cmin, the control code may be C4.

When the free oscillation frequency of the quadrature oscillator 120 is constant, that is, when the capacitance value Ctank is not adjusted, in order to synchronize the frequency of the input signal Ffmcw and the frequency of the output signal, the quadrature oscillator 120 The injection synchronization frequency range may require a minimum frequency range of Fmin to Fmax. However, as the free oscillation frequency Fr of the quadrature oscillator 120 is shown as in the graph of FIG. 6A by the capacitor array 300 having a capacitance value Ctank as shown in FIG. 6B, the frequency of the input signal Ffmcw The minimum synchronization range for synchronizing the frequency of the output signal with Frlr can be reduced with Frlr. Consequently, since the output signal can be synchronized with respect to a wider frequency of the input signal, a wideband phase shifter can be implemented.

7 is a block diagram illustrating a phase shifter according to another embodiment of the present invention. The phase shifter 1000 may be used for beamforming of a radar device (not shown). The radar device may be a frequency modulated continuous wave radar device. The phase shifter 1000 may receive an input signal Sin from a radar device. The input signal Sin input to the phase shifter 1000 may be a modulated wave signal frequency modulated by a radar device. The phase shifter 1000 may output a final output signal Sout having a phase different from that of the input signal Sin by the phase shift, but having the same frequency as the input signal Sin. Referring to FIG. 7, the phase shifter 1000 includes a phase converter 1100, a quadrature oscillator 1200, a phase selector 1300, a control unit 1400, and first and second switches SW1 and SW2. It may include.

The phase converter 1100 may be substantially the same as the phase converter 110 of FIG. 1. However, the phase converter 1100 may receive the input signal Sin when the switch controller 1410 of the controller 1400 turns on the first switch SW1.

The quadrature oscillator 1200 may be substantially the same as the quadrature oscillator 1200 of FIG. 1. However, the quadrature oscillator 1200 is from the phase converter 1100 when the switch controller 1410 of the controller 1400 turns on the first switch SW1 and the phase converter 1100 receives an input signal. The first and second converted input signals Si and Sq may be received. When the switch controller 1410 of the control unit 1400 turns off the first switch SW1, the input signal Sin is not input to the phase converter 1100, so the quadrature oscillator 1200 sets the free oscillation frequency. Branches can output an output signal. A process in which the switch controller 1410 of the controller 1400 controls the first switch SW1 will be described later. The quadrature oscillator 1200 may include first and second mixers 1210 and 1220, and an in-phase oscillator 1230 and a quadrature oscillator 1240.

The free oscillation frequency of the quadrature oscillator 1200 may be synchronized with the frequency of the input signal Sin according to control codes CN provided from the controller 1400. The phases of the first and second output signals Ti' and Tq' output from the quadrature oscillator 1200 may be controlled by the phase control signal PC provided from the phase controller 1440 of the controller 1400. I can. The control codes CN provided from the controller 1400 and the phase control signal PC provided from the controller 1440 will be described later.

The first and second mixers 1210 and 1220 may be substantially the same as the first and second mixers 121 and 122 of FIG. 1. However, when the switch controller 1410 of the control unit 1400 turns on the first switch SW1, the first and second mixers 1210 and 1220 have the first and second converted input signals ( Si, Sq) can be received.

The in-phase oscillator 1230 and the quadrature oscillator 1240 may be substantially the same as the in-phase oscillator 123 and the quadrature oscillator 124 of FIG. 1. The in-phase oscillator 1230 and the quadrature oscillator 1240 may adjust the free oscillation frequency of the quadrature oscillator 1200 according to control codes CN provided from the controller 1400. The in-phase oscillator 1230 and the quadrature oscillator 1240 can and control the first and second phase control signals Ci and Cq according to the phase control signal PC provided from the control unit 1400. have. Phases of the first and second output signals Ti' and Tq' may be determined by the first and second phase control signals Ci and Cq.

The phase selector 1300 may be substantially the same as the phase selector 130 of FIG. 1. However, when the switch controller 1410 of the control unit 1400 turns on the first switch SW1, the phase selector 1300 receives a final output signal Sout having a frequency synchronized with the input signal Sin. Can be printed.

The controller 1400 may synchronize the free oscillation frequency of the quadrature oscillator 1200 with the frequency of the input signal Sin. The controller 1400 may synchronize the frequency of the final output signal Sout with the frequency of the input signal Sin and control the phase of the final output signal Sout. The control unit 1400 may include a switch controller 1410, a frequency locked loop (FLL) 1420, a lookup table 1430, a phase controller 1440, and a second switch SW2. The controller 1400 may receive setting data necessary for frequency synchronization from an external device. In this case, the radar device may determine the setting data in advance. Alternatively, the controller 1400 may determine and store setting data necessary for frequency synchronization in advance. The setting data includes the operation mode of the phase shifter 1000, the start time of the input signal Sin, the slope of the input signal Sin, the period of the input signal Sin (eg, the PRI period in FIG. 6A), and sampling. For example, it may include a time interval (Tstep of FIG. 6A).

The switch controller 1410 may control the first switch SW1, the second switch SW2, and the third switch SW3. The switch controller 1410 may turn on the first switch SW1 and the phase converter 1100 may receive the input signal Sin according to the turn-on of the first switch SW1. In the switch controller 1410, the second switch SW2 connects the quadrature oscillator 1200 to the frequency locked loop 1420, or the second switch SW2 connects the quadrature oscillator 1200 to the look-up table 1430. The second switch SW2 may be controlled to be connected. When the first switch SW1 is turned off and the second switch SW2 connects the quadrature oscillator 1200 to the frequency locked loop 1420, the phase shifter 1000 is set to a calibration mode. It can work. On the other hand, when the first switch SW1 is turned on and the second switch SW2 connects the quadrature oscillator 1200 to the lookup table 1430, the phase shifter 1000 is in the FMCW mode (frequency modulation continuous wave). mode).

The switch controller 1410 controls the third switch SW3 so that the controller 1400 receives an input signal Sin or the controller 1400 receives a signal Sref having a lower frequency than the input signal Sin. You can choose to do it. That is, the third switch SW3 inputs the frequency lock loop 1420 so that the frequency lock loop 1420 receives and samples a signal Sref having a lower frequency than the input signal Sin instead of the input signal Sin. A terminal supplying a signal Sref having a lower frequency than the signal Sin may be connected. When the input signal Sin is a high frequency, the operation of the frequency locked loop 1420 may be unstable. That is, it may be difficult for the frequency locked loop 1420 to operate according to the frequency of the input signal Sin. Accordingly, since the control unit 1400 receives the signal Sref having a lower frequency than the input signal Sin, the frequency lock loop 1420 may operate normally. The signal Sref having a lower frequency than the input signal Sin may have a frequency obtained by dividing the frequency of the input signal Sin by N (N is a natural number greater than 1). For convenience of explanation, the following description will be based on a case where the controller 1400 receives the input signal Sin.

The frequency locked loop 1420 may generate control codes. The frequency locked loop 1420 may dynamically convert control codes generated as the frequencies of the input signal are varied so that the sampled frequencies and the free oscillation frequencies of the quadrature oscillator 1200 are the same. The frequency locked loop 1420 may store dynamically converted control codes CN in a lookup table. The frequency locked loop 1420 may sample the frequency of the input signal Sin according to a time interval for sampling included in the setting data. The frequency locked loop 1420 may monitor the quadrature oscillator 1200 operating at a free oscillation frequency in the measurement mode. The frequency locked loop 1420 includes control codes based on the sampled frequency of the input signal, the monitored output signal Smon, and the start time of the input signal Sin included in the setting data, and the period of the input signal Sin. (CN) can be created. That is, the frequency locked loop 1420 may generate control codes CN for adjusting the free oscillation frequency of the quadrature oscillator 1200 according to the sampled frequency of the input signal.

The lookup table 1430 may provide control codes CN to the quadrature oscillator 1200 in the FMCW mode. That is, the lookup table 1430 may dynamically input the control codes CN to the quadrature oscillator 1200. In this case, the lookup table 1430 may provide control codes CN corresponding to a plurality of time intervals. The quadrature oscillator 1200 may change or adjust the free oscillation frequency according to the control codes CN. The quadrature oscillator 1200 includes first and second output signals Ti' and Tq' having a frequency synchronized with the sampled frequency of the input signal based on the free oscillation frequency adjusted according to the control codes CN. Can be created. Here, the quadrature oscillator 1200 may selectively use a plurality of control codes CN. For example, when the control codes CN stored in the lookup table 1430 include the first to fifth control codes C0 to C4, the quadrature oscillator 1200 is 5 The free oscillation frequency can be synchronized with the sampled frequency of the input signal using only the control codes C0, C2, and C4. In this case, the free oscillation frequency may not be equal to the sampled frequency of the input signal in all time ranges. Nevertheless, when the frequency of the input signal is within the synchronization range of the quadrature oscillator 1200, the free oscillation frequency may be synchronized with the frequency of the input signal.

The phase controller 1440 may provide a phase control signal PC to the quadrature oscillator 1200. The phase controller 1440 may control phases of the first and second output signals Ti' and Tq' output from the quadrature oscillator 1200 according to the phase control signal PC. As a result, the final output signal Sout based on the first and second output signals Ti' and Tq' may have a phase shifted from the phase of the input signal Sin.

FIG. 8 is a circuit diagram showing the phase shifter of FIG. 7 in more detail. 8 will be described in conjunction with FIGS. 2, 5, and 7. In FIG. 8, the phase shifter 2000 may be substantially the same as the phase shifter 1000 of FIG. 7. In FIG. 8, for convenience of description, the first and second switches SW1 and SW2 of FIG. 7 and the control unit 1400 are omitted. The phase shifter 2000 may be interpreted as including the first and second switches SW1 and SW2 of FIG. 7 and the control unit 1400. The phase shifter 2000 may include a phase converter 2100, a quadrature oscillator 2200, and a phase selector 2300.

The phase converter 2100 and the phase selector 2300 may be substantially the same as the phase converter 110 and the phase selector 130 of FIG. 1, respectively. Alternatively, the phase converter 2100 and the phase selector 2300 may be substantially the same as the phase converter 1100 and the phase selector 1300 of FIG. 7, respectively. Although not shown in FIG. 8, the phase converter 2100 may be connected to a switch like the phase converter 1100 of FIG. 7.

The quadrature oscillator 2200 may correspond to the quadrature oscillator 1200 of FIG. 7. The quadrature oscillator 2200 includes first and second input transconductors 2211 and 2214, first and second phase transconductors 2212 and 2213, an inverter 2215, an in-phase oscillator 2221, and It may include a rectangular phase oscillator 2222.

The first and second input transconductors 2211 and 2214 may transmit the first and second converted input signals Si and Sq to the in-phase oscillator 2221 and the quadrature oscillator 2222, respectively. The first and second phase transconductors 2212 and 2213 may adjust the first and second phase control signals Ci and Cq. Although not shown in FIG. 8, the first and second phase transconductors 2212 and 2213 may receive a phase control signal (eg, from the phase controller 1440 of FIG. 7) and are controlled by the phase control signal. The first and second phase control signals Ci and Cq may be adjusted. The first and second phase transconductors 2212 and 2213 may transmit the first and second phase control signals Ci and Cq to the in-phase oscillator 2221 and the quadrature oscillator 2222, respectively.

The in-phase oscillator 2221 and the quadrature oscillator 2222 may include compensation transconductors Gmi and Gmq, and RLC resonators 2222_1 and 2, respectively. The in-phase oscillator 2221 and the quadrature oscillator 2222 may correspond to the in-phase oscillator 1230 and the quadrature oscillator 1240 of FIG. 7, respectively. Components and signals of the in-phase oscillator 2221 and the quadrature oscillator 2222 may correspond to the components of FIG. 2. That is, the compensation transconductors Gmi and Gmq are in the compensation transconductor Gm of FIG. 2, and the resistors Ri and Rq, the inductors Li and Lq, and the capacitors Ci and Cq are respectively It may correspond to the resistance (R) of 2, the inductor (L), and the capacitor (C). Compensation signals (Gi, Gq), converted input signals (Si, Sq), phase control signals (Ci, Cq), and output signals (Ti', Tq') are respectively a compensation signal (Gj) and converted It may correspond to an input signal Sj, a phase control signal Cj, and an output signal Tj.

Although not shown in FIG. 8, capacitance values of the capacitors Ci and Cq may be changed by the control codes CN of FIG. 7. The capacitors Ci and Cq may be composed of the capacitor array 300 of FIG. 5, and in this case, turn-on and turn-on of a plurality of switches respectively connected to a plurality of unit capacitors according to the control codes CN. Off can be controlled and the free oscillation frequency of the quadrature oscillator 2200 can be synchronized with the frequency of the input signal Sin.

9 is a flowchart illustrating an exemplary phase shifting method for beamforming of a radar device according to an embodiment of the present invention. FIG. 9 will be described with reference to FIG. 7.

In step S110, the frequency locked loop 1420 may sample the frequency of the input signal Sin received by the phase shifter 1000 according to a plurality of time intervals. In this case, a plurality of time intervals may be stored in the controller 1400 or included in setting data input from the outside. As an example, a plurality of time intervals may have the same length.

In step S120, the frequency locked loop 1420 may generate a plurality of control codes CN for adjusting the free oscillation frequency of the quadrature oscillator 1200 according to the sampled frequency of the input signal. The control codes CN of can be dynamically converted. The plurality of control codes CN may correspond to a plurality of time intervals. The frequency locked loop 1420 may monitor the quadrature oscillator 1200 operating at a free oscillation frequency in the measurement mode. The frequency locked loop 1420 controls a plurality of controls based on the sampled frequency of the input signal, the monitored output signal Smon, the start time of the input signal Sin included in the setting data, and the period of the input signal Sin. Codes CN can be generated. The frequency locked loop 1420 may store a plurality of generated control codes CN in a lookup table.

In step S130, the control unit 1400 dynamically inputs a plurality of control codes CN corresponding to the frequency of the input signal to the quadrature oscillator 1200 according to a plurality of time intervals. The value of the capacitor in) can be adjusted. Accordingly, the synchronization area of the quadrature oscillator 1200 may be extended. As a result, the quadrature oscillator 1200 may output a signal synchronized with the frequency of the input signal.

The lookup table 1430 may provide a plurality of control codes CN to the quadrature oscillator 1200 in the FMCW mode. In this case, the lookup table 1430 may provide each of the plurality of control codes CN to the quadrature oscillator 1200 during a corresponding time interval. For example, a first control code among the plurality of control codes CN may correspond to a first time interval, and the lookup table 1430 includes the first control code during the first time interval, and the quadrature oscillator 1200 Can be provided. The quadrature oscillator 1200 may change or adjust a free oscillation frequency according to a plurality of control codes CN. The quadrature oscillator 1200 generates first and second output signals Ti'Ti' having a frequency synchronized with a frequency of an input signal based on a free oscillation frequency adjusted according to a plurality of control codes CN. Can be generated.

In step S140, the phase controller 1440 may control the phase of the output signal. Referring to FIG. 8, the phase controller 1440 provides a phase control signal PC to the quadrature oscillator 1200 to provide first and second output signals Ti' Tq' output from the quadrature oscillator 1200. ) Phases can be controlled.

The phase selector 1300 may receive an output signal from the quadrature oscillator 1200. The phase selector 1300 includes first and second output signals Ti' and Tq' positioned in a first quadrant on a phase diagram, and phase inversion signals of the first and second output signals Ti' and Tq'. By selecting one of (~Ti', ~Tq'), the final output signal Sout can be output. When the output signal is located in the first to fourth quadrants on the phase diagram as shown in FIG. 3, the phase selector 1300 may select the output signal by selecting the output signal located in the first quadrant on the phase diagram. The phase selector 130 may output the final output signal Sout by selecting the first and second output signals Ti' and Tq' located in the first quadrant on the phase diagram. In the end, the final output signal Sout has the same frequency as the input signal Sin by frequency locking of the quadrature oscillator 120, but the input signal Sin has a different phase due to phase shift. I can.

The contents described above are specific examples for carrying out the present invention. The present invention will include not only the embodiments described above, but also embodiments that can be changed in design or easily changed. In addition, the present invention will also include techniques that can be easily modified and implemented in the future using the above-described embodiments.

100, 1000, 2000: phase shifter
110, 1100, 2100: phase converter
120, 1200, 2200: quadrature oscillator
130, 1300, 2300, phase selector

Claims (16)

Free oscillation of the sampled frequencies and a quadrature oscillator by sampling the frequencies of the input signal, generating control codes according to the sampled frequencies, and dynamically converting the generated control codes A frequency locked loop (FLL) that makes frequencies equal;
The quadrature oscillator generating an output signal having frequencies synchronized with the frequencies of the input signal based on the free oscillation frequencies adjusted according to the dynamically converted control codes; And
Phase shifter comprising a phase controller for controlling the phase of the output signal. The method of claim 1,
The quadrature oscillator includes a resonator, and
The frequency locked loop is a phase shifter that controls the free oscillation frequencies of the quadrature oscillator by controlling the resonator according to the dynamically converted control codes. The method of claim 2,
The resonance unit includes a plurality of unit capacitors, and
The frequency locked loop is a phase shifter for changing the free oscillation frequencies by controlling turn-on and turn-off of a plurality of switches respectively connected to the plurality of unit capacitors according to the dynamically converted control codes. The method of claim 1,
The quadrature oscillator includes a plurality of transconductors, and
The phase controller is a phase shifter for controlling the phase of the output signal by controlling each of the plurality of transconductors by a phase control signal. The method of claim 1,
The phase shifter further comprising a phase selector for selecting one of the output signal and the phase inversion signal of the output signal. The method of claim 1,
A phase shifter further comprising a lookup table for storing the dynamically converted control codes. The method of claim 6,
The phase shifter further comprises a switch connecting the quadrature oscillator to the frequency locked loop or connecting the quadrature oscillator to the lookup table. The method of claim 1,
Further comprising a phase converter for converting the phase of the input signal, and
A phase shifter further comprising a switch connecting the quadrature oscillator and the phase converter so that the quadrature oscillator receives the converted input signal. The method of claim 1,
Further comprising a switch connecting the frequency-locked loop and the first signal supply terminal so that the frequency-locked loop receives a first signal having a lower frequency than the input signal instead of the input signal and samples the frequency of the first signal. Phase shifting. Sampling frequencies of an input signal received by a phase shifter by a frequency locked loop (FLL);
Generating a plurality of control codes to adjust free oscillation frequencies of a quadrature oscillator according to the sampled frequencies by the frequency locked loop, and dynamically converting the generated plurality of control codes ;
Converting the phase of the input signal by a phase converter;
Generating an output signal having frequencies synchronized with the frequencies of the converted input signal based on the free oscillation frequencies adjusted according to the plurality of dynamically converted control codes by the quadrature oscillator ;
Controlling the phase of the output signal by a phase controller; And
And outputting the output signal by the quadrature oscillator. The method of claim 10,
The quadrature oscillator includes a resonator, and
The step of generating the output signal comprises adjusting the free oscillation frequencies by controlling the resonator according to the plurality of dynamically converted control codes. The method of claim 11,
The resonance unit includes a plurality of unit capacitors, and
The generating of the output signal includes controlling turn-on and turn-off of a plurality of switches respectively connected to the plurality of unit capacitors according to the plurality of control codes dynamically converted by the frequency locked loop. The phase shifting method further comprising adjusting free oscillation frequencies. The method of claim 10,
The phase shifter includes a plurality of transconductors, and
Controlling the phase of the output signal comprises controlling the phase of the output signal as controlling the plurality of transconductors by the phase controller. The method of claim 10,
The phase shifter comprises a phase selector, and
The step of outputting the output signal comprises selecting one of the output signal and a phase inversion signal of the output signal by the phase selector. The method of claim 10,
The phase shifter includes a lookup table, and
The step of dynamically converting the generated plurality of control codes includes dynamically converting the generated plurality of control codes and then storing the dynamically converted plurality of control codes in the lookup table by the frequency lock loop. Phase shift method comprising a. The method of claim 10,
The step of dynamically converting the generated plurality of control codes includes generating the plurality of control codes for adjusting the free oscillation frequency according to the sampled input signal by monitoring the quadrature oscillator, and generating the plurality of control codes A method of phase shifting comprising dynamically transforming codes.

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