KR20120121185A - Apparatus and method for equalizing a signal using canceling interference in a wiless communication system - Google Patents

Abstract

PURPOSE: An equalization apparatus and method thereof are provided to promote the reliability of equalization by equalizing a signal which interference is eliminated in the received signal. CONSTITUTION: The equalization unit of a receiver estimates a transmission signal by equalizing a received signal(703). The interference elimination signal generator of the receiver generates the received signal which interference is eliminated between transmission signals using the estimated transmission signal(705). The equalization unit creates x' and x" in order to independently equalize the signal which the interference is eliminated. The equalization unit outputs x' using a linear addition method. The equalization unit generates the estimated transmission signal(707). [Reference numerals] (701) Receiving a signal(Y) from a transmitter; (703) Estimating a transmission signal(X^) by equalizing the received signal; (705) Generating signals(Y',Y") from which interference between transmission signals is eliminated using the estimated transmission signal; (707) Equalizing Y' and Y" independently to produce X' and X" and adding to generate the finally estimated transmission signal; (AA) Start; (BB) End

Description

Equalization device and method using interference cancellation in wireless communication system {APPARATUS AND METHOD FOR EQUALIZING A SIGNAL USING CANCELING INTERFERENCE IN A WILESS COMMUNICATION SYSTEM}

The present invention is directed to interference cancellation in a wireless communication system. More particularly, the present invention relates to a method and apparatus for equalizing a received signal using interference cancellation in a wireless communication system.

The High Speed Packet Access (HSPA) system of the 3rd Generation Partnership Project (3GPP) provides high-order modulation (HOM) with 64-QAM (Quadrature Amplitude Modulation) and 2x2 multiple-input and It is characterized by the use of multiple-output systems.

The HSPA system is characterized by the use of a data channel (HS-PDSCH) having a small spreading factor (SF) compared to a WCDMA mobile communication system. Symbols with a large SF could effectively suppress intersymbol interference (ISI) or inter-cell interference (ICI) through de-spreading of the receiver. In case of SF symbol, it is fundamentally difficult to protect the symbol by only reverse spreading. Therefore, a method of estimating a transmission signal using an equalizer instead of a commonly used rake reception method is used.

However, the equalizer has an advantage of effectively inversion of the channel matrix. However, when the interference signal is large, the inversion performance deteriorates and the reliability of the estimated transmission signal is lowered. For this reason, performing equalization while properly processing interference signals has been an important research issue for increasing reception performance. Meanwhile, techniques for performing signal detection while eliminating interference in a CDMA system or a MIMO system using interference cancellation have been studied in the past, but are not suitable for application to an HSPA system that suffers from frequency selective fading. Therefore, there is a request for an equalizer for interference cancellation.

The present invention provides an equalization method and apparatus for improving reception performance in a MIMO system.

The present invention provides a method and apparatus for removing interference in a received signal generated by mutually independent transmission signals in a MIMO system.

The present invention provides a method and apparatus for minimizing an approximation error when approximating an exponential function to a linear function in a process of cleaning equalized symbols in a receiver in a MIMO system.

The method of the present invention is characterized in that the first channel data and the first channel data are spread using a first spreading coefficient (SF) in a multiple input multiple output (MIMO) wireless communication system comprising at least two transmit antennas and at least two receive antennas. CLAIMS 1. A method of equalizing a received signal including at least one of second channel data spread using two spreading coefficients, the method comprising: estimating a transmitted signal by equalizing the received signal; A channel determining process of determining whether the estimated transmission signal is the first channel data or the second channel data; If the estimated transmission signal is the first channel data, removing the interference according to the characteristics of the first channel; If the estimated transmission signal is the second channel data, removing the interference according to the characteristics of the second channel.

The apparatus of the present invention is characterized in that the first channel data and first channel data are spread using a first spreading coefficient (SF) in a multiple input multiple output (MIMO) wireless communication system comprising at least two or more transmit antennas and at least two receive antennas. An apparatus for equalizing a received signal including at least one of second channel data spread using two spreading coefficients, the apparatus comprising: an equalizing unit for equalizing the received signal and estimating a transmission signal; A channel determination unit determining whether the estimated transmission signal is the first channel data or the second channel data; A first channel signal processor to remove interference according to the characteristics of the first channel if the estimated transmission signal is the first channel data; And a second channel signal processor for removing interference according to the characteristics of the second channel if the estimated transmission signal is the second channel data.

Representative effects of the configuration of the present invention are as follows.

The present invention can generate a signal which eliminates interference in a received signal generated by mutually independent transmission signals in a MIMO system, and equalizes the signal from which the interference is removed to improve the reliability of equalization, thereby improving reception performance. In addition, the present invention can reduce the approximation error when approximating the exponential function to a linear function in the process of cleaning the equalized symbol.

In addition, when the received signal includes the HS channel and the R99 channel signal, it is possible to determine which channel signal for each code channel and to increase reception performance by individually processing interference according to the determined channel signal.

1 is a diagram illustrating the structure of a 2x2 MIMO transmitter in an HSPA system;
2 is a diagram illustrating an equalization process of a channel and a receiver between a transmitter and a receiver in a 2x2 MIMO transmitter in an HSPA system;
3 is a scatter plot showing the output of an LMMSE equalizer when there is interference between transmission signals;
4 is a scatter plot showing the output of the LMMSE equalizer when there is no interference between the transmitted signals;
5 is a diagram illustrating a configuration of an LMMSE equalizer in a receiver of a 2X2 MIMO transceiving system according to a first embodiment of the present invention;
6 is a view for explaining the configuration of the interference cancellation signal generator 520 of FIG.
7 is a diagram for explaining an equalization method in a receiver according to the first embodiment of the present invention;
8 is a view for explaining step 705 of FIG. 7 in detail;
9 is a diagram illustrating the contribution of each constellation point to the nonlinear MMSE estimate in the constellation of 16-QAM modulation scheme;
FIG. 10 is a diagram illustrating an approximation result using a Taylor series of an exponential function. FIG.
FIG. 11 is a view for explaining a method of dividing an exponential function by sections in order to approximate an exponential function to a linear function according to the first embodiment of the present invention; FIG.
12 is a view for explaining a process of determining an approximated linear function for an exponential function in a corresponding interval according to the first embodiment of the present invention;
FIG. 13 is a diagram for explaining a tree structure of an OVSF code that is a channelization code of a WCDMA downlink; FIG.
14 is a diagram illustrating a configuration of an LMMSE equalizer in a receiver of a 2X2 MIMO transceiving system according to a second embodiment of the present invention.
FIG. 15 is a diagram illustrating a configuration of the HS channel signal processing unit 1421 of FIG. 14.
FIG. 16 is a view for explaining the configuration of the non-HS channel signal processing unit 1425 of FIG. 14;
17 is a view for explaining an equalization method according to a second embodiment of the present invention;
18 is a diagram showing simulation results according to the second embodiment of the present invention.

Hereinafter, exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings. It is to be noted that the same elements among the drawings are denoted by the same reference numerals whenever possible. Also, detailed descriptions of well-known functions and configurations that may unnecessarily obscure the subject matter of the present invention will be omitted.

In the following description, the signal processing will be described based on the signal in the frequency domain unless otherwise specified as a time domain signal. However, this is merely for convenience of description, and the signal processing of the present invention may be performed in the frequency domain or the time domain.

1 is a diagram illustrating the structure of a 2x2 MIMO transmitter in an HSPA system.

Referring to FIG. 1, each data that has undergone two independent transport channel processing (101, 103) is subjected to precoding matrix after spreading and scrambling (103, 104) The weight according to the matrix is multiplied. The weight is generated in the weight generator 120, and the precoding matrix is defined by Equation 1.

For reference, in Equation 1, w may have four kinds of values, and a precoding matrix having the largest energy among four signals composed of a combination of a channel and all possible precoding matrices at the receiver may be used. And the receiver is defined in the HSPA standard to send the selected precoding matrix to the transmitter.

The precoded signals are summed alternately in summers 109 and 110 and transmitted to the receiver through a total of four wireless channels through transmit antennas 113 and 114. Hereinafter, an equalization process of a channel and a receiver between a transmitter and a receiver will be described.

2 is a diagram illustrating an equalization process of a channel and a receiver between a transmitter and a receiver in a 2x2 MIMO transmitter in an HSPA system.

Referring to FIG. 2, X1 and X2, which are independent transmission signals output from the transmitter 200, are output through the transmission antenna 1 201 and the transmission antenna 2 202 and are over a wireless channel represented by a channel matrix H. Transmitted to receiver 210.

The channel matrix H is composed of H ₁ , H ₂ , H ₃ , and H ₄ , where H ₁ represents a channel between transmit antenna 1 201 and receive antenna 1 214, and H ₂ represents transmit antenna 1 201. ) Means a channel between transmit antenna 2 (215) and H ₃ denotes a channel between transmit antenna 2 (202) and receive antenna 1 (214), and H ₄ denotes a transmit antenna transmit antenna 2 (202) and receive antenna It means a channel between two (215).

The receiver 210 receives signals Y ₁ and Y ₂ through the receiving antenna 1 214 and the receiving antenna 2 215, respectively, and the received signals Y ₁ and Y ₂ are the LMMSE equalizer 111 and the MIMO decoder 113. Is input to the receiver.

In FIG. 2, the reception signals Y ₁ and Y ₂ may be represented by Equation 2 below.

N ₁ denotes a noise signal in the received signal Y ₁ , and N ₂ denotes a noise signal in the received signal Y ₂ .

When Equation 2 is expressed as a discrete-time signal, Equation 3 is expressed.

Meanwhile, the LMMSE equalizer 211 of the receiver 210 estimates a transmission signal using cross-correlation between the received signal and the desired signal and auto-correlation between the received signals. Output

Specifically, the LMMSE equalizer 111 is designed to minimize the mean square error (MSE) between the transmission signal and the equalizer output signal, where MSE is defined as in Equation 4. .

The MMSE weight that satisfies Equation 4 is expressed as a product of cross-correlation and auto-correlation inverse matrix as shown in Equation 5 below.

Therefore, the output of the equalizer in the 2x2 MIMO system is represented by Equation 6 below.

)을 살펴보면,

가 포함되어 있다. 한편,

에서 상기 <수학식 2>와 같이 송신 스트림 X₁의 관점에서는 송신 스트림 X₂의 신호가 간섭으로 작용하고, 송신 스트림 X₂의 관점에서는 송신 스트림 X₁의 신호가 간섭으로 작용하게 된다.On the other hand, the output of the conventional LMMSE equalizer (

),

Is included. Meanwhile,

In view of the <Equation 2> X ₁ and the transmission stream transmitted stream of the transmission signal X is X ₂ stream acts as interference from the viewpoint, and the _second as a signal of a transmission stream, X ₁ is acting as an interference in.

In general, however, the distance between the MIMO transmitting antennas is much smaller than the distance between the transceivers in a mobile communication environment. In this case, according to the plane wave theory, the channel experienced by the signal output from each transmitting antenna is known to have the same phase difference only according to the incident angle of the plane wave. The same channel experienced by the output signal of the transmitting antenna may occur even when the correlation between the receiving antennas is high.

As a result, in this environment, the H ₁ and H ₂ channels are almost identical, and the H ₃ and H ₄ channel values are almost identical. Therefore, Equation 2 may be expressed by Equation 7 below.

In this environment where the H ₁ and H ₂ channels are almost the same and the H ₃ and H ₄ channels are almost the same, the inversion of the autocorrelation matrix of the LMMSE has a large error. The error rate of the output signal of the equalization is also very large. Therefore, the LMMSE equalizer output signal in the case where there is interference between the transmission signals is much larger in the estimation of the transmission signal than the LMMSE equalizer output signal in the absence of mutual interference.

3 is a scatter plot showing the output of the LMMSE equalizer when there is interference between transmission signals, and FIG. 4 is a scatter plot showing the output of the LMMSE equalizer when there is no interference between transmission signals. plot). 4 shows that there is less error in the estimation of the transmission signal than in FIG. Thus, there is a need to remove or reduce inter-stream interference to reduce the error rate of the LMMSE equalizer output.

&Lt; Embodiment 1 >

According to this need, the first embodiment of the present invention generates a received signal from which mutual interference has been removed and uses it for equalization. The basic concept of the first embodiment of the present invention will be briefly described.

The first embodiment of the present invention estimates a transmission signal by equalizing the reception signal in Equation (7), and generates a plurality of reception signals from which interference is removed from the reception signal using the estimated transmission signal, Independently equalize the plurality of received signals from which the interference has been removed and combine them to estimate a transmission signal from which the interference has been removed. That is, a diversity gain can be obtained at the same time as estimating a transmission signal having high reliability by generating received signals from which interference has been removed, and adding and equalizing them. On the other hand, by repeatedly performing this process it is possible to further improve the quality of the received signal.

Although the following description of the present invention has been described using a 2X2 MIMO transmission / reception system as an example, this is merely for convenience of description and may be equally applied to other multiple transmission / reception systems other than 2X2.

5 is a diagram illustrating a configuration of an LMMSE equalizer in a receiver of a 2X2 MIMO transceiving system according to a first embodiment of the present invention.

The equalizer 500 of the present invention includes an equalizer 510 and an interference cancellation signal generator 520.

',

''은 각각 (

₁',

₂')과 (

₁'',

₂'')로 구성된다.In FIG. 5, the input signal Y is composed of (Y ₁ , Y ₂ ) received by two antennas. In addition, the outputs Y 'and Y''of the interference cancellation signal generator 520 are composed of (Y ₁ ', Y ₂ ') and (Y ₁ '', Y ₂ ''), respectively. In addition, the output of the first LMMSE equalizer 511

',

'' Is each (

₁ ',

₂ ') and (

₁ ``,

₂ '').

The received signal Y is input to the equalizer 510 and the interference cancellation signal generator 520.

)를 간섭 제거 신호 생성부(520)로 전달한다.The equalizer 510 equalizes the received signal Y using one of the first LMMSE equalizer 511 or the second LMMSE equalizer 513 to estimate a transmission signal, and estimates the estimated transmission signal (

) Is transmitted to the interference cancellation signal generator 520.

)를 이용하여 서로 다른 안테나에서 생성된 송신 신호들 상호 간 간섭을 제거한 신호들(Y', Y'')를 생성한다. 이후, 상기 Y'과 Y''은 등화부(510)로 입력되며, 제1 LMMSE 등화기(511)와 제2 LMMSE 등화기(513) 각각에서 독립적으로 등화되어

',

''이 출력된다. The interference cancellation signal generator 520 may transmit the estimated transmission signal in the received signal Y (

) To generate the signals (Y ', Y'') to remove the interference between the transmission signals generated from different antennas. Then, Y 'and Y''are input to the equalizer 510 and are independently equalized in each of the first LMMSE equalizer 511 and the second LMMSE equalizer 513.

',

Is printed.

'과

''는 가산기(515)에서 선형 가산 되어

'''이 출력된다. 상기

'''는 최종적으로 추정된 송신 신호가 된다. 이렇게 본 발명의 제1 실시예에서는 송신 신호들 상호 간에 의한 간섭이 제거된 복수 개의 수신 신호들을 생성하고 이를 독립적으로 등화한 이후 등화된 신호들을 선형 가산하여

'''을 출력하여, 등화기의 출력, 즉, 추정된 송신 신호의 신뢰성을 높일 수 있다. 한편, 상기

'''를 다시 간섭 제거 신호 생성부(520)으로 입력하여 상기 동작들을 반복하면 추정된 송신 신호의 신뢰성은 더 높아진다.After that

'and

'' Is linearly added in adder 515

'''Is printed. remind

'''Becomes the finally estimated transmission signal. As described above, in the first embodiment of the present invention, a plurality of received signals from which interference between transmission signals are removed are generated and equalized independently, and then linearized addition of the equalized signals.

By outputting ''', the output of the equalizer, that is, the reliability of the estimated transmission signal can be improved. On the other hand,

Inputting '''back to the interference cancellation signal generator 520 and repeating the above operations increases the reliability of the estimated transmission signal.

Y 'is represented by Equation (8), and Y' 'is represented by Equation (9).

On the other hand, since the transmission signal estimated by the LMMSE appears in the form of attenuation of the original transmission signal, Equation 8 and Equation 9 are represented by Equation 10 when the Equation 8 is expressed again as a matrix-vector model.

In Equation 10, the attenuation of the transmission signal is expressed by substituting the attenuation of the channel matrix. By redesigning the filter coefficients of the equalizer after the interference is removed according to Equation 10, the performance of the equalization can be improved.

을

'으로 치환하면 되고, 하기 <수학식 11>에서 Y를 Y''으로, Y₁을 Y₁'', Y₂를 Y₂''으로 치환하고,

을

''으로 치환하면 된다.On the other hand, the result of equalizing Y 'and the result of equalizing Y''are expressed based on Equation 11 below. That is, substituted by the following <Equation 11> Y to Y 'as, for Y ₁ Y _1', a Y ₂ Y ₂ 'in,

of

And Y, Y ₁ to Y ₁ '', Y ₂ to Y ₂ '' in Equation 11 below,

of

Replace with ''.

FIG. 6 is a diagram illustrating a configuration of the interference cancellation signal generator 520 of FIG. 5.

The interference cancellation processor 520 includes a symbol generator 610, a symbol cleaner 620, a virtual transmitter 630, a virtual channel unit 640, and an interference cancellation processor 650.

로부터 심볼을 생성한다. 상기 심볼 생성부(610)는 심볼을 생성하기 위하여

에 상기 도 1에서 설명한 송신기의 신호 처리 절차의 역 절차를 적용한다. 즉,

에 역 프리코딩 행렬을 곱한 이후, 역 스크램블링을 수행하고, 역 확산을 수행하여 심볼을 생성한다. 상기 역 확산은 역 하다마드(Hadamard) 변환을 통하여 수행될 수 있고, 상기 역 스크램블링은 PN 코드를 곱하여 수행되는 것이 일반적이다.The symbol generator 610 estimates the transmitted signal transmitted from the interference cancellation signal generator 520.

Create a symbol from. The symbol generator 610 generates a symbol

The reverse procedure of the signal processing procedure of the transmitter described above with reference to FIG. In other words,

After multiplying the inverse precoding matrix by, inverse scrambling is performed and inverse spreading is performed to generate a symbol. The inverse spreading may be performed through an inverse Hadamard transformation, and the inverse scrambling is generally performed by multiplying the PN code.

The symbols generated by the symbol generator 610 are multiplied by the gain g of the code channel and the symbol s, and are represented by Equation 12 below.

The symbol cleaner 620 may perform symbol cleaning on the generated symbols to generate more reliable symbols than the symbols generated by the symbol generator 610. A first embodiment of the present invention proposes a method for efficiently performing symbol cleaning, which will be described later with reference to FIG. 9.

The virtual transmitter 630 applies the signal processing procedure of the transmitter of FIG. 1 to the symbol on which the cleaning is performed. That is, after spreading and scrambling are performed on the symbols, the precoding matrix is multiplied. The spreading may be performed through a Hadamard transform, and the scrambling is generally performed by multiplying a PN code.

On the other hand, the virtual channel unit 640 is a channel to the output signal of the virtual transmitter 630 in order for the output signal of the virtual transmitter 630 to experience the channel (H) between the transmitter 200 and the receiver 210. Multiply the function. If the signal is processed in the time domain, convolution of the channel function will be performed.

'''을 출력한다.The interference cancellation processor 650 outputs Y 'of Equation 8 and Y''of Equation 9 by using the output signal of the virtual channel unit 640 in the received signal Y. It generates reliable received signals with interference between transmission signals removed. Since Y 'and Y''are input to the equalizer 510 as described in FIG. 5 and independently equalized, linear addition is performed.

Print '''.

)의 신뢰성을 더욱 향상시키기 위한 구성들이다. 따라서 시스템의 복잡성을 감소시키고자 하는 경우에 상기 801 내지 807 단계는 생략될 수도 있다. 이 경우 가상 채널부(640)에 입력되는 신호는 수신 신호(Y)가 등화부(510)에서 등화되어 출력된 신호(

)가 될 것이다.Meanwhile, the symbol generator 610, the symbol cleaner 620, and the virtual transmitter 630 are equalized and estimated by the equalizer 510.

) To further improve the reliability. Therefore, steps 801 to 807 may be omitted in order to reduce the complexity of the system. In this case, the signal input to the virtual channel unit 640 is a signal output after the received signal Y is equalized by the equalizer 510.

Will be

7 is a diagram illustrating an equalization method in the receiver according to the first embodiment of the present invention.

) 출력한다.If the receiver receives the signal (Y) transmitted from the transmitter in step 701, the equalizer 510 equalizes the received signal to estimate the transmission signal in step 703 (

)

In step 705, the interference cancellation signal generator 520 generates received signals Y ′ and Y ″ from which interference between transmission signals is removed using the estimated transmission signal.

',

''을 생성하고, 이를 선형 가산하여

'''을 출력하여 최종적으로 추정된 송신 신호를 생성한다.In step 707, the equalizer 510 independently equalizes Y 'and Y''.

',

'' And add it linearly

'''Is output to generate the finally estimated transmission signal.

8 is a view for explaining operation 705 of FIG.

)로부터 심볼을 생성한다. 상기 심볼 생성부(610)는 심볼을 생성하기 위하여

에 상기 도 1에서 설명한 송신기의 신호 처리 절차의 역 절차를 적용한다. 즉,

에 역 프리코딩 행렬을 곱한 이후, 역 스크램블링을 수행하고, 역 확산을 수행하여 심볼을 생성한다. 상기 역 확산은 역 하다마드(Hadamard) 변환을 통하여 수행될 수 있고, 상기 역 스크램블링은 PN 코드를 곱하여 수행되는 것이 일반적이다.In operation 801, the symbol generator 610 may estimate the transmitted signal (sent from the interference cancellation signal generator 520).

Create a symbol from. The symbol generator 610 generates a symbol

The reverse procedure of the signal processing procedure of the transmitter described above with reference to FIG. In other words,

After multiplying the inverse precoding matrix by, inverse scrambling is performed and inverse spreading is performed to generate a symbol. The inverse spreading may be performed through an inverse Hadamard transformation, and the inverse scrambling is generally performed by multiplying the PN code.

In operation 803, the symbol cleaner 620 may perform symbol cleaning on the generated symbols to generate more reliable symbols than the symbols generated by the symbol generator 620. The present invention proposes a method for efficiently performing symbol cleaning, which will be described later with reference to FIG. 9.

In operation 805, the virtual transmitter 630 applies the signal processing procedure of the transmitter of FIG. 1 to the symbol on which the cleaning is performed. That is, after spreading and scrambling are performed on the symbols, the precoding matrix is multiplied. The spreading may be performed through a Hadamard transform, and the scrambling is generally performed by multiplying a PN code.

In operation 807, the virtual channel unit 640 outputs the output signal of the virtual transmitter 630 to the output signal of the virtual transmitter 630 in order to experience the channel H between the transmitter 200 and the receiver 210. Apply the channel by multiplying the channel function.

In operation 809, the signals Y 'and Y ″ are generated from the received signal Y by removing the interference between the transmission signals using the symbols to which the channel is applied.

)의 신뢰성을 더욱 향상시키기 위한 구성들이다. 따라서 시스템의 복잡성을 감소시키고자 하는 경우에 상기 801 내지 807 단계는 생략될 수 도 있다. 이 경우 807단계에서 가상 채널부(640)는 수신 신호(Y)가 등화부(510)에서 등화되어 추정된 송신 신호(

)에 채널 함수를 곱할 것이다. 만일 신호가 시간 영역에서 처리되는 경우라면 채널 함수의 컨벌루션이 수행될 것이다.Meanwhile, in steps 801 to 805, the transmission signal estimated by being equalized by the equalizer 510 (

) To further improve the reliability. Therefore, steps 801 to 807 may be omitted in order to reduce the complexity of the system. In this case, in operation 807, the virtual channel unit 640 receives the estimated transmission signal (E) after the received signal Y is equalized by the equalizer 510.

) Will be multiplied by the channel function. If the signal is processed in the time domain, convolution of the channel function will be performed.

Hereinafter, a symbol cleaning scheme proposed in the first embodiment of the present invention mentioned in FIG. 6 will be described.

Symbol cleaning is commonly referred to as soft slicing, and may be used interchangeably in the following. The soft slicing scheme is generally used as a linear MMSE scheme and a general, that is, a non-linear MMSE scheme.

The result of soft slicing using the linear MMSE method is known to be represented by Equation (13), and the result of soft slicing using the nonlinear MMSE method is known to be represented by Equation (14).

Meanwhile, if Equation 14 is rearranged, Equation 15 may be expressed.

 Referring to Equation 15, it is necessary to implement an exponential function in order to implement soft slicing of the nonlinear MMSE method. However, there is a problem that it is very complicated to implement the exponential function in hardware. Therefore, the present invention proposes a method of approximating an exponential function to a piecewise linear function that is easy to implement in hardware.

The reason for performing symbol cleaning, that is, soft slicing, is to increase the reliability of the symbol generated by the symbol generator 610. On the other hand, the non-linear MMSE method may be expressed as a linear combination of all constellation points on the constellation. In this case, contributions to the result value of Equation 15 are different from each other according to positions on the constellations of the received symbols.

For example, in the constellation of the 16-QAM modulation scheme as shown in FIG. 9, the contribution of each constellation point to the nonlinear MMSE estimate is higher as it is closer to the received signal y 900 in the time domain, and as it is farther away. In FIG. 9, the contribution of the constellation 901 is highest, the constellation 903 is next highest, and the contribution of the farthest constellations 905 is low.

On the other hand, the easiest way to think about when approximating an exponential function is to first approximate the exponential function using a Taylor series, as shown in Fig. 10, or to consider the constellation point that is farthest from the received signal. There is a way not. Reference numerals 1010, 1020, and 1030 of FIG. 10 denote actual exponential functions, approximation results using a first order Taylor series, and approximation results using a second order Taylor series.

As can be seen in FIG. 10, the simulation results show that the schemes do not work as desired or exhibit significantly worse results when using the actual nonlinear MMSE function.

Therefore, in order to approximate the exponential function more precisely, the present invention proposes a method of dividing the exponential function according to its importance and expressing the exponential function as an optimal linear function for each interval.

In the present invention, in order to minimize the error when approximating the exponential function to a linear function, the size of the interval of the exponential function is set differently according to the magnitude of the function value of the exponential function. This will be explained with reference to FIG.

In addition, when the linear function is determined from the exponential function of each of the differently set intervals, the linear function determined by the start and end points of the actual exponential function in the corresponding interval is appropriately parallelized to determine the finally approximated linear function. The parallel movement is set so as to satisfy a condition that the mean square error (MSE) between the exponential function and the finally approximated linear function is minimized. That is, the parallel movement is determined to satisfy the MMSE. This will be explained with reference to FIG. 12.

FIG. 11 is a diagram illustrating a method of dividing an exponential function by sections in order to approximate an exponential function as a linear function according to an embodiment of the present invention.

Referring to FIG. 11, it can be seen that an exponential function is divided into four sections, that is, D1 1101, D2 1103, D3 1105, and D4 1107.

In the first section D1 (1101), because the size of the function value is large, a more accurate approximation can be obtained by setting the interval interval narrower, and in the second section D2 (1103), the section value is smaller than the D1 (1101) section. The interval of is made larger than D1 (1101). Therefore, when the value of the exponential function is large, a more accurate function of the result of soft slicing according to Equation 15 can be obtained, and as a result, the nonlinear MMSE value is more accurate.

FIG. 12 is a diagram illustrating a process of determining an approximated linear function of an exponential function in a corresponding section according to the first embodiment of the present invention.

Reference numeral 1201 denotes a point mapping for determining a linear function by using a start point and an end point of an actual exponential function in a corresponding section. When the linear function is determined according to the method of point mapping 1201, it can be seen that the error becomes very large in the remaining portions except for the start point and the end point of the linear function. This results in a large error between the linear and exponential functions. Therefore, in order to minimize this error, the present invention allows the linear function determined by the point mapping to be a linear function satisfying the MMSE condition. Reference numeral 1203 denotes the determination of the linear function such that the MMSE condition is satisfied according to an embodiment of the present invention. Hereinafter, a process of determining a linear function to satisfy the MMSE condition will be described.

When the coordinates of the start point and the end point of the exponential function are [x ₀ , y ₀ ], [x ₁ , y ₁ ] in an arbitrary interval, [x ₀ , y ₀ ], [x ₁ , y ₁ ] The linear function as the start point and the end point is determined by Equation 16 below.

In order to minimize the error between the linear function and the exponential function in the linear function of Equation 16, the slope value α and the y intercept value β may be determined to minimize the error between the linear function and the exponential function. In the present invention, the gradient value α and the y intercept value β are determined to minimize Mean Square Error (MSE).

MSBs according to the α and β values are represented by the following Equations 17 and 18.

According to Equation 18, Equation 17 is rearranged as in Equation 19.

In Equation 19, MSE (α, β) of Equation 17 is expressed as J (α, β) for convenience. On the other hand, in order to find α and β in which J (α, β) of Equation 19 is minimized, that is, MSE (α, β) is minimized, the values of partial derivatives of Equation 19 with respect to α and β are It is necessary to find α and β values that become zero. This process is represented by Equation 20 below.

To rearrange Equation 20, Equation 21 is represented.

According to Equation 21, J and α, ie, α and β may be determined to minimize MSE (α, β). Therefore, the linear function determined according to the determined α and β becomes the final approximated linear function of the exponential function of the interval.

An example of approximating an exponential function according to the above-described exponential function approximation method of the present invention will be described. If the interval of the exponential function is divided into six stages, the linear function approximated in each interval is expressed by Equation 22 below.

That is, referring to Equation 22, since the value of the exponential function decreases as x increases, the length of the corresponding section becomes longer as x increases. The α and β values for determining the linear function for each exponential function in the divided sections are determined by Equation 21.

Hereinafter, the result of simulating the equalization performance of the present invention as described above.

Simulation was performed in the HSPA 2x2 MIMO system environment, and simulation was performed for QPSK, 16-QAM, and 64-QAM, and soft slicing is used for linear slicing, in order to identify equalization performance according to the symbol cleaning, that is, symbol slicing. Nonlinear Slicing An approximate nonlinear (ALN) slicing scheme was used. The pseudo nonlinear slicing method divides the exponential function into 16 sections.

Simulation results show that using the equalizer proposed in the present invention, a reception gain of about 2 to 3 dB can be obtained compared to the conventional LMMSE equalizer.

Specifically, in the case of using the QPSK modulation method, the received signal from which the interference signal has been removed according to the present invention is equalized, and when the linear function approximation method is used in soft slicing, the performance improvement of about 2.5 dB was confirmed. An additional 2dB gain was found when using the method or similar nonlinear slicing method.

On the other hand, when 16-QAM modulation is used, the received signal from which the interference signal is removed according to the present invention is equalized, and when the linear function approximation method is used for soft slicing, the performance improvement of about 2 dB was confirmed. The gain of 3dB was confirmed when using the slicing method or the pseudo nonlinear slicing method.

In addition, when the 64-QAM modulation is used, the received signal from which the interference signal is removed is equalized according to the present invention, and when the linear function approximation method is used during soft slicing, the performance improvement of about 2 dB was confirmed. When using the slicing method or the pseudo nonlinear slicing method, an additional 2dB gain was confirmed.

Second Embodiment

In the second embodiment of the present invention described below, the equalization scheme using the interference cancellation proposed in the first embodiment is applied to the HS channel (HS-PDSCH) signal and the non-HS channel separately to further improve performance. It is an improvement.

The "HS channel" refers to an HSDPA data channel (HS-PDSCH) defined in 3GPP Release 7 or later. The HS data channel uses 16 fixed as a spreading factor (SF). Meanwhile, QPSK, 16QAM, 64QAM, etc. may be used as the modulation scheme. In the 2x2 multi-antenna system, a precoding scheme is used for the HS channel.

The non-HS channel is a pilot channel (CPICH), a control channel defined in 3GPP Release 99 (R 99) such as P-CCPCH, S-CCPCH, HS control channel (HS-SCCH) and R99 data channel (DPCH) And the like. In the non-HS channel, SF is not fixed and various SFs from 4 to 256 can be used. For example, SF 256 is used for the pilot channel, and SF 128 is typically used for the R99 control channel, the R99 data channel, and the HS control channel. On the other hand, the modulation scheme is fixed to QPSK. In addition, precoding is not applied to the non-HS channel like the HS channel in a 2x2 multi-antenna system. Instead, space-time transmit diversity (STTD) is applied. To this end, the STTD encoding is performed at the transmitting side, and the STTD decoding and STTD re-encoding process is required at the receiving side in order to recover the signal.

FIG. 13 is a diagram for explaining a tree structure of an OVSF code that is a channelization code of WCDMA downlink.

In the case of SF 16 in FIG. 13, 16 code channels of the SF 16 branch of the OVSF code are selected from Ch0,... , Ch15, is assigned to CPICH, PICH, AICH, PCCPCH, HS-SCCH 1303, and the like. Thus, Ch0 1301 cannot be assigned to the HS data channel.

On the other hand, the remaining 15 channels 1305 may be assigned to the HS channel or to the non-HS channel. For example, the current terminal is a terminal that receives an HS-DSCH channel signal of SF 16, and the terminal is allocated Ch1 to CH5 among 16 channels of SF, and the remaining CH6 to CH15 are R99 data channels of SF 128. This is the case when it is allocated to other terminals receiving a signal. For reference, if an OVSF code is assigned and used by one user in a higher layer (for example, 0000 in SF4), all codes in the lower layer of the branched tree in that code (for example, 00000000 in SF 8). , 0000 .... 0000 of SF 64) can no longer be used. This is because orthogonality is not maintained between the code of the characteristic upper layer of the OVSF code tree structure and the code of the lower layer branched from the code. Similarly, if the code in the lower layer is used first, the code in the upper layer of the code cannot all be used.

When the HS channel signal is received using the SF 16 as described above, not all code channels of the SF 16 are allocated to the HS channel using the SF 16. However, the first embodiment may exhibit optimal performance when all code channels are configured as HS channels. Therefore, in the second embodiment of the present invention, when the code channels include both the HS channel signal and the non-HS channel signal, the function block configured to determine which channel signals are the corresponding code channels, and, if the HS channel is an HS channel, to process the HS signal. Eliminate interference, and if it is a non-HS channel, remove the interference from a functional block configured to handle the non-HS channel.

Hereinafter, the second embodiment will be described in detail.

14 is a diagram illustrating a configuration of an LMMSE equalizer in a receiver of a 2X2 MIMO transceiving system according to a second embodiment of the present invention.

14 is a modification of the configuration of the equalizer according to the first embodiment described in FIG.

Comparing the equalizer of FIG. 14 with the equalizer of FIG. 5, since the first embodiment is assumed to process the HS channel, the interference cancellation signal generator 520 processes all the HS channel signals. However, in the second embodiment, the HS channel signal and the non-HS channel signal are each processed in separate blocks. That is, the HS channel signal processor 1421 processes the HS channel signal and the non-HS channel signal processor 1425 processes the non-HS channel signal.

Meanwhile, the channel determiner 1423 uses the symbol received from the HS channel signal processor 1421 and the symbol received from the non-HS channel signal processor 1425, according to the method proposed by the present invention, and the corresponding signal is an HS channel signal. Determine whether it is a non-HS channel signal.

The specific operation is as follows.

)는 HS 채널 신호 처리부(1421)과 비HS 채널 신호 처리부(1425)로 각각 전달된다. The received signal Y is input to the equalizer 1410 and is equalized in one of the first LMMSE 1411 or the second LMMSE 1413 included in the equalizer 1410 to estimate the transmitted signal. Estimated transmission signal (

) Are transmitted to the HS channel signal processor 1421 and the non-HS channel signal processor 1425, respectively.

)가 SF 16을 이용하여 송신된 것으로 가정하고, SF 16을 이용하여 심볼을 생성한다. 한편, 비 HS 채널 신호 처리부(1425)는 상기 추정된 송신 신호(

)가 SF 128을 이용하여 송신된 것으로 가정하고, SF 128을 이용하여 심볼을 생성한다. 각각 생성된 심볼은 채널 결정부(1423)으로 전달된다. The HS channel signal processor 1421 may transmit the estimated transmission signal (

Is assumed to be transmitted using SF 16 and generates a symbol using SF 16. Meanwhile, the non-HS channel signal processor 1425 may transmit the estimated transmission signal (

Is assumed to be transmitted using SF 128, and a symbol is generated using SF 128. Each generated symbol is transferred to the channel determiner 1423.

The channel determiner 1423 determines whether each code of SF 16 is allocated to HS channel data using SF 16 or non-HS channel data using SF 128 according to a method described below. If the channel is determined, the channel information for each code determined according to the determined channel is transmitted to the HS channel signal processor 1421 or the non-HS channel signal processor 1425. That is, if Ch1 is an HS channel and Ch2 is a non-HS channel, this information is transmitted to the HS channel signal processor 1421 and the non-HS signal processor 1425, respectively.

The HS channel signal processing unit 1421 and the non-HS channel signal processing unit 1425 generate a signal from which interference is removed with respect to code data to be processed according to the transmitted channel information for each code.

',

''이 출력된다. 상기

'과

''는 가산기(515)에서 선형 가산 되어

'''이 출력된다. 비HS 채널 신호 처리부(1425)에서도 동일한 방식으로 (Y_{N - HS}', Y_{N - HS}'')을 생성한다. 이렇게 각 코드 별로 간섭이 제거된 추정된 송신 신호(

''')가 출력되고 이를 모두 합하여 송신측, 즉, 기지국이 송신한 신호를 최종적으로 추정하게 된다.That is, the HS channel signal processor 1421 generates a signal (Y _{HS '} , Y _HS'' ) for each code, and the generated (Y _HS' , Y _{HS ''} ) is the first LMMSE of the equalizer 1410. 1411 and the second LMMSE (1413) and then each independently equalized

',

Is printed. remind

'and

'' Is linearly added in adder 515

'''Is printed. The non-HS channel signal processor 1425 generates (Y _{N - HS} ′, Y _{N - HS} '') in the same manner. In this way, the estimated transmission signal whose interference is removed for each code

''') Is output and the sum is added to finally estimate the signal transmitted from the transmitting side, that is, the base station.

FIG. 15 is a diagram illustrating a configuration of the HS channel signal processing unit 1421 of FIG. 14. In FIG. 14, the configuration of the HS channel signal processor 1421 is substantially the same as that of FIG. 6.

That is, the HS channel signal processor 1421 includes a symbol generator 1510, a symbol cleaner 1520, a virtual transmitter 1530, a virtual channel unit 1540, and an interference cancellation processor 1550.

로부터 심볼을 생성한다. 상기 심볼 생성부(1520)는 심볼을 생성하기 위하여

에 상기 HS 송신기의 신호 처리 절차의 역 절차를 적용한다. 즉,

에 역 프리코딩 행렬을 곱한 이후, 역 스크램블링을 수행하고, 역 확산을 수행하여 심볼을 생성한다. 상기 역 확산은 역 하다마드(Hadamard) 변환을 통하여 수행될 수 있고, 상기 역 스크램블링은 PN 코드를 곱하여 수행되는 것이 일반적이다.The symbol generator 1510 estimates the transmitted signal transmitted from the equalizer 1410.

Create a symbol from. The symbol generator 1520 generates a symbol

The reverse procedure of the signal processing procedure of the HS transmitter is applied. In other words,

After multiplying the inverse precoding matrix by, inverse scrambling is performed and inverse spreading is performed to generate a symbol. The inverse spreading may be performed through an inverse Hadamard transformation, and the inverse scrambling is generally performed by multiplying the PN code.

The symbol generated by the symbol generator 1510 is transferred to the symbol cleaner 1520, the channel determiner 1423, and the signal of the corresponding code channel is HS channel signal according to the method described later by the channel determiner 1423. If determined to be, the generated symbol is transferred back to the symbol cleaner 1520.

The symbol cleaner 1520 may generate a symbol that is more reliable than a symbol generated by the symbol generator 1510 by performing symbol cleaning on the generated symbol. The symbol cleaning scheme is the same as described above with reference to FIG. 9.

The virtual transmitter 1530 applies the signal processing procedure of the transmitter of FIG. 1 to the symbol on which the cleaning is performed. That is, after spreading and scrambling are performed on the symbols, the precoding matrix is multiplied. The spreading may be performed through a Hadamard transform, and the scrambling is generally performed by multiplying a PN code.

On the other hand, the virtual channel unit 1540 is a channel to the output signal of the virtual transmitter 630 in order for the output signal of the virtual transmitter 1530 to experience the channel (H) between the transmitter 200 and the receiver 210. Multiply the function. If the signal is processed in the time domain, convolution of the channel function will be performed.

'''을 출력한다.The interference cancellation processing unit 1550 outputs Y _HS ′ and Y _HS ″ using the output signal of the virtual channel unit 640 from the received signal Y, thereby providing a reliable reception signal from which interference between transmission signals is removed. Create them. After the Y _HS 'and Y _HS ''is input to the equalizing unit 1410 of FIG.

Print '''.

FIG. 16 is a diagram illustrating a configuration of the non-HS channel signal processing unit 1425 of FIG. 14.

The non-HS channel signal processor 1621 includes a symbol generator 1610, a symbol cleaner 1620, a virtual transmitter 1630, a virtual channel unit 1640, and an interference cancellation processor 1650. Since the above configurations perform almost the same operations as the respective corresponding configurations described with reference to FIG. 15, a description of the same operations will be omitted.

However, the symbol generator 1510 of FIG. 15 multiplies the inverse precoding matrix and performs descrambling and despreading, whereas the symbol generator 1610 of FIG. 16 performs STTD decoding and performs descrambling and despreading. There is a difference in performance. As described above, since the transmitter performs STTD encoding on the non-HS channel (R99 data channel) data in the multi-antenna system, the receiver performs STTD decoding instead of inverse precoding. In the STTD decoding, since the non-HS channel uses the SF 128 under the HSPA-MIMO system, non-HS channel signals can be detected using the SF 128. However, as an exception, since the pilot channel is STTD coded based on SF 256, SF 256 should also be used for STTD decoding.

In addition, although the virtual transmitter 1530 of FIG. 15 performs spreading and scrambling on the symbol and multiplies the precoding matrix, the virtual transmitter 1630 of FIG. 16 processes the non-HS channel signal, so that the STTD after spreading and scrambling is performed. Perform the encoding.

Hereinafter, a method of determining a channel by the channel determiner 1423 will be described.

In the second embodiment of the present invention, hypothesis testing is performed. The first hypothesis is that the transmitter transmits HS channel (HS-PDSCH) data using the corresponding code. The second hypothesis is that the transmitter transmits non-HS channel data (eg DPCH of SF128) using the corresponding code. Assume the case.

Hypothesis 1

Assume that the transmitter has transmitted HS channel data using an OVSF code of SF16. The received symbol of the HS channel data of SF 16 is a form in which the gain g of the code channel and the symbol s are multiplied and the noise v is added, as shown in Equation 23 below.

First, when the receiver receives a signal using the OVSF code of SF 16, the power of the recovered symbol is calculated as follows.

The value z multiplied by the corresponding OVSF code of SF16 to the signal of Equation 23 is expressed by Equation 24 below.

In Equation 24, N is a value of SF and N = 16.

Meanwhile, the square mean of the signal z in Equation 24 is expressed by Equation 25 below.

Next, with respect to <Hypothesis 1>, when the receiver receives a signal using the OVSF code of SF 128, the power of the recovered symbol is calculated as follows.

A total of eight codes of SF 128 branched from one code of SF 16. Therefore, consider the entire symbol, which is the sum of all eight symbols. On the other hand, the signal z of the following equation (8) is multiplied by each of the eight corresponding OVSF codes of SF 128 and summed is represented by the following equation (26).

Meanwhile, the square mean of the signal z in Equation 26 is expressed by Equation 27 below.

In Equation 27, N1 = 128/16 = 8 and N2 = 128.

On the other hand, according to the Law of large numbers, the sum of eight different (average zero) symbols Si can be approximated to a single Gaussian random variable with an average of zero, which is Equation 28>

Equation (28) has a distribution of Gaussian random variables as shown in Equation (29).

In the two cases of <Hypothesis 1>, i.e., the transmitting side transmits a signal using SF 128 and the receiving side uses SF 16 and each case using SF 128, Comparing <Equation 25> and <Equation 27>, it can be seen that the squared mean value (ie, power value) of the restored signal z in the case of SF 128 is smaller than the value in the case of SF 16. In addition, it can be seen that the ratio of the square mean value at SF 128 and the square mean value at SF 16 is approximately the inverse of the ratio of SF (16/128 = 1/8).

Hypothesis 2

Assume that the transmitter has transmitted non-HS channel (DPCH) data using an OVSF code of SF 128. The received symbol of the HS channel data of SF 128 is a form in which the gain g of the code channel and the symbol s are multiplied and the noise v is added, as shown in Equation 30 below.

First, with respect to Hypothesis 2, when the receiver receives a signal using an OVSF code of SF 128, the power of the recovered symbol is calculated as follows.

A value z multiplied by the corresponding OVSF code of SF16 to the signal of Equation 30 is expressed by Equation 31 below.

Where N = 128.

Meanwhile, the square mean of the signal z in Equation 31 is expressed by Equation 32 below.

Next, for the hypothesis 2, when the receiver receives a signal using the OVSF code of SF 16, the power of the recovered symbol is calculated as follows.

The value z multiplied by the corresponding OVSF code of SF 16 for the symbol is expressed by Equation 33 below.

In Equation 33, N ₁ is a value of SF and N = 16.

Meanwhile, in Equation 33, the square mean of the signals z is expressed by Equation 34 below.

In the two cases for Hypothesis 2, that is, the transmitting side transmits a signal using SF 128, and the receiving side uses SF 128 and each case using SF 16, Comparing Eq 32 and Eq 34, it can be seen that the noise power of the restored signal z in the case of SF 128 is smaller than the noise power in SF 16.

The results of Hypothesis 1 and Hypothesis 2 can be used as follows.

First, when the transmitter actually transmits the corresponding channel using SF 16, the ratio r of the power of the signal obtained by the receiver using SF 16 and the power of the signal obtained by using SF 128 is expressed as Eq 35>

In Equation 35, r is greater than 8. In addition, when the power of the signal is greater than the power of the noise, the r value can be approximated to 8.

Second, when the transmitter actually transmits the corresponding channel using SF 128, the ratio r of the power of the signal obtained by the receiver using SF 16 and the power of the signal obtained by using SF 128 is given by Eq 36>

In Equation 36, r is slightly larger than 1. If the power of the signal is greater than the power of noise, the r value can be approximated to one.

The r value is slightly larger than 1, and the r value can be approximated to 1 when the signal square is greater than the noise power.

As described above, in the second embodiment of the present invention, a method of determining whether the corresponding channel is a channel using SF 16 or a SF 128 is as follows.

That is, the sum of the power values of the signals received using the corresponding SF 16 codes in the branches of SF 16 and the power of the eight signals received using the 8 codes of SF 128 branched from the corresponding SF 16 branches. If the comparison result exceeds a predetermined reference value, the corresponding channel is a channel (that is, an HS channel) signal transmitted using SF 16, and thus, the corresponding channel is determined as an HS channel. To determine the transmitted channel (ie, R99 channel).

The hypothesis test described above is expressed by Equation 37 below.

In Equation 37, the τ value is a value that can be appropriately selected between 1 and 8 according to the system setting. Based on the result of Equation 37, it is determined whether the corresponding channel is the HS channel or the non-HS channel for the 16 code channels except Ch0 among the 16 channels of the SF 16.

If it is determined whether each code channel is an HS channel or a non-HS channel for each, the operation after symbol generation is performed in a block corresponding to the determined channel.

For example, if the Ch1 code channel is determined to be an HS channel, and the Ch2 code channel is determined to be a non-HS channel, a series of blocks after symbol generation in the HS signal processor 1421 for the signal of the Ch1 code channel (ie, By performing the operations of the symbol cleaning 1520, the virtual transmitter 1530, the virtual channel unit 1540, and the interference cancellation processor 1550, the signals whose interference is removed from the code channel signal (Y _HS ′, Y _HS ''). ) Is estimated and output. On the other hand, the non-HS signal processor 1425 sets the output to 0 for the Ch1 code channel. On the other hand, with respect to the signal of the Ch2 code channel, the non-HS signal processor 1425 has a series of blocks after symbol generation (ie, symbol cleaning 1620, virtual transmitter 1630, virtual channel unit 1640, and interference cancellation processor). In operation 1650, the interference _{- reduced} signals Y _{N - HS} ′ and Y _N-HS ″ may be estimated and output for the code channel signal. On the other hand, the HS signal processor 1421 sets the output to 0 for the Ch2 code channel.

In this way, a channel of each code channel is determined, and a signal from which interference is removed is generated by removing interference of a corresponding code channel signal in a signal processing block corresponding to the determined channel. Will be estimated.

17 is a view for explaining an equalization method according to a second embodiment of the present invention.

In operation 1701, the transmission signal is estimated by equalizing the reception signal. In operation 1703, the channel of the transmission signal is determined using the estimated reception signal. In operation 1705, signals generated by removing interference between transmission signals are generated using a separate signal processor according to the determined characteristics of the channel. In operation 1707, equalized and equalized signals obtained by removing the interference are independently added. To generate a final estimated transmission signal.

18 is a diagram showing simulation results according to the second embodiment of the present invention. In addition, the simulation results are shown in Table 1 below.

As described above, the first embodiment is a method including only HS-channel signal processing, and the second embodiment is a method including both HS channel signals and non-HS channel signals, and Table 1 is the second embodiment. The experimental results are compared. As can be seen in FIG. 18, it can be seen that the result of the second iteration of the second embodiment is improved by 2 dB or more than the result of the second iteration of the first embodiment.

On the other hand, in the second embodiment, since the HS channel data and the non-HS channel data, that is, the R99 channel and the control channel data can be restored at the same time, the inter-stream interference can be effectively eliminated when used, thereby reducing reception performance. It can greatly improve.

Claims (12)

In a multiple input multiple output (MIMO) wireless communication system comprising at least two transmit antennas and at least two receive antennas, the first channel data and the second spread coefficients are spread using the first spreading factor (SF). A method of equalizing a received signal comprising at least one of spread second channel data,
Estimating the transmission signal by equalizing the reception signal;
A channel determining process of determining whether the estimated transmission signal is the first channel data or the second channel data;
If the estimated transmission signal is the first channel data, removing the interference according to the characteristics of the first channel; And
If the estimated transmission signal is the second channel data, removing the interference according to the characteristics of the second channel.
The method of claim 1, wherein the channel determination process,
Comparing a ratio of a power value of a symbol generated by applying the first SF to the estimated transmission signal and a power value of a symbol generated by applying the second SF to the estimated transmission signal with a predetermined reference value; ,
The estimated transmission signal is determined as the first channel data when the ratio exceeds the reference value as a result of the comparison, and the estimated transmission signal is determined as the second channel data when the ratio does not exceed the reference value. A method of equalizing a received signal comprising a process.
The method of claim 1,
And wherein the first SF is 16 and the first channel is an HS channel (HS-PDSCH).
The method of claim 3, wherein the removing of the interference according to the characteristics of the first channel comprises:
Generating a symbol by applying an inverse procedure of transmitter signal processing of the first channel data;
Cleaning the generated symbol;
Applying a virtual transmission procedure to the cleaned symbols;
Applying a channel between the transmitter and the receiver to a symbol to which the virtual transmission procedure is applied;
And generating a plurality of received signals from which the interference has been removed from the received signal by using the symbol to which the channel is applied.
The method of claim 1,
And wherein the second SF is 128 and the first channel is an R99 data (DTCH) channel.
The method of claim 5, wherein the removing of the interference according to the characteristics of the second channel comprises:
Generating a symbol by applying an inverse procedure of the transmitter signal processing of the second channel data;
Cleaning the generated symbol;
Applying a virtual transmission procedure to the cleaned symbols;
Applying a channel between the transmitter and the receiver to a symbol to which the virtual transmission procedure is applied;
And generating a plurality of received signals from which the interference has been removed from the received signal by using the symbol to which the channel is applied.
In a multiple input multiple output (MIMO) wireless communication system comprising at least two transmit antennas and at least two receive antennas, the first channel data and the second spread coefficients are spread using the first spreading factor (SF). An apparatus for equalizing a received signal comprising at least one of spread second channel data, the apparatus comprising:
An equalizer which equalizes the received signal and estimates the transmitted signal;
A channel determination unit determining whether the estimated transmission signal is the first channel data or the second channel data;
A first channel signal processor to remove interference according to the characteristics of the first channel if the estimated transmission signal is the first channel data; And
And a second channel signal processing unit for removing interference according to the characteristics of the second channel if the estimated transmission signal is the second channel data.
The method of claim 7, wherein the channel determination unit,
A power value of a symbol generated by the first channel signal processor applying the first SF to the estimated transmission signal, and a symbol generated by the second signal processor applying the second SF to the estimated transmission signal Comparing the ratio of the power value to a predetermined reference value, and determining the estimated transmission signal as the first channel data when the ratio exceeds the reference value, and if the ratio does not exceed the reference value, the estimation And determine the transmitted signal as the second channel data.
The method of claim 7, wherein
And wherein the first SF is 16 and the first channel is an HS channel (HS-PDSCH).
10. The method of claim 9,
The first channel signal processor,
A symbol generation unit generating a symbol by applying an inverse procedure of the transmitter signal processing of the first channel data;
A symbol cleaning unit cleaning the generated symbols;
A virtual transmitter for applying a virtual transmission procedure to the cleaned symbol;
A virtual channel unit applying a channel between the transmitter and the receiver to a symbol to which the virtual transmission procedure is applied; And
And an interference cancellation processor to generate a plurality of received signals from which interference is removed from the received signal by using the symbol to which the channel is applied.
The method of claim 7, wherein
Wherein the second SF is 128 and the first channel is an R99 data (DTCH) channel.
The method of claim 11, wherein the second channel signal processing unit,
A symbol generator for generating a symbol by applying a reverse procedure of transmitter signal processing of the second channel data;
A symbol cleaning unit cleaning the generated symbols;
A virtual transmitter for applying a virtual transmission procedure to the cleaned symbol;
A virtual channel unit applying a channel between the transmitter and the receiver to a symbol to which the virtual transmission procedure is applied; And
And an interference cancellation processor to generate a plurality of received signals from which interference is removed from the received signal by using the symbol to which the channel is applied.

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