KR20120100739A - Method of demapping in a wireless communication system and apparatus thereof - Google Patents

Abstract

PURPOSE: A demapping method in a wireless communication system and an apparatus thereof are provided to efficiently demap a signal using a modular arithmetic in the wireless communication system. CONSTITUTION: An antenna receives information, data, a signal, or a message. Transmitters(100a,100b) transmit the information, the data, the signal, or the message by controlling the antenna. Receivers(300a,300b) receive the information, the data, the signal, or the message by controlling the antenna. Memories(200a,200b) temporarily or permanently store various information in a wireless telecommunication system. Processors(400a,400b) are provided to control each element. The transmitter, the receiver, the memory, and the processor are respectively formed as an independent element by each individual chip. [Reference numerals] (100a,100b) Transmitter; (200a,200b) Memory; (300a,300b) Receiver; (400a,400b) Processor; (AA) Terminal; (BB) Base station

Description

METHOD OF DEMAPPING IN A WIRELESS COMMUNICATION SYSTEM AND APPARATUS THEREOF

The present invention relates to a wireless communication system, and more particularly, to a method and apparatus for demapping in a wireless communication system using a modular operation.

Wireless communication systems are widely deployed to provide various kinds of communication services such as voice and data. In general, a wireless communication system is a multiple access system capable of supporting communication with multiple users by sharing available system resources (bandwidth, transmission power, etc.). Examples of multiple access systems include code division multiple access (CDMA) systems, frequency division multiple access (FDMA) systems, time division multiple access (TDMA) systems, orthogonal frequency division multiple access (OFDMA) systems, and single carrier frequency (SC-FDMA). division multiple access) system.

An object of the present invention is to provide a method and an apparatus for efficiently performing demapping in a wireless communication system using a modular operation. The technical problems to be solved by the present invention are not limited to the technical problems and other technical problems which are not mentioned can be understood by those skilled in the art from the following description.

In order to solve the above problems, in one aspect of the present invention, in a method of demapping a receiving end in a wireless communication system, whether a first modulo operation is performed on an input signal and the input signal from a transmitting end. Receiving first information about the first information, when the first information is information indicating that the first modulo operation is performed, obtaining a received signal by performing a second modulo operation on the input signal, and receiving the received signal Generating a maximum function value most likely to correspond to the candidate constellation point on the extended constellation and generating a log-likelihood ratio (LLR) using the generated maximum function value. Can be.

In addition, when the power of the interference included in the input signal is smaller than the loss power generated by performing the first modulo operation, the transmitter does not perform the first modulo operation. Can be.

The method may further include receiving information on the total number of streams used from the transmitting end and the order of the first stream received by the receiving end.

In addition, when the order of the first stream is the first or the last, it may be determined that the first modulo operation is not performed.

In addition, the extended constellation may include a basic constellation point of the basic constellation and at least one extended constellation point.

In addition, the extended constellation point may be a constellation point repeatedly arranged to repeat the basic constellation point.

The extended constellation point may be selected from among constellation points disposed close to the basic constellation points of the basic constellation after repeatedly arranging the modulo box including the basic constellation point to be adjacent to the modulo box. Can be.

In addition, the candidate constellation point may be a constellation point representing the same bit string on the extended constellation.

In addition, the maximum function value may use the following equation.

는 잡음 분산,

는 상기 수신 신호,

는 채널 응답 및

의 값은 상기 후보 성상점의 상기 확장 성상점의 상기 확장 성상도 상에서의 좌표이다.here,

Is noise variance,

Is the received signal,

Is the channel response and

Is a coordinate on the extended constellation of the extended constellation point of the candidate constellation point.

In addition, the LLR may use the following equation.

는 α의 실수 값에 맵핑되는 k-번째 비트가 가리키는 것이 0인 심벌이며,

는 α의 실수 값에 맵핑되는 k-번째 비트가 가리키는 것이 1인 심벌이다.Where α is any one of the coordinates of the constellation symbol on the constellation diagram,

Is a symbol whose zero is indicated by the k-th bit, mapped to the real value of α,

Is a symbol of which the k-th bit, mapped to the real value of α, points to 1.

The receiving terminal may further include generating an effective signal-to-interference-plus-noise ratio (SINR) and transmitting the effective SINR to the transmitting terminal, wherein the effective SINR is represented by the following equation. It is available.

은

번째

,

은

의 총 개수,

은 심볼 채널의 뮤추얼 정보(mutual information)를 이용하여 결정되는 함수이고,

은

의 인벌스(inverse) 함수이다.here,

silver

th

,

silver

Total number of,

Is a function determined using the mutual information of the symbol channel,

silver

Inverse function of.

In addition, the mutual information may use the following equation.

는 i-번째 코드 비트,

은 코드 비트의 총 개수,

는

의 LLR이다.here,

Is the i-th code bit,

Is the total number of code bits,

The

LLR.

On the other hand, in another aspect of the present invention for solving the above problems, in the receiving end de-mapping in the wireless communication system, whether the first modulo operation is performed on the input signal and the input signal from the transmitting end; And a receiving module for receiving first information on the first information and a processor for performing a second modulo operation on the input signal to obtain a received signal when the first information is information indicating that the first modulo operation has been performed. The processor generates a maximum function value most likely to correspond to the candidate constellation point on the extended constellation, and uses the generated maximum function value to form a log-likelihood ratio (LLR). Can be generated.

In addition, when the power of the interference included in the input signal is smaller than the loss power generated by performing the first modulo operation, the transmitter does not perform the first modulo operation. Can be.

The receiving module may further receive information on the total number of streams used from the transmitting end and the order of the first stream received by the receiving end, and the processor may first order the first stream. In the first or last case, it may be determined that the first modulo operation is not performed.

In addition, the extended constellation may include a basic constellation point of the basic constellation and at least one extended constellation point.

In addition, the processor may determine the maximum function value using the following equation.

는 잡음 분산,

는 상기 수신 신호,

는 채널 응답 및

의 값은 상기 후보 성상점의 상기 확장 성상점의 상기 확장 성상도 상에서의 좌표이다.here,

Is noise variance,

Is the received signal,

Is the channel response and

Is a coordinate on the extended constellation of the extended constellation point of the candidate constellation point.

In addition, the processor may determine the LLR using the following equation.

는 α의 실수 값에 맵핑되는 k-번째 비트가 가리키는 것이 0인 심벌이며,

는 α의 실수 값에 맵핑되는 k-번째 비트가 가리키는 것이 1인 심벌이다.Where α is any one of the coordinates of the constellation symbol on the constellation diagram,

Is a symbol whose zero is indicated by the k-th bit, mapped to the real value of α,

Is a symbol of which the k-th bit, mapped to the real value of α, points to 1.

The apparatus may further include a transmission module configured to transmit an effective signal to interference-plus-noise ratio (SINR) to the transmitter, and the processor may generate the effective SINR using the following equation.

은

번째

,

은

의 총 개수,

은 심볼 채널의 뮤추얼 정보(mutual information)를 이용하여 결정되는 함수이고,

은

의 인벌스(inverse) 함수이다.here,

silver

th

,

silver

Total number of,

Is a function determined using the mutual information of the symbol channel,

silver

Inverse function of.

In addition, the processor may determine the mutual information using the following equation.

는 i-번째 코드 비트,

은 코드 비트의 총 개수,

는

의 LLR이다.here,

Is the i-th code bit,

Is the total number of code bits,

The

LLR.

According to the present invention, it is possible to efficiently perform demapping of signals in a wireless communication system using modular arithmetic. Effects obtained in the present invention are not limited to the above-mentioned effects, and other effects not mentioned above may be clearly understood by those skilled in the art from the following description. will be.

1 shows a configuration of a terminal and a base station to which the present invention is applied.
2 illustrates a signal processing procedure for transmitting an uplink signal by a terminal.
3 illustrates a signal processing procedure for transmitting a downlink signal by a base station.
4 illustrates an SC-FDMA scheme and an OFDMA scheme to which the present invention is applied.
5 illustrates examples of mapping input symbols to subcarriers in the frequency domain while satisfying a single carrier characteristic.
FIG. 6 illustrates a signal processing procedure in which DFT process output samples are mapped to a single carrier in clustered SC-FDMA.
7 and 8 illustrate a signal processing procedure in which DFT process output samples are mapped to multi-carriers in a clustered SC-FDMA.
9 illustrates a signal processing procedure of a segmented SC-FDMA.
10 illustrates examples of a radio frame structure used in a wireless communication system.
11 shows an uplink subframe structure.
12 is a flowchart illustrating a demapping method according to an embodiment of the present invention.
13 is a diagram illustrating an example of generating an extended constellation according to an embodiment of the present invention.
FIG. 14 illustrates an example in which candidate constellation points are arranged in the extension generation diagram generated in FIG. 13.
15 is an example of an extended constellation diagram in QPSK.
FIG. 16 is a diagram illustrating a received signal in which a modulo operation is performed on a symbol mapped to QPSK when noise is a Gaussian distribution.
17 is a diagram illustrating an example of DPC (Dirty Paper Coding) using a modular operator combined with channel encoding.
FIG. 18 is a view showing an example in which a QPSK symbol is embedded in a constellation in accordance with the present invention.
19 is a diagram showing an example of a system model in the case of MISO-THP in accordance with the present invention.
FIG. 20 is a diagram showing an example of mutual information which is a maximum transmission value in each SNR in accordance with the present invention.
21 is a diagram illustrating an example of the performance of a system coded at each SNR in connection with the present invention.
22 illustrates an example of a coding and decoding process using code bits in accordance with the present invention.
FIG. 23 is a diagram illustrating an example of an LLR distribution in connection with the present invention. FIG.
FIG. 24 is a diagram showing an example of mutual information in each SNR according to the present invention.
FIG. 25 illustrates an example of mutual information in an LLR distribution and an SNR calculated using modular Gaussian noise in accordance with the present invention.
FIG. 26 is a diagram illustrating an example of mutual information when and when a modular operation is applied according to the present invention.
FIG. 27 is a diagram illustrating an example of mutual information for each SNR in a short term in connection with the present invention.
28 illustrates an example of an LLR distribution according to a simplified calculation in connection with the present invention.
29 illustrates an example of link performance in connection with the present invention.
30 illustrates an example of mutual information according to different modular boxes in accordance with the present invention.

Hereinafter, preferred embodiments according to the present invention will be described in detail with reference to the accompanying drawings. DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS The following detailed description, together with the accompanying drawings, is intended to illustrate exemplary embodiments of the invention and is not intended to represent the only embodiments in which the invention may be practiced. The following detailed description includes specific details in order to provide a thorough understanding of the present invention. However, those skilled in the art will appreciate that the present invention may be practiced without these specific details.

In addition, the techniques, devices, and systems described below may be applied to various wireless multiple access systems. Examples of multiple access systems include code division multiple access (CDMA) systems, frequency division multiple access (FDMA) systems, time division multiple access (TDMA) systems, orthogonal frequency division multiple access (OFDMA) systems, and single carrier frequency (SC-FDMA). division multiple access (MCD) systems and multi-carrier frequency division multiple access (MC-FDMA) systems. CDMA may be implemented in a radio technology such as Universal Terrestrial Radio Access (UTRA) or CDMA2000. TDMA may be implemented in wireless technologies such as Global System for Mobile Communication (GSM), General Packet Radio Service (GPRS), Enhanced Data Rates for GSM Evolution (EDGE), and the like. OFDMA may be implemented in wireless technologies such as Institute of Electrical and Electronics Engineers (IEEE) 802.11 (Wi-Fi), IEEE 802.16 (WiMAX), IEEE 802-20, evolved-UTRA (E-UTRA), and the like. UTRAN is part of Universal Mobile Telecommunication System (UMTS), and 3rd Generation Partnership Project (3GPP) Long Term Evolution (LTE) is part of E-UMTS using E-UTRAN. 3GPP LTE adopts OFDMA in downlink and SC-FDMA in uplink. LTE-advanced (LTE-A) is an evolution of 3GPP LTE. For convenience of explanation, hereinafter, it will be described on the assumption that the present invention is applied to 3GPP LTE / LTE-A. However, the technical features of the present invention are not limited thereto. For example, although the following description is described based on a wireless communication system in which the wireless communication system corresponds to a 3GPP LTE / LTE-A system, any other wireless communication except for those specific to 3GPP LTE / LTE-A Applicable to the system as well.

In some instances, well-known structures and devices may be omitted or shown in block diagram form centering on the core functions of the structures and devices in order to avoid obscuring the concepts of the present invention. In addition, the same components will be described with the same reference numerals throughout the present specification.

In the present invention, a terminal may be fixed or mobile, and collectively refers to devices that transmit and receive various data and control information by communicating with a base station. The terminal may be a user equipment (UE), a mobile station (MS), a mobile terminal (MT), a user terminal (UT), a subscriber station (SS), a wireless device, a personal digital assistant (PDA), or a wireless modem. modem, handheld device, and the like.

Also, a base station generally means a fixed station communicating with a terminal or another base station, and communicates with the terminal and other base stations to exchange various data and control information. The base station may be named in other terms such as an evolved-NodeB (eNB), a base transceiver system (BTS), an access point, and the like.

In the present invention, the specific signal is assigned to the frame / subframe / slot / carrier / subcarrier means that the specific signal is transmitted through the corresponding carrier / subcarrier in the period or timing of the frame / subframe / slot.

In the present invention, the rank or transmission rank refers to the number of layers multiplexed or allocated on one OFDM symbol or one resource element.

In the present invention, Physical Downlink Control CHannel (PDCCH) / Physical Control Format Indicator CHannel (PCFICH) / PHICH (Physical Hybrid automatic retransmit request Indicator CHannel) / PDSCH (Physical Downlink Shared CHannel) are respectively DCI (Downlink Control Information) / CFI ( Means a set of resource elements that carry ACK / NACK (ACKnowlegement / Negative ACK) / downlink data for uplink transmission.

In addition, PUCCH (Physical Uplink Control CHannel) / PUSCH (Physical Uplink Shared CHannel) / PRACH (Physical Random Access CHannel) means a set of resource elements that carry uplink control information (UCI) / uplink data / random access signals, respectively .

In particular, resource elements (REs) assigned to or belonging to PDCCH / PCFICH / PHICH / PDSCH / PUCCH / PUSCH / PRACH are assigned to PDCCH / PCFICH / PHICH / PDSCH / PUCCH / PUSCH / PRACH RE or PDCCH / PCFICH / PHICH, respectively. Named / PDSCH / PUCCH / PUSCH / PRACH resource.

Therefore, the expression that the terminal transmits the PUCCH / PUSCH / PRACH may be used in the same meaning as transmitting the uplink control information / uplink data / random access signal on the PUSCH / PUCCH / PRACH. In addition, the expression that the base station transmits the PDCCH / PCFICH / PHICH / PDSCH may be used in the same meaning as transmitting downlink control information / downlink data and the like on the PDCCH / PCFICH / PHICH / PDSCH.

On the other hand, mapping the ACK / NACK information to a specific constellation point is used in the same meaning as mapping the ACK / NACK information to a specific complex modulation symbol. In addition, mapping ACK / NACK information to a specific complex modulation symbol is used in the same sense as modulating ACK / NACK information to a specific complex modulation symbol.

1 shows a configuration of a terminal and a base station to which the present invention is applied. The terminal operates as a transmitter in uplink and as a receiver in downlink. In contrast, the base station operates as a receiver in uplink and as a transmitter in downlink.

Referring to FIG. 1, a terminal and a base station are antennas 500a and 500b capable of receiving information, data, signals or messages, and a transmitter 100a for controlling the antennas to transmit information, data, signals or messages, and the like. 100b), receivers 300a and 300b for controlling the antenna to receive information, data, signals or messages, and memories 200a and 200b for temporarily or permanently storing various information in the wireless communication system. In addition, the terminal and the base station are operatively connected to components such as a transmitter, a receiver, and a memory, and include processors 400a and 400b configured to control each component.

The transmitter 100a, the receiver 300a, the memory 200a, and the processor 400a in the terminal may be embodied as separate components by separate chips, respectively, and two or more may be included in one chip. It may be implemented by. In addition, the transmitter 100b, the receiver 300b, the memory 200b, and the processor 400b in the base station may be implemented as independent components by separate chips, respectively, and two or more chips may be used as one chip. It may also be implemented by). The transmitter and the receiver may be integrated to be implemented as one transceiver in the terminal or the base station.

The antennas 500a and 500b transmit a signal generated by the transmitters 100a and 100b to the outside or receive a signal from the outside and transmit the signal to the receivers 300a and 300b. Antennas 500a and 500b are also called antenna ports. The antenna port may correspond to one physical antenna or may be configured by a combination of a plurality of physical antennas. A transceiver supporting a multi-input multi-output (MIMO) function for transmitting and receiving data using a plurality of antennas may be connected to two or more antennas.

Processors 400a and 400b typically control the overall operation of various components or modules within a terminal or base station. In particular, the processor 400a or 400b includes various control functions for performing the present invention, a medium access control (MAC) frame variable control function according to service characteristics and a propagation environment, a power saving mode function for controlling idle mode operation, and a hand. Handover, authentication and encryption functions can be performed. The processors 400a and 400b may also be referred to as controllers, microcontrollers, microprocessors or microcomputers. Meanwhile, the processors 400a and 400b may be implemented by hardware or firmware, software, or a combination thereof.

When implementing the present invention using hardware, application specific integrated circuits (ASICs) or digital signal processors (DSPs), digital signal processing devices (DSPDs), programmable logic devices (PLDs), FPGAs ( field programmable gate arrays) may be provided in the processors 400a and 400b.

In addition, when the present invention is implemented using firmware or software, the firmware or software may be configured to include a module, a procedure, or a function that performs the functions or operations of the present invention, and is configured to perform the present invention. The firmware or software may be provided in the processors 400a and 400b or may be stored in the memory 200a and 200b to be driven by the processors 400a and 400b.

The transmitters 100a and 100b perform a predetermined coding and modulation on a signal or data to be transmitted to the outside, which is scheduled from the processor 400a or 400b or a scheduler connected to the processor, and then the antennas 500a and 500b. To pass). The transmitters 100a and 100b and the receivers 300a and 300b of the terminal and the base station may be configured differently according to a process of processing a transmission signal and a reception signal.

The memories 200a and 200b may store a program for processing and controlling the processors 400a and 400b and may temporarily store information input and output. In addition, the memory 200a or 200b may be utilized as a buffer. The memory may be a flash memory type, a hard disk type, a multimedia card micro type or a card type memory (e.g. SD or XD memory, etc.), RAM Access Memory (RAM), Static Random Access Memory (SRAM), Read-Only Memory (ROM), Electrically Erasable Programmable Read-Only Memory (EEPROM), Programmable Read-Only Memory (PROM), Magnetic Memory, Magnetic Disk, It can be implemented using an optical disk or the like.

2 illustrates a signal processing procedure for transmitting an uplink signal by a terminal. Referring to FIG. 2, the transmitter 100a in the terminal may include a scramble module 201, a modulation mapper 202, a precoder 203, a resource element mapper 204, and an SC-FDMA signal generator 205. have.

In order to transmit the uplink signal, the scramble module 201 may scramble the transmission signal using the scramble signal. The scrambled signal is input to the modulation mapper 202 to perform Binary Phase Shift Keying (BPSK), Quadrature Phase Shift Keying (QPSK), or 16 QAM / 64 Quadrature Amplitude Modulation (QAM) modulation methods, depending on the type of the transmission signal or the channel state. Modulated by a complex modulation symbol. The modulated complex modulation symbol is processed by the precoder 203 and then input to the resource element mapper 204, which may map the complex modulation symbol to a time-frequency resource element. The signal thus processed may be transmitted to the base station through the antenna port via the SC-FDMA signal generator 205.

3 illustrates a signal processing procedure for transmitting a downlink signal by a base station. Referring to FIG. 3, a transmitter 100b in a base station includes a scramble module 301, a modulation mapper 302, a layer mapper 303, a precoder 304, a resource element mapper 305, and an OFDMA signal generator 306. It may include.

In order to transmit a signal or one or more codewords in downlink, the signal or codeword may be modulated into a complex modulation symbol through the scramble module 301 and the modulation mapper 302 similar to FIG. 2. The complex modulation symbols are mapped to a plurality of layers by the layer mapper 303, and each layer may be multiplied by the precoding matrix by the precoder 304 and assigned to each transmit antenna. The transmission signal for each antenna processed as described above is mapped to a time-frequency resource element by the resource element mapper 305 and transmitted through each antenna port through an orthogonal frequency division multiple access (OFDMA) signal generator 306. have.

In a wireless communication system, when a terminal transmits a signal in uplink, a Peak-to-Average Ratio (PAPR) is a problem as compared with a case in which a base station transmits a signal in downlink. Accordingly, as described above with reference to FIGS. 2 and 3, the uplink signal transmission uses a single carrier-frequency division multiple access (SC-FDMA) scheme unlike the OFDMA scheme used for the downlink signaling.

4 illustrates an SC-FDMA scheme and an OFDMA scheme to which the present invention is applied. The 3GPP system employs OFDMA in downlink and SC-FDMA in uplink.

Referring to FIG. 4, both a terminal for uplink signal transmission and a base station for downlink signal transmission have a serial-to-parallel converter (401), a subcarrier mapper (403), and an M-point IDFT module (404). ) And the Cyclic Prefix (CP) addition module 406 are the same. However, the terminal for transmitting a signal in the SC-FDMA scheme further includes an N-point DFT module 402. The N-point DFT module 402 partially offsets the IDFT processing impact of the M-point IDFT module 404 so that the transmitted signal has a single carrier property.

SC-FDMA must satisfy the single carrier property. 5 illustrates examples of mapping input symbols to subcarriers in the frequency domain while satisfying a single carrier characteristic. According to one of FIGS. 5A and 5B, when a DFT symbol is allocated to a subcarrier, a transmission signal satisfying a single carrier property may be obtained. FIG. 5 (a) shows a localized mapping method and FIG. 5 (b) shows a distributed mapping method.

Meanwhile, a clustered DFT-s-OFDM scheme may be adopted for the transmitters 100a and 100b. Clustered DFT-s-OFDM is a variation of the conventional SC-FDMA scheme. The clustered DFT-s-OFDM is a method in which a signal passed through a precoder is split into several subblocks and then discontinuously mapped to a subcarrier. 6 to 8 illustrate examples in which input symbols are mapped to a single carrier by clustered DFT-s-OFDM.

FIG. 6 illustrates a signal processing procedure in which DFT process output samples are mapped to a single carrier in clustered SC-FDMA. 7 and 8 illustrate a signal processing procedure in which DFT process output samples are mapped to multi-carriers in a clustered SC-FDMA. FIG. 6 illustrates an example of applying intra-carrier clustered SC-FDMA, and FIGS. 7 and 8 correspond to an example of applying inter-carrier clustered SC-FDMA. FIG. 7 illustrates a case in which a signal is generated through a single IFFT block when subcarrier spacing between adjacent component carriers is aligned in a situation in which component carriers are contiguous in the frequency domain. FIG. 8 illustrates a case where a signal is generated through a plurality of IFFT blocks in a situation in which component carriers are allocated non-contiguous in the frequency domain.

9 illustrates a signal processing procedure of a segmented SC-FDMA.

Segment SC-FDMA is simply an extension of the conventional SC-FDMA DFT spreading and IFFT frequency subcarrier mapping configuration as the number of IFFTs equal to the number of DFTs is applied and the relationship between DFT and IFFT is one-to-one. Sometimes referred to as -FDMA or NxDFT-s-OFDMA. This specification collectively names them Segment SC-FDMA. Referring to FIG. 9, the segment SC-FDMA performs a DFT process on a group basis by grouping all time-domain modulation symbols into N (N is an integer greater than 1) groups to alleviate a single carrier characteristic condition.

10 illustrates examples of a radio frame structure used in a wireless communication system. In particular, FIG. 10 (a) illustrates a radio frame according to frame structure type 1 (FS-1) of 3GPP LTE / LTE-A system, and FIG. 10 (b) shows a frame structure type of 3GPP LTE / LTE-A system. 2 illustrates a radio frame according to (FS-2). The frame structure of FIG. 10 (a) may be applied to a frequency division duplex (FDD) mode and a half FDD (H-FDD) mode. The frame structure of FIG. 10B may be applied in a time division duplex (TDD) mode.

Referring to FIG. 10, a radio frame used in 3GPP LTE / LTE-A has a length of 10 ms (307200 Ts) and is composed of 10 equally sized subframes. Each of 10 subframes in one radio frame may be assigned a number. Here, T _s represents a sampling time and is expressed as T _s = 1 / (2048x15 kHz). Each subframe has a length of 1 ms and consists of two slots. 20 slots may be sequentially numbered from 0 to 19 in one radio frame. Each slot is 0.5ms long. The time for transmitting one subframe is defined as a transmission time interval (TTI). The time resource may be classified by a radio frame number (or radio frame index), a subframe number (or subframe number), a slot number (or slot index), and the like.

The radio frame may be configured differently according to the duplex mode. For example, in the FDD mode, since downlink transmission and uplink transmission are divided by frequency, a radio frame includes only one of a downlink subframe and an uplink subframe.

On the other hand, in the TDD mode, downlink transmission and uplink transmission are classified by time, and thus, subframes within a frame are divided into downlink subframes and uplink subframes.

11 shows an uplink subframe structure to which the present invention is applied. Referring to FIG. 11, an uplink subframe may be divided into a control region and a data region in the frequency domain. At least one physical uplink control channel (PUCCH) may be allocated to the control region for transmitting uplink control information (UCI). In addition, at least one physical uplink shared channel (PUSCH) may be allocated to the data area for transmitting user data. However, when the UE adopts the SC-FDMA scheme in LTE release 8 or release 9, PUCCH and PUSCH cannot be simultaneously transmitted in order to maintain a single carrier characteristic.

The uplink control information (UCI) transmitted by the PUCCH differs in size and use according to the PUCCH format. In addition, the size of the uplink control information may vary according to the coding rate. For example, the following PUCCH format may be defined.

(1) PUCCH Format 1: On-Off Keying (OOK) Modulation, Used for Scheduling Request (SR)

(2) PUCCH formats 1a and 1b: used to transmit ACK / NACK (Acknowledgment / Negative Acknowledgment) information

1) PUCCH format 1a: 1 bit ACK / NACK information modulated with BPSK

2) PUCCH format 1b: 2-bit ACK / NACK information modulated by QPSK

(3) PUCCH format 2: modulated with QPSK and used for CQI transmission

(4) PUCCH formats 2a and 2b: used for simultaneous transmission of CQI and ACK / NACK information

Table 1 shows a modulation scheme and the number of bits per subframe according to the PUCCH format. Table 2 shows the number of RSs per slot according to the PUCCH format. Table 3 shows SC-FDMA symbol positions of the RS according to the PUCCH format. In Table 1, PUCCH formats 2a and 2b correspond to a case of normal CP.

In an uplink subframe, subcarriers having a long distance based on a direct current (DC) subcarrier are used as a control region. In other words, subcarriers located at both ends of the uplink transmission bandwidth are allocated for transmission of uplink control information. The DC subcarrier is a component that is not used for signal transmission and is mapped to a carrier frequency f ₀ during frequency upconversion by the OFDMA / SC-FDMA signal generator.

PUCCH for one UE is allocated to an RB pair in a subframe, and the RBs belonging to the RB pair occupy different subcarriers in two slots. The PUCCH allocated in this way is expressed as that the RB pair allocated to the PUCCH is frequency hopped at the slot boundary. However, if frequency hopping is not applied, the RB pair occupies the same subcarrier in two slots. Regardless of whether or not frequency hopping, since the PUCCH for the UE is allocated to the RB pair in the subframe, the same PUCCH is transmitted twice, once through one RB in each slot in the subframe.

Hereinafter, an RB pair used for PUCCH transmission in a subframe is called a PUCCH region. In addition, a PUCCH region and a code used in the region are referred to as PUCCH resources. That is, different PUCCH resources may have different PUCCH regions or different codes within the same PUCCH region. In addition, for convenience of description, a PUCCH for transmitting ACK / NACK information is called ACK / NACK PUCCH, a PUCCH for transmitting CQI / PMI / RI information is called CSI (Channel State Information) PUCCH, and SR information is called. The PUCCH to be transmitted is called SR PUCCH.

The terminal is allocated a PUCCH resource for transmission of uplink control information from the base station by an explicit method or an implicit method.

The uplink link control information (UCI) such as ACK / NACK (ACKnowlegement / negative ACK) information, channel quality indicator (CQI) information, precoding matrix indicator (PMI) information, rank information (RI) information, and scheduling request (SR) information It can be transmitted on the control region of the uplink subframe.

In a wireless communication system, a terminal and a base station transmit and receive signals or data. When the base station transmits data to the terminal, the terminal decodes the received data and, if the data decoding is successful, transmits an ACK to the base station. If the data decoding is not successful, send a NACK to the base station. The opposite case is also the case when the terminal transmits data to the base station. In a 3GPP LTE system, a terminal receives a PDSCH from a base station and transmits an ACK / NACK for the PDSCH to the base station through an implicit PUCCH determined by the PDCCH carrying scheduling information for the PDSCH. In this case, if the terminal does not receive the data, it may be regarded as a discontinuous transmission (DTX) state, and is treated as if there is no data received according to a predetermined rule or NACK (data is received, but decoding is not successful. Case).

Meanwhile, the transmitters 100a and b may include an encoder, a mapper, and a DPC group.

At this time, the encoder encodes the input information bits through the channel code. In addition, the channel code may include a function for correcting an error caused by the channel. Encoding by the encoder may be a variety of methods, such as block-type encoding and trellis-type encoding. Trellis-type encoding may include turbo coding or convolutional coding.

The mapper also maps the coded data to a data symbol w that represents a location on the signal constellation. The data symbol w is a symbol encoded and constellation mapped to data to be transmitted to each user. If there is no restriction on the modulation scheme performed by the mapper, it may be m-Phase Shift Keying (m-PSK) or m-Quadrature Amplitude Modulation (m-QAM). For example, m-PSK may be Binary Phase Shift Keying (BPSK), Quadrature Phase Shift Keying (QPSK), or 8-PSK. The m-QAM may be 16-QAM, 64-QAM, or 256-QAM.

In addition, the DPC machine generates a transmission signal M by performing DPC on the data symbol w input from the mapper. The DPC may include an interference deductor and a modulo operator. The interference deductor deducts the interference S known to the transmitters 100a, b for the mapped data symbol w. When the data symbol to be sent to the user is called w and the interference previously known by the transmitters 100a and b is S, the interference deductor performs w-S to subtract the interference S.

It is necessary to change the shape of the transmission signal set in order for the DPC to operate on any interference S. This is because if the interference S is large, it may exceed the transmit power constraint. In consideration of this, a modulo operation for changing the shape of the transmission signal set is performed.

The modulo operator performs a modulo operation on the output w-S of the interference deductor and outputs the transmission signal M. FIG. For example, when a QPSK symbol has a form of ± 1 ± j in a constellation using QPSK, modulo operation allows -1 + j to be -5-3j, -5 + j, + 3 + j, +3+. 5j and the like operate as if they are the same symbol to be prepared for any interference S. The transmission signal M output from the modulo operator is expressed by Equation 1 below.

[Equation 1]

Where a and b are integers and L is half the size of the modulo box. The transmitter 100a subtracts and transmits an interference signal in advance. In this case, since transmission power may increase due to interference, transmission power may be limited by modulo operation.

The receivers 300a and b may include a modulo operator, a demapper and a decoder. The modulo operator performs a modulo operation on the input signal R to output the received signal Y. The input signal R is the sum of the noise N and the interference S as the transmitted signal M passes through the channel, resulting in R = M + S + N. Therefore, the input signal R is as follows.

&Quot; (2) "

In addition, when the modulo operation is performed on the input signal R, the following equation 3 may be obtained.

&Quot; (3) "

Where c and d are integers and L is half the size of the modulo box. The modulo operator of the receiver moves the received signal Y into a rectangular modulo box having a length of 2L each.

The demapper demaps the received signal Y. The demapper may convert the symbol level information into the bit level by estimating the received signal as one of coordinates in the constellation. The decoder decodes the symbols demapped by the demapper and restores the original information bits.

12 illustrates a demapping method according to an embodiment of the present invention.

Referring to FIG. 12, a modulo operation is performed on the received signal (S100). Modulo operation is performed on the input signal R to the receiver to output the received signal Y.

A candidate function value is generated for the received signal Y, which is a signal on which the modulo operation is performed, using the candidate constellation point on the extended constellation, and a maximum function value is selected from the generated candidate function values (S110). The candidate function value refers to a value indicating the likelihood that the received signal Y corresponds to the candidate constellation point on the extended constellation. An extended constellation is a constellation that includes an extended constellation as well as basic constellations on the basic constellation. The maximum value of the candidate function value is called the maximum function value, and the maximum function value is selected from the candidate function value.

13 shows an example of generating an extended constellation according to an embodiment of the present invention. With reference to FIG. 13, an example using the basic constellation of 16-QAM is demonstrated. A constellation composed of a plurality of constellation points corresponding to symbols in the modulo box 300 is called a basic constellation, and the basic constellation in 16-QAM is 16 constellation points in the modulo box 300. It becomes the constellation which consists of.

The extended constellation includes the basic constellations and the extended constellations of the basic constellation. Extended constellation points refer to constellation points other than the basic constellation points. The modulo box 300 refers to an edge indicating an area where basic constellation points are located.

An example of generating an extended constellation point is to repeat the modulo box and place it adjacent to the modulo box 300 of the basic constellation. Eight modulo boxes can be arranged adjacent to one basic constellation, and the arrangement of constellation points included in the eight modulo boxes is the same as the arrangement on the basic constellation. Repeating the modulo box here means copying the constellation points inside the modulo box.

Herein, the constellation points included in the eight modulo boxes may be selected to be extended constellation points. For example, by repeatedly arranging four modulo boxes up, down, left, and right, a total of 16 extended constellation points can be obtained by selecting four constellation points close to the origin of constellation among constellation points located in each modulo box. have.

As another example, one of the basic constellation points, which is close to the constellation points near the edge, may be selected as the extended constellation point. Therefore, it is possible to select an extended constellation point located close to the modulo box of the basic constellation.

As another example, the basic constellation points may be repeatedly arranged vertically or horizontally. Repeated placement means that the basic constellation points are copied up, down, left, or right, with the same spacing and with the same spacing. Therefore, one or more extended constellation points can be selected from these copied basic constellation points.

As described above, by generating one or more extended constellation points, the extended constellation can be obtained by placing the constellation together with the basic constellation point. In generating extended constellations, various m-PSK or m-QAM basic constellations such as BPSK, QPSK, 8-QAM, 16-QAM, 64-QAM, 128-QAM, and 256-QAM can be used. In addition, in the basic constellation, the position of the constellation point on the constellation corresponding to the bit string may vary.

FIG. 14 illustrates an example in which candidate constellation points are arranged in the extension generation diagram generated in FIG. 13. Referring to FIG. 4, the extended generation diagram includes basic constellation points on the basic generation diagram and extended constellation points located close to the modulo box 300. Eight modulo boxes are arranged adjacent to one basic constellation, and the arrangement of constellation points included in the eight modulo boxes is the same as the arrangement of constellation points on the basic constellation.

One or more constellation points that represent the same bit string are called candidate constellation points, and the candidate constellation points may include one or more constellation points on the extended constellation diagram.

비트열을 나타내는 기본 성상점은 왼쪽 측면에 위치하는

비트열을 나타내는 확장 성상점을 생성하여 배치할 수 있다. 이는 모듈로 박스(300)가 반복 배치되더라도, 모듈로 박스(300) 내의 성상점들의 배치가 기본 성상도 상의 배치와 동일하기 때문이다. 따라서 (0010) 비트열을 나타내는 후보 성상점은

비트열을 나타내는 기본 성상점과

비트열을 나타내는 확장 성상점이다.Side constellations located on the side of the base constellation rather than on the corners may generate one extended constellation point. For example, on the right side

The default constellation point representing the string of bits is located on the left side.

An extended constellation point representing a bit string may be generated and arranged. This is because even if the modulo box 300 is repeatedly arranged, the arrangement of constellation points in the modulo box 300 is the same as the arrangement on the basic constellation. Therefore, the candidate constellation point representing the bit string is

The default constellation point representing the string of bits

An extended constellation point representing a bit string.

비트열을 나타내는 기본 성상점은 왼쪽 상부에 위치하는

비트열, 오른쪽 하부에 위치하는

비트열 및 왼쪽 하부에 위치하는

비트열을 나타내는 각각의 확장 성상점을 생성할 수 있다. 따라서 (0011) 비트열을 나타내는 후보 성상점은

비트열을 나타내는 기본 성상점과

비트열,

비트열 및

비트열을 나타내는 세 개의 확장 성상점이다.The corner constellation point located at the corner of the basic constellation point may generate three extended constellation points. For example, located in the upper right corner

The default constellation point representing the string of bits is located at the top left

Bit string, located in the lower right corner

Bit row and lower left

Each extended constellation point representing a bit string can be generated. Therefore, the candidate constellation point representing the bit string is

The default constellation point representing the string of bits

Bit String,

Bit string and

Three extended constellation points representing bit strings.

비트열을 나타내는 기본 성상점은 확장 성상점을 생성하지 않는다. 왜냐하면, 모듈로 박스를 반복하여 배치하면서 모듈로 박스(300) 내의 성상점들의 배치가 기본 성상도 상의 배치와 동일하기 때문이다. 따라서 (0000) 비트열을 나타내는 후보 성상점은

비트열을 나타내는 기본 성상점 하나이다.Among the basic constellation points, the constellation points other than the corner constellation point or the peripheral constellation point become inner constellation points and do not generate extended constellation points. For example, located near the origin on the constellation

The default constellation point representing the bit string does not create an extended constellation point. This is because the arrangement of constellation points in the modulo box 300 is the same as the arrangement on the basic constellation while repeatedly arranging the modulo boxes. Therefore, the candidate constellation point representing the (0000) bit string is

One basic constellation point that represents a bit string.

As such, the candidate constellation points are one or more constellation points representing the same bit string, and may include basic constellation points and extended constellation points. However, when there is no extended constellation point representing a specific bit string, only the basic constellation point may be a candidate constellation point.

15 is an example of an extended constellation diagram in QPSK. Referring to FIG. 15, in a typical QPSK constellation diagram, the length and width of the module are mapped to one of four points in the modulo box 300 having a length of 2L. The basic constellation in QPSK is a constellation of four basic constellation points in the modulo box 300.

The symbols mapped on the basic constellation map are mapped to the real part I and the imaginary part Q by 1 bit, respectively, and correspond to a total of 2 bits of information. For example, consider the case where the real axis affects the preceding bit and the imaginary axis affects the later bit. In other words, if the preceding bit is zero, the real value of the signal has A, and if it has 1, it has -A. On the other hand, if the next bit is zero, the imaginary value of the signal has A, and if it has 1, it has -A. The basic constellation point located on the basic constellation map corresponds to a specific bit string, and the same constellation point may correspond to another basic constellation point on the basic constellation in some cases.

The extended constellation includes the basic constellations and the extended constellations of the basic constellation. The extended constellation of FIG. 5 is generated by repeating the modulo box of the basic constellation and arranging adjacent to the modulo box of the basic constellation, and then selecting the extended constellation points disposed close to the constellation points of the basic constellation. .

및

의 네 개이다. 디맵핑 될 비트열이 (11)이라면, 확장 성상도에서의 후보 성상점들은

및

의 네 개이다.The extended constellation has candidate constellation points, which are one or more constellation points corresponding to the same bit string. In the basic constellation diagram, one constellation point corresponding to the bit string to be demapped in the modulo box 300 is located. In the extended constellation diagram, one or more constellation points corresponding to the bit string to be demapped are located. For example, if the bit string to be demapped is (00), then the candidate three points in the extended constellation

And

Four of them. If the bit string to be demapped is (11), then the candidate constellation points in the extended constellation

And

Four of them.

Using the candidate constellation point on the extended constellation diagram, the candidate function value Pi can be calculated by the following equation. The candidate function value Pi expresses numerically the likelihood that the received signal on which the modulo operation is performed corresponds to a specific bit string.

&Quot; (4) "

은 잡음 분산, y는 모듈로 연산이 수행된 수신 신호 및 h는 채널 응답을 말한다. 수학식 4에서의 후 보 함수 값(Pi)은 α의 비트열을 만족하는 후보 성상점의 좌표 (α)i 를 대입하여 구할 수 있다. 예를 들어, 디맵핑 될 비트열 α가 (00)이라면,

의 후보 성장점들은

및

의 네 개의 성상점이고,

은 기본 성상점이며

및

은 세 개의 확장 성상점이다. 따라서 이들 네 개의 성상점here,

Is the noise variance, y is the received signal on which the modulo operation is performed, and h is the channel response. The candidate function value Pi in Equation 4 can be obtained by substituting the coordinate α of the candidate constellation point satisfying the bit string of α. For example, if the bit string α to be demapped is (00),

Candidate Growth Points

And

Are four constellations of,

Is the default constellation point

And

Are three extended constellation points. Therefore, these four constellation points

Four candidate function values Pi may be generated according to Equation 4 above.

및

의 네 개의 성상점들이 될 수 있다. 각각에 후보 성상점에 대한 성상도 상에서의 좌표인

의 값을 대입하여In other words, if the bit string α to be demapped is (00), the corresponding candidate constellation points are (

And

Can be four constellation points. Each of which is the coordinate on the constellation for the candidate constellation

By substituting for

를 만족하는 경우에 후보 함수 값(Pi)는 최대가 될 수The candidate function value Pi is obtained. here,

The candidate function value Pi can be maximized when

have. The maximum candidate function value Pi means that the constellation symbol transmitted from the transmitter corresponds to the corresponding candidate constellation point.

This means that it is most likely a positive bit string. Thus, among one or more candidate constellation points in the extended constellation

를 가장 근사하게 만족시키는 후보 성상점에 의한 후보 함수 값(Pi)을 구하면, 그 때의 성Format

If the candidate function value Pi with the candidate constellation point that satisfies most is found, then

The store is most likely to be a specific bit string.

)을 다음 수학식에 의해 구할 수 있다.The maximum function value that becomes the maximum of the generated candidate function values (Pi) (

) Can be obtained by the following equation.

[Equation 5]

)을 이용하여 로그 우도 비율(Log-likelihood ratio; 이하 LLR)을 계산한다(S120). 디맵퍼에서의 출력으로는 2진수로 표현되는 것이 아닌 입력 비트열에서의 1과 0에 대한 신뢰도 또는 확률 값으로 표현될 수 있다. 따라서 다음 수학식을 이용하여 k-번째 비트에 대한 1 또는 0에 대한 신뢰도 또는 확률을 비율로 나타내는 LLR을 계산할 수 있다.Referring back to FIG. 12, the maximum function value (

Log-likelihood ratio (hereinafter referred to as LLR) is calculated using (S120). The output from the demapper can be expressed in terms of confidence or probability values for 1s and 0s in the input bitstream, not in binary. Therefore, the following equation can be used to calculate the LLR representing the reliability or probability of 1 or 0 for the k-th bit as a ratio.

&Quot; (6) "

는 α의 실수 값에 맵핑되는 k-번째 비트가 가리키는 것이 0 인 심벌이고,

는 α의 실수 값에 맵핑되는 k-번째 비트가 가리키는 것이 1 인 심벌이다. 수학식 6에서의 분모는 성상 심벌의 좌표 중 하나인 α가 맵핑되는 k-번째 비트가 0 이 되는 최대 함수 값들의 합이며, 분자는 성상 심벌의 좌표 중 하나인 α가 맵핑되는 k-번째 비트가 1 이 되는 최대 함수 값들의 합이다. 따라서 분모가 커지면 LLR은 낮아지고, 분자가 커지면 LLR은 높아진다. LLR이 낮다는 의미는 성상심벌의 좌표 중 하나인 α가 맵핑되는 k-번째 비트가 0 이 될 확률이 높다는 의미이고, LLR이 높다는 의미는 성상 심벌의 좌표 중 하나인 α가 맵핑되는 k-번째 비트가 1 이 될 확률이 높다는 의미이다.Where α is one of the coordinates of the constellation symbol on the constellation map,

Is a symbol whose 0 is the k-th bit mapped to the real value of α,

Is a symbol whose k-th bit is mapped to the real value of α. The denominator in Equation 6 is the sum of the maximum value of the function whose k-th bit to which α, one of the coordinates of the constellation symbol is mapped, becomes 0, and the numerator is the k-th bit to which α, which is one of the coordinates of the constellation symbol, is mapped. Is the sum of the maximum function values, where 1 is 1. Therefore, the larger the denominator, the lower the LLR, and the larger the numerator, the higher the LLR. The low LLR means that the k-th bit that α is mapped to one of the constellation symbols is highly likely to be zero, and the high LLR means the k-th to which α, one of the coordinates of the constellation symbol is mapped. This means that the bit is likely to be one.

)을 계산할 수 있다.The maximum function value Pmax in Equation 6 may be calculated using Equation 5. The maximum function value satisfying Equation 5 using the candidate constellation points on the extended constellation corresponding to a specific bit string (

) Can be calculated.

For example, in order to calculate the LLR of the first bit when demapping a QPSK mapped symbol, the following equation may be calculated using Equation 7 below.

[Equation 7]

)의 합과 분자에는 첫 번째 비트가 1이 되는 비트열인 (10)과 (11)에 대한 최대 함수 값(

)의 합으로 이루어진 수에 로그를 취하여 계산한다.In calculating the LLR of the first bit, the denominator contains the maximum function values for (00) and (01), which are the string of bits in which the first bit is zero

) And the sum of the numerators contain the maximum function values ((10) and (11) for the bit strings where the first bit is 1

Calculate by taking the logarithm of the sum of

)을 수학식 7에 대입하면 첫 번째 비트에 대한 LLR 값을 계산할 수 있다. 첫 번째 비트가 0 이 될 가능성이 크다면 LLR은 낮아지게 되고, 첫 번째 비트가 1 이 될 가능성이 크다면 LLR은 높아진다.In other words, the maximum function value of each symbol (

) Into Equation 7 to calculate the LLR value for the first bit. LLR goes low if the first bit is likely to be zero, and LLR goes high if the first bit is likely to be 1.

To calculate the LLR of the second bit can be calculated using Equation 8.

[Equation 8]

)의 합과 분자에는 첫 번째 비트가 1이 되는 비트열인 (01)과 (11)에 대한 최대 함수 값(

)의 합으로 이루어진 수에 로그를 취하여 계산한다. 두 번째 비트가 0 이 될 가능성이 크다면, 수학식 8의 LLR은 낮아지게 되고, 두 번째 비트가 1 이 될 가능성이 크다면, 수학식 8의 LLR은 높아진다.In calculating the LLR for the second bit, the denominator contains the maximum function values for (00) and (10), which are the string of bits in which the first bit is 1

) And the sum of the numerators contain the maximum function values (() and (11)

Calculate by taking the logarithm of the sum of If the second bit is likely to be zero, the LLR of Equation 8 is low, and if the second bit is likely to be one, the LLR of Equation 8 is high.

Therefore, by calculating the LLR (bR, 1) and the LLR (bR, 2) it can be calculated the ratio of the reliability or probability that will be 0 or 1 for the first bit and the second bit. Therefore, the demapper may demap the received signal on the constellation by LLR (bR, 1) and LLR (bR, 2).

)을 구하고, 이를 바탕으로 LLR 값을 구하여 디맵핑한다. 확장 성상도를 도입함으로써 수학식 4에 의해 계산되는 후보 함수 값(

)을 더욱 정확히 계산할 수 있다. 또한 모듈로 연산에 의하여 성상 심벌이 모듈로 박스 내로 이동됨에 의하여 유발되는 오류를 하나 이상의 후보 성상점을 적용하여 최대 함수 값(

)을 구함으로써 줄일 수 있다. 따라서 최대 함수 값()을 이용하는 LLR 계산에 있어서, 상대적으로 정확한 LLR 값을 계산할 수 있다.As described above, the demapping method according to an embodiment of the present invention uses a maximum function value using the candidate constellation point of the extended constellation.

), And then demap the LLR value based on this. By introducing extended constellations, candidate function values calculated by

) Can be calculated more accurately. In addition, by applying one or more candidate constellation points to errors caused by moving constellation symbols into the modulo box by modulo operation, the maximum function value (

Can be reduced by Therefore, the maximum function value ( For LLR calculations using < RTI ID = 0.0 >), a relatively accurate LLR value can be calculated.

Therefore, the receiver can demap the constellation symbols transmitted from the transmitter by extracting the position of the received signal on the constellation derived by modulo operation relatively accurately. Therefore, the performance deterioration can be reduced and the performance can be improved in the demapper.

Table 4 shows Signal to Noise Ratio (SNR) according to MCS level in order to satisfy the 10% Frame Error Rate (FER). Here, the conventional technique is an SNR value (dB) required for demapping by a general demapping method by receiving a signal on which the DPC is performed, and the proposed technique is a demapping method according to an embodiment of the present invention. It represents the SNR value (dB) required at the time of demapping.

Referring to Table 4, the 22 MCS levels have different coding schemes and modulation schemes. The rate is determined by the MCS level, which is the level for a predefined modulation and channel coding combination. The MCS level is determined according to the received SNR, and the MCS level having the highest efficiency is selected according to the SNR.

The SNR value in the proposed scheme to satisfy the 10% frame error rate at all MCS levels is lower than the SNR value in the conventional scheme. It is shown that the proposed technique according to an embodiment of the present invention under the condition of satisfying the same frame error rate may have a lower SNR value than that of the conventional technique.

For example, at the MCS 0 level, an SNR of about 3.46 dB is required for the conventional scheme, and an SNR of 2.31 dB is required for the proposed scheme. Therefore, it can be seen that the performance improvement of about 1.15 dB is achieved by the proposed technique. In the MCS 17 level, the proposed technique produces about 8.80 dB in SNR compared to the conventional scheme. On average, a performance improvement of about 4.23 dB occurs.

As such, the SNR for satisfying the 10% frame error rate is relatively lower than that of the conventional technique in the case of the proposed technique. In order to satisfy the same frame error rate, it can be seen that a relatively higher SNR value is required in the case of performing the conventional scheme. In other words, it can be seen that the performance deterioration can be reduced in the case of demapping by the proposed technique, compared to the case of demapping by the conventional technique.

This is because the function value is calculated using only the basic constellation point on the basic constellation in case of demapping by performing the existing technique. The function value P by performing the DPC can be obtained by the following equation.

[Equation 9]

Equation 9 uses the received signal y that is moved into the modulo box when performing a modulo operation on the input signal.

However, the received signal obtained by performing the modulo operation may be moved to a position other than the position on the constellation at the time of transmission while the noise is included. In other words, while moving the received signal into the modulo box by modulo operation, the function value P may be changed by Equation 9 by moving to a position other than the position of the constellation symbol during transmission. Therefore, this may cause a significant error in obtaining the LLR value.

Hereinafter, an error caused in demapping transmission symbols transmitted by the DPC will be described in detail.

로서 약 0.707의 값을 가진다.FIG. 16 shows a received signal in which a modulo operation is performed on a symbol mapped to QPSK when noise is a Gaussian distribution. In the transmitter, when the symbol of (00) is transmitted and the SNR is 5 dB, the distribution of the real value of the received signal after the modulo operation is obtained by simulation. Where the value of A is

It has a value of about 0.707.

Here, the horizontal axis represents a real value in constellation and the vertical axis represents a received signal on which a modulo operation is performed.

Referring to FIG. 16, the (00) symbol may include a noise of a Gaussian distribution while being transmitted. Modulo operation is performed on the received signal containing noise. After the modulo operation is performed, the noise no longer has a Gaussian distribution.

This is because a received signal that is out of the modulo box by a modulo operation is forced into the modulo box.

Accordingly, a received signal having a Gaussian distribution is obtained in the range of ± 0.5 based on the value of A, about 0.707, and if it is out of this range, it no longer has a Gaussian distribution. It can also be seen that the right end is forcibly cut off by modulo operation.

As described above, when the real value of the signal received by the modulo operation is A, the LLR value is highest, and when the real value of the received signal is outside A, the LLR value is lowered like the Gaussian distribution. However, it can be seen that the left portion does not have a Gaussian distribution and thus has a tail that increases again after the LLR of the received signal becomes zero.

17 is a diagram illustrating an example of DPC (Dirty Paper Coding) using a modular operator combined with channel encoding.

Referring to FIG. 17, a DPC using a Modulo operator combined with channel encoding operates in the following order.

First, the information data of the user becomes D through channel coding, and is mapped to a constellation point to X.

Subsequently, the known interference S is subtracted here. By subtracting the known interference, the energy of the signal can increase, so a modulo operation can send the signal to a point in a particular box.

The signal after the modular operation is passed through the channel, adding to the known interference and adding additional white Gaussian noise (AWGN) to the receiver's received signal (called AWGN). M + S + N).

)가 된다.This received signal is again subjected to a modular operation (

)

) 채널 코드 디코딩(channel code decoding) 이 수행된 후, 전송된 데이터가 추정된다.In addition, through de-mapping (demapping) to convert the symbol level information (information) to bit level information (

After channel code decoding is performed, the transmitted data is estimated.

In the above-described process of FIG. 17, the demapper estimates the received signal as one of the coordinates in the constellation and converts the symbol level information into the bit level. At this time, a hard decision method (hard output) for determining a value of 1 or 0 and a soft decision method (soft output) for a probability value may be applied.

이고 이때 noise의 variance를

라고 하면 확률 값, Log-likelihood ratio(LLR)은 다음과 같이 표현된다.Equation at the time of outputting the soft value to the output is the same as the above equations (4) and (6). In other words, when a signal to be transmitted is x, a channel response is h and an additive white Gaussian noise (AWGN) is n, the received signal y is

Where the variance of noise

In this case, the probability value, Log-likelihood ratio (LLR) is expressed as follows.

은 α의 실수 값에 매핑(mapping)되는 k 번째 bit가 가리키는 것이 0 bit일 symbol이고,

은 α의 실수 값에 mapping되는 k 번째 bit가 가리키는 것이 1 bit일 symbol이다.Where α is one of the coordinates on the constellation

Is the symbol where the kth bit mapped to the real value of α is 0 bit,

Denotes the symbol where the kth bit mapped to the real value of α is 1 bit.

For example, assume that QPSK symbols are mapped to constellations as shown in FIG. 18.

Referring to FIG. 18, the number of bits mapped to the total real part and the imaginary part is 1 bit each, so that 2 bits of information correspond to the points on the four constellations.

The real axis affects the preceding bit and the imaginary axis affects the later bit. In other words, if the preceding bit is 0, the real value of the signal has A, and if it has 1, it has -A. On the other hand, if the following bit is 0, the imaginary value of the signal has A, and if it has 1, it has -A.

In other words, to calculate the probability value of the first bit, the following equation is solved.

That is, the probability that x = α is the largest probability value among the candidate ( α ) _i of repeated α . candidate ( α ) _i are constellation points (repeated) extended by modulo operation.

In other words,

It can be calculated as Here the original constellation is BPSK (+ -1 transmission) and L is the size of the modulo box.

In addition, in terms of encoding and decoding, channel encoding includes block code sequences such as LDPC code and Hamming code, convolutional code, and turbo. There is a trellis family such as code. In addition, the LLR value derived from the demapping enters the decoder stage and decoding is performed.

One of algorithms for implementing DPC (dirty paper coding) is a modular operation method. In other words, it is necessary to remove the interference to be added from the transmitting end and transmit. In this case, the transmission power may be increased by the interference, thereby limiting the transmission power through a modular operation. This is described as an equation.

, 여기서, L은 modulo box 반의 크기, α와 b는 정수이다.

Where L is the modulo box half size and α and b are integers.

이 되고, 다시 modulo 연산을 수행하면,

이 된다. 이때 c와 d 역시 정수이다.therefore

If you perform the modulo operation again,

. C and d are also integers.

Next, a system model in the case of MISO-THP will be described with reference to FIG. 19.

There are two ways to calculate B matrix in the system model to which MISO-THP is applied.

First, the B matrix may be obtained using duality.

The method of obtaining the B matrix using duality is expressed as follows.

Where K is the number of users encoding at a time (assuming one stream is transmitted per user), Li is the size of a user's modulo box, and t is the number of transmit antennas. do.

At this time, the signal of the K-th user may be restored as follows.

Also, if you want to completely eliminate the interference, the G matrix can be set as follows.

If the interference is to be partially destroyed, the G matrix may be set as follows.

At this time, the i-th column of the B matrix may be obtained as follows.

In this case, the encoding may be encoded in the order from the K th user to the first user, and the K th user may not be modulo encoded.

In addition, the B matrix may be obtained through the ZF method or the QR method. The method of obtaining the B matrix through the ZF method or the QR method may be expressed as follows.

In addition, the signal of the second user is restored as follows.

At this time, if you want to completely eliminate the interference (Interference) G matrix is set as follows.

In conclusion, the B matrix is obtained as follows.

In this case, encoding is performed from the first user to the K-th user, and the first user does not perform a modulo operation.

Meanwhile, FIG. 20 is a diagram illustrating an example of mutual information which is the maximum transmission value in each SNR according to the present invention.

Referring to FIG. 20, it can be seen that performance is low at low SNR due to a modulo operation.

This is because the constellation is repeatedly generated on the constellation as described above, and thus the number of nearest neighbors affecting the performance increases.

In addition, through a modulo operation, power is uniformly distributed in the modulo box. Therefore, the actual transmit power E [M ^H M] = 2/3 L ² is larger than the original E [X ^H * X] (where L means the size of the modulo box).

Next, FIG. 21 shows an example of the performance of a system coded at each SNR in connection with the present invention.

21 shows the performance of an actual coded system.

Referring to FIG. 21, it can be seen that QPSK exhibits a performance difference of at least 1.5 dB to as much as 3.5 dB.

However, if the power of the interference is smaller than the loss coming from the modulo, it is not necessary to perform the modulo and to transmit the signal with the interference. You can see that there is a performance benefit.

Accordingly, in the present invention, it is determined whether to perform a modulo operation for each terminal adaptively according to the situation of the terminal (for example, CQI, channel state, etc.), the situation of another terminal, and modulo operation. It provides a method for notifying the terminal whether to perform).

In this case, the information that can be received by the terminal is the sum of the mutual information received by each user, the sum of the throughput, the priority of the user (for example, the priority obtained by the proportional fair algorithm). Weighted sum mutual information, weighted sum rate, weighted sum throughput, and the like.

In this case, when encoding by the THP method, the user may be informed about how many streams are transmitted in total.

In addition, the user may also transmit information on the number of streams.

If the system is ZF-THP, the first stream may not perform a task related to a modulo operation.

In addition, in the case of a THP system using duality, the last stream may not perform a task related to a modulo operation.

In addition, regardless of the THP method in the system, the first or last stream may be set to not perform an operation related to a modulo operation.

In this case, in case of another stream, whether or not to perform a modular operation may be indicated by 1 bit.

In addition, it is also possible to indicate whether or not to perform a modular operation for all streams in one bit.

Next, in the case of QR-THP, an example to which the present invention is applied will be described.

If the total available power is P, it is assumed that a total of two users are supported at the same time.

In the case of performing power control, if p_i is assigned to the i-th user, sum (p_i) <= P.

In addition, it is assumed that MI (snr_i) is mutual information in snr_i when no modulo is taken, and MI_M (snr_i) is mutual information in snr_i when modulo is taken.

In this case, since the first user does not perform a modulo operation, the maximum rate obtainable by the first user is MI (snr_1).

That is, snr_1 = p1 * | R (1,1) | ＾ 2 (Assuming that both power of noise and power of signal are scaled to 1, R (1,1) is 1,1 of matrix R. Second element).

Next, when the second user performs a modulo operation, the second user becomes MI_M (snr_2), where snr_2 = p2 * | R (2,2) | / 2 * 3 / (2L ² ).

If the second user does not perform a modulo operation, it becomes MI (snr_2 '), where snr_2' = p2 * | R (2,2) | ＾ 2 / (p1 * | R (1 , 2) | ＾ 2 + 1).

In addition, if MI_M (snr_2) is larger than MI (snr_2 '), a modulo operation may be performed for the second user, and if not, a modulo operation may not be performed.

In addition, in the example related to the present invention described above, the B matrix may be multiplied by a dedicated pilot and transmitted.

In addition, a precoding matrix B transmitted to each user or a precoding vector bi of the user may be transmitted.

On the other hand, PHY (physical layer) extraction is used to predict the performance of the link layer through an accurate and simple method.

The PHY extraction must be accurate, computer-simple, independent of any channel model, applicable in a variety of interference environments, and in the multi-antenna process. Should be applicable.

In particular, the role of the PHY extraction method is to accurately predict the coded block error rate (BLER) for OFDM subcarriers used to transmit the coded FEC block. will be.

In order to accurately predict the block error rate (BLER), the SINR values of the inputs to the FEC decoder by applying PHY compression can be considered as inputs to the PHY absorption mapping (MAPPING). have.

에 대한 새로운 개념이 필요하다.In particular, since link level curve values are generated assuming a frequency flat channel response corresponding to a given SINR, an effective SINR, i.e.

New concepts are needed.

는 환경과 무관하게 적용될 수 있는 결과적인 BLER을 결정하기 위해, 시스템 레벨의 SINR을 링크 레벨 커브들로 정확하게 매핑(mapping)하기 위한 것이다.

Is to accurately map the system level SINR to link level curves to determine the resulting BLER that can be applied regardless of the environment.

을 이용한 매핑 방법은 이펙티브 SINR 매핑(effective SINR mapping, ESM)이라는 용어를 통해 정의될 수도 있다. 이하에서는, 설명의 편의를 위해

을 이용한 매핑 방법을 ESM으로 호칭한다.Also,

The mapping method using may be defined through the term effective SINR mapping (ESM). Hereinafter, for convenience of explanation

The mapping method using is called ESM.

For example, there are two methods that can be applied to ESM.

That is, an ESM method using CQI measurement (link estimation) based on measurement and an ESM method using packet error rate estimation based on a received signal may be used.

에 대해 구체적으로 설명한다.In the following, an example of a system model is given.

It demonstrates concretely about.

인 것으로 가정한다. 이때,

는 수신한 신호 벡터인

이고,

는 채널 메트릭스인

이며,

는 전송된 심볼 스트림인

이고,

는 노이즈 벡터(또는, 인터페런스(interference)를 포함한 값일 수도 있다)는

를 의미한다.First, the system model

Assume that At this time,

Is the received signal vector

ego,

Is the channel metric

,

Is the transmitted symbol stream

ego,

Is a noise vector (or may be a value including interference)

Means.

는 다음의 수학식 10과 같이 표현될 수 있다.In the system model described above

May be expressed as Equation 10 below.

&Quot; (10) &quot;

함수는 제한된 커패시티를 통해 획득될 수 있다. 반면, EESM의 경우엔느 상기

함수는 에러 확률에 대한 체르노프 바운드(Chernoff bound on the probability of error)를 통해 구해질 수 있다.In multi-input ESM (MI-ESM, MMIB)

The function can be obtained through limited capacity. On the other hand, in the case of EESM,

The function can be obtained through the Chernoff bound on the probability of error.

함수는 다른 요소와는 상관없이 제한된 커패시터를 통해 결정된다.As mentioned above, in a MI-ESM (MMIB) environment

The function is determined by a limited capacitor regardless of other factors.

In this case, the limited capacitor may be regarded as Gaussian channel-based mutual information using Gaussian inputs.

In particular, the modulation constrained capacity may be viewed as a symbol channel (e.g., a channel constrained from input symbols due to a complex set).

This will be described in detail with reference to FIG. 23.

22 illustrates an example of a coding and decoding process using code bits in accordance with the present invention.

는 다음의 수학식 11과 같이 표현될 수 있다.Referring to Figure 22,

May be expressed as Equation 11 below.

&Quot; (11) &quot;

In addition, the mutual information in the equivalent channel may be expressed as in Equation 12 below.

[Equation 12]

는 변조 맵(modulation map)에서의 i 번째 인풋 비트와 아웃풋 LLR 사이의 뮤추얼 정보를 의미한다.here,

Denotes mutual information between the i th input bit and the output LLR in the modulation map.

In addition, mean mutual information may be expressed as in Equation 13 below.

&Quot; (13) &quot;

함수가 결정되고, 이에 따라

를 산출해낼 수 있다.Equation 10 using the above-described mutual information (mutual information)

Function is determined and accordingly

Can be calculated.

Hereinafter, mutual information applied to BPSK and QPSK will be calculated.

일 수 있다.The mutual information between the coded bit value b and the LLR can be regarded as the uncertainty for b minus the uncertainty for b in which a given LLR can be used. Also,

Lt; / RTI &gt;

Therefore, the mutual information may be expressed as in Equation 14 below.

&Quot; (14) &quot;

Equation 14 is used to calculate the mutual information applied to the BPSK and QPSK.

이고,

이며,

이고,

이 된다.

ego,

,

ego,

.

That is, y = x + n _{is, SNR = E x / N 0} = 1 / σ n 2 = 1 / (σ n, R 2/2) = With ² / σ _{n, R 2,} definition of the LLR Can be done as follows:

이고, b = 1인 경우에는

이 되며, b = 0인 경우에는 다음과 같이 된다.For BPSK, if b = 0

, If b = 1

If b = 0, it becomes as follows.

와 베리언스

를 갖는 가우시안 R.V이다.Where z is the mean

And variance

Gaussian RV with

If b = 1, simplifying the above equation,

이 된다.

.

와 베리언스

를 갖는 가우시안 R.V이다.Where z is the mean

And variance

Gaussian RV with

이고,

이 된다.In other words,

ego,

.

와 관련하여,Also,

In connection with

를 이용하면, 하기의 수학식 15와 같은 BPSK에 적용되는 뮤추얼 정보(mutual information)에 대한 식을 얻을 수 있다.therefore

By using Equation 15, an equation for mutual information applied to BPSK, such as Equation 15 below, can be obtained.

&Quot; (15) &quot;

In addition, equations for mutual information applied to QPSK can be obtained as shown in Equation 16 below.

&Quot; (16) &quot;

에서,

이 된다.Also,

in,

.

와

의 값을 갖는다.At this time,

Wow

Has the value of.

이다.Also,

to be.

이고,

이 된다. 표 5에 이에 대한 구체적인 내용을 정리하였다.At this time,

ego,

. Table 5 summarizes the details.

On the other hand, the LLR distribution (distribution) to apply the MI-ESM to the system model will be described below.

FIG. 23 is a diagram illustrating an example of an LLR distribution in connection with the present invention. FIG.

23A shows the LLR distribution when the SNR is 10 dB, a in FIG. 23 shows the LLR distribution when the SNR is 5 dB, and c in FIG. 23 shows the LLR distribution when the SNR is 0 dB.

FIG. 24 is a diagram showing an example of mutual information in each SNR according to the present invention.

In FIG. 24A, the mutual information according to the SNR is divided into BPSK, QPSK, BPSK, J1, and J2 to which a modular operation is applied.

In addition, in FIG. 24B, the following values can be obtained.

이고, QPSK에서는

이 된다.Also, in BPSK

In QPSK

.

Next, Figs. 25A and 25B show LLR distributions accurately calculated using the Mauler Gaussian noise pdf.

That is, (a) and (b) of FIG. 25 illustrate the above-described equation (10).

를 이용하여 표현된 것이다.

It is expressed using.

Considering the LLR distribution, it can be seen that applying MI-ESM to the system model shows more stable performance.

Meanwhile, FIG. 26 is a diagram illustrating an example of mutual information when a modular operation is applied and when it is not applied according to the present invention.

First, (a) of FIG. 26 shows mutual information when a modular operation is applied and when it is not applied.

Results of the graph shown in FIG. 26A are shown in Table 6 below.

In addition, Fig. 26B shows a graph of the BLER value corresponding to the SNR.

Referring to FIG. 26B, when an inverse function for mutual information that does not use a modular operation is used, an AWGN reference curve that does not use a modular operation may be used. This means that after acquiring the mutual information, the mutual information is not associated with modulation (QPSK, 16QAM, 64QAM, etc., with or without performing a modular operation).

27 is a diagram illustrating an example of mutual information for each SNR in a short term in connection with the present invention.

Next, we check the contents of the simplified LLR calculation in a graph.

28 illustrates an example of an LLR distribution according to a simplified calculation in connection with the present invention.

와 관련하여,first,

In connection with

가 된다.ego,

.

Also, with respect to z,

It is expressed as

이고, 이를 기초로 도 28을 나타낸 것이다.here

And FIG. 28 based on this.

Next, FIG. 29 is a diagram showing an example of link performance in connection with the present invention.

FIG. 30 is a diagram illustrating an example of mutual information according to different modular boxes according to the present invention.

In addition, Table 7 shows the mutual information by setting the range of the SNR from -20 to 27 and the interval to 0.5.

Regarding the mutual information, in Table 7, the first column is the result for QPSK, the second column is the result for 16QAM, and the third column is for 64QAM. Show the result.

Combined. Each component or feature shall be considered optional unless otherwise expressly stated. Each component or feature may be implemented in a form that is not combined with other components or features. It is also possible to construct embodiments of the present invention by combining some of the elements and / or features. The order of the operations described in the embodiments of the present invention may be changed. Some configurations or features of certain embodiments may be included in other embodiments, or may be replaced with corresponding configurations or features of other embodiments. It is clear that the claims that are not expressly cited in the claims may be combined to form an embodiment or be included in a new claim by an amendment after the application.

In this document, embodiments of the present invention have been described mainly based on a signal transmission / reception relationship between a terminal and a base station. This transmission / reception relationship is extended to the same / similarly for signal transmission / reception between the UE and the relay or the BS and the relay. The specific operation described herein as being performed by the base station may be performed by its upper node, in some cases. That is, it is apparent that various operations performed for communication with a terminal in a network including a plurality of network nodes including a base station can be performed by a network node other than the base station or the base station. A base station may be replaced by terms such as a fixed station, a Node B, an eNode B (eNB), an access point, and the like. In addition, the terminal may be replaced with terms such as a user equipment (UE), a mobile station (MS), a mobile subscriber station (MSS), and the like.

Embodiments in accordance with the present invention may be implemented by various means, for example, hardware, firmware, software, or a combination thereof. In the case of hardware implementation, an embodiment of the present invention may include one or more application specific integrated circuits (ASICs), digital signal processors (DSPs), digital signal processing devices (DSPDs), programmable logic devices (PLDs) field programmable gate arrays, processors, controllers, microcontrollers, microprocessors, and the like.

In the case of implementation by firmware or software, an embodiment of the present invention may be implemented in the form of a module, procedure, function, etc. that performs the functions or operations described above. The software code can be stored in a memory unit and driven by the processor. The memory unit may be located inside or outside the processor, and may exchange data with the processor by various well-known means.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS The foregoing description of the preferred embodiments of the invention disclosed herein has been presented to enable any person skilled in the art to make and use the present invention. Although the above has been described with reference to preferred embodiments of the present invention, those skilled in the art will variously modify and change the present invention without departing from the spirit and scope of the invention as set forth in the claims below. I can understand that you can. Accordingly, the present invention is not intended to be limited to the embodiments shown herein but is to be accorded the widest scope consistent with the principles and novel features disclosed herein.

Claims (20)

A method of demapping by a receiving end in a wireless communication system,
Receiving an input signal from a transmitting end and first information on whether a first modulo operation is performed on the input signal;
If the first information is information indicating that the first modulo operation is performed, obtaining a received signal by performing a second modulo operation on the input signal;
Generating a maximum function value that is most likely to correspond to a candidate constellation point on an extended constellation; And
Generating a log-likelihood ratio (LLR) using the generated maximum function value. The method of claim 1,
When the power of the interference included in the input signal is smaller than the loss power generated by performing the first modulo operation, the transmitter does not perform the first modulo operation. A demapping method. The method of claim 1,
And receiving information about the total number of streams used from the transmitting end and the order of the first stream received by the receiving end. The method of claim 3, wherein
And if the order of the first stream is first or last, the first modulo operation is determined not to be performed. The method of claim 1,
The extended constellation comprises a basic constellation point of the basic constellation and at least one extended constellation point. 6. The method of claim 5,
The extended constellation point is characterized in that the constellation point is arranged to repeat the basic constellation point, demapping method. 6. The method of claim 5,
The extended constellation point is selected from among constellation points disposed close to the basic constellation points of the basic constellation after repeatedly arranging the modulo box including the basic constellation point adjacent to the modulo box. A demapping method. The method of claim 1,
The candidate constellation point is a constellation point representing the same bit string on the extended constellation. The method of claim 1,
The maximum function value uses the following equation.

here,

Is noise variance,

Is the received signal,

Is the channel response and

Is a coordinate on the extended constellation of the extended constellation point of the candidate constellation point. The method of claim 1,
The LLR uses the following equation.

Where α is any one of the coordinates of the constellation symbol on the constellation diagram,

Is a symbol whose zero is indicated by the k-th bit, mapped to the real value of α,

Is a symbol of which the k-th bit, mapped to the real value of α, points to 1. The method of claim 1,
Generating, by the receiving end, an effective signal-to-interference-plus-noise ratio (SINR); And
Transmitting the effective SINR to the transmitting end;
The effective SINR uses the following equation.

here,

silver

th

,

silver

Total number of,

Is a function determined using the mutual information of the symbol channel,

silver

Inverse function of. 12. The method of claim 11,
The mutual information is demapping method using the following equation.

here,

Is the i-th code bit,

Is the total number of code bits,

The

LLR. A receiving end for demapping in a wireless communication system,
A receiving module for receiving an input signal from a transmitting end and first information on whether or not an operation has been performed on the input signal with a first module; And
If the first information is information indicating that the first modulo operation is performed, a processor for performing a second modulo operation on the input signal to obtain a received signal,
The processor generates a maximum function value most likely to correspond to the candidate constellation point on the extended constellation, and generates a log-likelihood ratio (LLR) using the generated maximum function value. Receiving end, characterized in that. The method of claim 13,
When the power of the interference included in the input signal is smaller than the loss power generated by performing the first modulo operation, the transmitter does not perform the first modulo operation. Receiving end. The method of claim 13,
The receiving module further receives information on the total number of streams used from the transmitting end and the order of the first stream received by the receiving end,
And the processor determines that the first modulo operation is not performed when the order of the first stream is first or last. The method of claim 13,
And the extended constellation comprises a basic constellation point of the basic constellation and at least one extended constellation point. The method of claim 13,
The processor, characterized in that for determining the maximum function value using the following equation.

here,

Is noise variance,

Is the received signal,

Is the channel response and

Is a coordinate on the extended constellation of the extended constellation point of the candidate constellation point. The method of claim 13,
And the processor determines the LLR using the following equation.

Where α is any one of the coordinates of the constellation symbol on the constellation diagram,

Is a symbol whose zero is indicated by the k-th bit, mapped to the real value of α,

Is a symbol of which the k-th bit, mapped to the real value of α, points to 1. The method of claim 13,
And a transmitting module for transmitting an effective signal to interference-plus-noise ratio (SINR) to the transmitting end.
And the processor generates the effective SINR using the following equation.

here,

silver

th

,

silver

Total number of,

Is a function determined using the mutual information of the symbol channel,

silver

Inverse function of. 20. The method of claim 19,
And the processor determines the mutual information by using the following equation.

here,

Is the i-th code bit,

Is the total number of code bits,

The

LLR.

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