KR20070021148A - Digital microphone - Google Patents

Abstract

)을 가지는 증폭기 섹션(301), 출력(

;

,

) 및 제 2 입력(φ') 사이에 접속된 피드백 필터 네트워크(Z1; Z1, Z1*, Z2)를 가지는 전치 증폭기(306)를 포함하는데, 상기 증폭기 섹션으로의 제 1 입력(φ)은 증폭기 섹션의 입력 임피던스에 의해 입력 임피던스와 관련하여 피드백 네트워크와 실질적으로 분리되는 입력 임피던스를 포함하고, 전치 증폭기는 낮은 주파수들을 억제하는 주파수-이득 변환 기능을 가지고, 안티 에일리어싱 필터링된 입력 신호를 수신하여 디지털 출력 신호(D_O) 신호를 제공하기 위해 접속된 아날로그-디지털 컨버터를 포함한다.The present invention relates to an integrated circuit configured to process microphone signals, the integrated circuit comprising a first input φ and a second input φ * and an output (

Amplifier section 301, with output (

;

,

) And a preamplifier 306 having a feedback filter network Z1 (Z1, Z1 *, Z2) connected between the second input φ ', the first input φ to the amplifier section being an amplifier. An input impedance substantially separated from the feedback network with respect to the input impedance by the input impedance of the section, the preamplifier has a frequency-gain conversion function that suppresses low frequencies, and receives the antialiased filtered input signal to It includes an analog-to-digital converter connected to provide an output signal (D _O ) signal.

Description

Digital microphone {DIGITAL MICROPHONE}

The present invention relates to an integrated circuit comprising a microphone preamplifier and an analog-to-digital converter for providing a digital output signal. The invention also relates to a digital microphone comprising a microphone element and the integrated circuit described above.

The sound pressure detected by the microphone element causes the film of the element to shake, thus changing the capacitance of the capacitor formed by the so-called backplate of the film and the microphone element. When the change in the capacitor formed by the two members continues constant, the voltage between the two capacitor members may change depending on the sound pressure acting on the film. Loading the microphone capacitance into the resistive load is not important because the change in the microphone capacitor must be constant to maintain a proportionality between the negative pressure and the voltage between the capacitor members. The resistive load discharges the capacitor, thus reducing or eliminating the capacitor performance as a microphone.

Thus, to pick up a microphone signal from a capacitor, amplifiers configured for the purpose of providing a high input resistance are preferred for buffering the capacitor from circuits used for other purposes. An amplifier connected to pick up a microphone signal is generally referred to as a preamplifier or buffer amplifier or simply a buffer. The preamplifier is generally connected physically very close to the capacitor-within a distance of several millimeters or part of a millimeter.

For small microphones, the active capacitance is very small (typically 1 to 10 pF). This generally increases the requirement for high input resistance and capacitance. Therefore, the input resistance of the preamplifiers for small microphones must be very high-the size of GΩ. In addition, the input capacitance of the amplifier must be very small in order to achieve adequate sensitivity to sound pressure.

In particular, so-called telecommunication microphones with preamplifiers are sold in bulk at very low prices. Since the price of the amplifier for the telecommunication microphone is directly related to the size of the preamplifier chip die, it is important to make the preamplifier chip die as small as possible for the purpose of reducing the price. Thus, special processing is required to make the circuits compact, which are in high demand. However, in this regard it is important to provide circuits at very low noise levels. Low noise is important because noise can be exchanged with the area, i.e. if the circuit has a low noise and the noise is lower than required, the noise level can be exchanged with a small chip die area and thus lower Preamps can be manufactured at cost.

When designing a preamplifier with CMOS technology for a microphone, there are typically three sources of noise. The noise sources are noise from bias resistors, 1 / f noise from input transistors and white noise from input transistors. We assume that input transistor noise is dominant. Both white noise and 1 / f noise can be minimized by utilizing the length and width of the input transistor (s). This fact applies to any input stage, for example a single transistor stage or a differential stage. Noise from the bias resistors can also be minimized. If the bias resistor is made very large, the noise from the resistor will be high pass filtered and the in-band noise will be very low. This is done so that the lower bandwidth limit of the amplifier is very small. This can be a problem because the input of the amplifier has settled to its nominal value after a very long period of time after the power up. Also, for example, signals with low intensity frequency content resulting from door closure or inaudible sound inside the vehicle can overload the amplifier. Another problem is that small leakage currents arise from the mounting of the die inside the microphone module. The currents are due to the extreme input impedance which forms the DC offset. The impedance reduces the overload margin of the amplifier.

Additionally, there is a need for a digital microphone that includes a microphone element and an integrated circuit with a preamplifier and an analog-to-digital converter to provide a digital output signal. In general, since telecommunication microphones are coupled to consumer electronic devices where the actual digital signal processing is performed by digital integrated circuit chips, it is desirable that signals from sensors (such as microphones) be provided as digital signals. This presents new challenges with respect to signal processing in integrated circuits in which microphones are inserted, especially with regard to distortion in the digital domain.

Recently, so-called sigma-delta modulators are very popular in implementing A / D converters. The modulators have a number of advantages: no need for high precision components; High linearity; And small die area for so-called single loop modulators. This is advantageous for forming sigma delta modulators well suited for single chip implementations.

Sigma delta modulators of a particular class are 1-bit sigma delta modulators. Modulators of this type are particularly suitable for low cost implementations because the complexity of the analog part of the A / D converter is low compared to other types of A / D converters. The complete 1-bit sigma delta converter consists of only a 1-bit analog sigma delta modulator and a digital decimation filter. Normally required higher order anti-aliasing filters may be implemented by a single RC-filter. This is because high oversampling is used, so the digital decimation filter performs anti-aliasing filtering.

1-bit sigma delta modulators are very simple to implement in the analog domain. The modulators are therefore well suited for low cost small digital microphones. Unfortunately, the modulators also have disadvantages. In particular, 1-bit sigma delta modulators generate so-called idle mode tones, which are low level tones in the audio band generated due to low frequency or DC levels at the input of the modulator. This is why one bit sigma delta modulators cannot be used despite a number of advantages. One solution can be used to eliminate the problem or to design complex modulators, but all of these solutions affect the design complexity to increase dynamically. Thus, power consumption and die area increase dynamically.

The effect of the idle-mode-tones is caused by sigma delta modulators that are less suitable for high quality audio applications. Clearly, the impact can be seen as less relevant in consumer / telecommunications applications. However, as the demand for low-cost digital microphones increases, so does the demand for performance, which can be almost equal to high-quality audio performance. Thus, the impact of idle-mode-tones causes an increase in problems for telecommunication applications.

To achieve the high performance of the digital microphone, the preamplifier of the digital microphone ASIC must have as high performance as possible, low noise, low distortion, and high dynamic ranges. According to the technology available recently, CMOS technology is essential for achieving low noise performance and shows that it can be utilized in relation to the noise of the input stage of the amplifier. In addition, the input impedance can be as large as possible to minimize noise. The input impedance is particularly potent for new thinner forms of telecommunication microphones with much lower sensitivity and cartridge capacity than previously experienced.

Unfortunately, the impedance may result in the preamplifier being able to amplify low frequency signals resulting from door closure, ie sound pressure such as vehicle noise, or the sensitivity of the microphone element changes due to humidity changes. This adds to the problem of the idle tone modes described above when a 1 bit sigma delta modulator is used. In fact, two-bit modulators with more levels will also act as above when exposed to such low frequency signals.

The low frequency signals also reduce the dynamic range and can cause intermodulation distortion because low frequency signals can cause excessive amplitude.

The problem is exacerbated as telecommunication microphones become smaller and thinner, thus requiring more gain from the preamplifier. In general, however, confused low frequency signals cannot be made smaller in amplitude. Thus the impact associated with confusion will increase.

Therefore, there is a need for the construction of a preamplifier and A / D converter suitable for thin ECM cartridges with very low cartridge sensitivity and capacity. In addition, the configuration should provide very high performance against noise, dynamic range, and distortion. In addition, the configuration can be implemented in a single chip die with less external components coupled or uncoupled and having a very small area.

In the following description, the term audio band is used. The term conventionally has various definitions with respect to the content thereof. However, the term below will generally be used to designate a frequency band having a lower corner frequency of 20 Hz to 500 Hz and an upper corner frequency of 5 KHz to 25 kHz. While specific limitations to the bands indicate design criteria, such limitations in the description below can be broadly understood.

Related Fields

The so-called two-stage preamplifier configuration, in which the filter is connected to a single buffer amplifier, has two disadvantages: the two stages result in more noise and the physical size of the filter is relatively small because no gain is provided in the first stage. It must be large. The size of the filter can be minimized by increasing the gain of the first stage, but the amplifier is sensitive to overload due to the low frequency components that do not decrease until the subsequent filter. Thus, the solution originally developed for the listening support microphone does not apply to new high sensitivity telecommunication microphones. The area of the amplifier die can be very large and therefore the device cost can be high.

The preamplifier should be as small as possible because the chip area occupied by the preamplifier must be as small as possible to obtain a relatively low cost. Thus, the amplifier configurations known from the listening support microphones are generally not used in the same range as the telecommunication microphones with respect to the chip area, so the above configurations are not applicable to telecommunications applications. It should also be appreciated that the buffers or amplifiers applied to the listening support microphones are not configured to provide high gain levels as required for the low sensitivity microphones used in telecommunications applications.

US 6,583,658 B1 discloses an adjusting circuit arrangement for converting an analog input signal from a first terminal of a capacitor microphone into a symmetrical output signal. The first operational amplifier is coupled as a voltage flower and its output is provided as the first output signal of the symmetrical output signal. The second terminal of the capacitor microphone is directly coupled to the output of the second operational amplifier, thus providing a second output signal of the symmetrical output signal. The non-inverting input of the second operational amplifier is connected to ground, but the inverting input is connected to a voltage divider providing a voltage halfway between the second and first output signals. The symmetrical output signal is provided to a sigma-delta analog-to-digital converter that provides a binary output signal.

The disclosed configuration is advantageous where relatively high voltage levels from the capacitor microphone can be processed while providing low noise. However, the capacitor microphone combined with the input impedance of the amplifier will form a filter with a very slow decreasing impulse response. When the microphone is exposed to transient sounds with large amplitude or low frequency signals, it provides signal components that in turn change very slowly, which are input to an sigma-delta type analog-to-digital converter. In a sigma-delta analog-to-digital converter, signal components will generate so-called idle mode tones in the binary output signal. The configuration is also sensitive to overload caused by non-uniform distribution of the signal spectrum generated by the microphone.

US 2002 / 0106091-A1 discloses a microphone unit having an internal analog-to-digital converter. The unit includes a sound transducer (a capacitor microphone with an electrical member), a preamplifier connected at the input to the sound transducer and at the output to an analog-to-digital converter to provide a signal input. A high pass filter located between the preamplifier and the analog-to-digital converter is configured to block the DC signals and reduce the noise level. In addition, the low pass filter may be formed as an anti-aliasing filter located between the preamplifier and the analog-to-digital converter.

Although the configuration addresses important signal processing aspects, important implementation issues related to performance with respect to cost and noise are not addressed. The disclosed configuration includes several signal processing stages. Each contributes to increasing the noise level. In addition, some stages occupy a large chip area that charges higher costs. The configuration is also sensitive to overload caused by a fairly non-uniform distribution of the signal spectrum generated by the microphone.

US 5,339, 286 discloses a preamplifier for a piezoelectric sensor. The preamplifier is configured as a fully differential amplifier with inputs connected to a piezoelectric sensor and outputs connected to an analog-to-digital converter. The preamplifier includes a common-mode filter feedback configuration that combines with the capacitance of the sensor to implement a high pass filter coupled with the preamplifier. The configuration is small in size, has relatively small noise, and the differential configuration allows silicon substrate noise (when a preamplifier is implemented on a chip) to be generated as a common mode signal which can then be removed. However, this configuration is inadequate because the gain of the amplifier is determined by the microphone. In addition, the noise of the preamplifier cannot be used independently of the microphone element.

It is therefore an object of the present invention to provide a preamplifier having the lowest possible input capacitance, the lowest possible consonant, the maximum output signal swing, and at the same time providing the minimum possible chip area.

It is an object of the present invention to provide a preamplifier with large rejection and low distortion.

It is an object of the present invention to provide an amplifier capable of processing slow changing signals with relatively large amplitude at an input terminal and at the same time amplifying low level signals to high frequencies with low distortion.

It is an object of the present invention to provide an amplifier whose performance is insensitive to leakage and plastic couplings connected to its input.

It is also an object to provide a digital output signal having a low distortion.

It is also an object to provide a preamplifier configuration that can be used independently of the microphone circuit.

There is provided an integrated circuit configured to process microphone signals, the integrated circuit including an amplifier section having a first input, a second input and an output and a feedback filter network connected between the output and the second input. The first input to the amplifier section has an input impedance that is substantially separated from the feedback network with respect to the input impedance by the input impedance of the amplifier section, and the preamplifier has a frequency-gain transmission function that suppresses low frequencies. The amplifier also includes an analog-to-digital converter connected to receive the anti-aliased filtered input signal and provide a digital output signal.

The anti-aliased filtered input signal is provided by an anti-aliasing filter connected to receive the output signal from the preamplifier, or as a result of a frequency-gain transmission function consisting of a band pass filter whose upper cutoff band prevents anti-aliasing. Is provided.

Regardless of anti-aliasing, the preamplifier is configured to provide a determined frequency-gain transmission function that is substantially unchanged regardless of the frequency-impedance characteristics of the microphone circuit connected to provide the input signal to the preamplifier. Do not. This is an important improvement because the microphone element itself is often a contradiction in the design of preamplifier and microphone circuits. In particular, because the microphone element is a mechanical component and is often difficult to control with respect to electrical properties, the independent nature of the microphone element for the frequency-gain transmission function is advantageous. This situation applies to both condenser microphone elements located as a unit spaced apart from the chip carrying the integrated circuit and MEMS microphone elements located as the micromechanical portion of the MEMS device. Since the preamplifier generally has a differential input stage (of the operational amplifier), very high impedance is achieved. Such high input impedance is not eliminated by the feedback filter, so the amplifier cannot load the microphone circuit.

In addition, the inter-modulation distortion generated at low frequencies outside the audio band by the frequency components will be very low. The loop-gain characteristic provided by the feedback arrangement provides lower distortion to other components.

The preamplifier may be implemented as a single-ended amplifier or differential amplifier or differential difference amplifier or other amplifier with several inputs and outputs.

However, in preferred embodiments the preamplifier may be configured as a differential amplifier. Thus, the preamplifier is advantageously configured to provide a differential output signal by the first and second amplifier sections, the preamplifier having a differential mode transmission function comprising a band pass characteristic. The preamplifier includes a feedback filter network that forms filter feedback paths connecting the outputs to the individual inverting inputs of the amplifier sections and a filter interconnection path that interconnects the inverting inputs.

Different configurations of preamplifiers generally provide large common-mode rejection and very high input impedance. The configuration of the feedback filter results in single circuit control of the frequency-gain transmission function. Thus, the transmission function is adjusted or adjusted to a larger free range. In addition, the different configurations and combined feedback filters allow the use of common mode rejection in a frequency dependent manner.

Preferably, the lower cutoff frequency of the filter implemented by the preamplifier is positioned to be below the lower corner frequency of the audio band. Thus, a compromise is achieved between an audio band that widens downward and a sufficiently short decay time of the impulse response of the microphone circuit and its preamplifier coupled thereto. The short decay time of the impulse response is advantageous when the influence of low frequency pulses resulting from sound pulses or electrical disturbances is reduced. Otherwise, the low frequency pulses can overload the amplifier and subsequent signal processing circuits, thus resulting in inappropriate nonlinear distortion. In a preferred embodiment, the cutoff frequency is located at about 10 Hz.

Advantageously, the preamplifier has a differential mode transmission function that includes a band pass characteristic with an upper cutoff frequency that is less than half the sampling frequency of the analog-to-digital converter. Thus, an efficient implementation of the anti-aliasing filter with respect to the chip area is provided by the preamplifier. For example, the sampling frequency may be about 2.4 MHz, and an upper cutoff frequency of about 40-70 KHz may be selected. In addition, circuit control of the cutoff frequency is provided that matches the band limit of the amplifier sections.

In one embodiment, the preamplifier has a differential mode transmission function that includes band pass characteristics having a nominal pass band and a gain plateau band, wherein the nominal pass band extends to audio band frequencies, and the gain stabilized band is It extends beyond the audio band to frequencies up to the upper cutoff frequency. Thus, noise components located beyond the audio band and resulting from acoustic / mechanical sources or electrical sources are damped. Thus, it is possible to prevent the preamplifier from being overloaded by the gain / amplitude effects of the noise signals and the resonance peak of the microphone circuit. The peak can, for example, have an amplitude of about 6 dB.

The preamplifier may have a common-mode transmission function that includes low pass characteristics. Thus, the common-mode DC output level of the preamplifier can be set at the input of the preamplifier, and the common-mode rejection ratio is used at the audio band frequencies. Due to the very high input impedance of the preamplifier, very high ohmic impedances (e.g. C <OS transistors in a weak inversion mode and linear region) that cannot substantially load the microcircuit with a DC input level through the pull-up resistor Can be set to 1-20GOhm).

In addition, the preamplifier may have a common-mode transmission function that includes cut-band characteristics, and a flat gain response is provided for low frequencies. Thus, the DC input setting is maintained while the preamplifier provides the common-mode signal at two frequency bands: low frequency (below the audio band) and high frequency (above the audio band). Thus, with respect to audio sound reproduction, efficient damping of unwanted frequency components is achieved. The cut-band will include the audio band.

In one embodiment, the preamplifier has a common-mode transfer function and a differential mode transfer function, wherein the preamplifier is configured to have a common-mode gain prevail at low frequencies, while the differential mode gain is configured to prevail at audio band frequencies. .

In addition, the common-mode gain may prevail at frequencies above the upper cutoff frequency of the bandpass characteristic.

In a preferred embodiment, the phase-shifter is cross connected between the output of the first amplifier section and the input of the second amplifier section. This is an efficient configuration that ensures that the second amplifier section operates at 180 degrees or adjacent phase difference with the first amplifier section when a dominant differential mode gain is required at the audio band frequencies. In addition, the phase-shifter may be configured to control the DC level of the input to the second amplifier section. This is achieved when the phase-shifter includes a resistive path between the input and the output of the individual amplifiers.

Alternatively, or in addition, a phase-shifter is connected between the individual inputs of the individual amplifier sections. The configuration may provide a 180 degree phase shift and optionally a DC level input to the second amplifier section.

Preferably, the preamplifier comprises a DC offset circuit coupled with a feedback filter to provide a DC shift at the output of the preamplifier. The coupling may be provided by a voltage divider connected to an AC feedback resistor, which voltage divider is actually lower than the AC feedback resistor, for example by a factor of about 1/5, 1/8, or 1/10. Has a low impedance. Optionally, DC shift can be performed by active current sources.

In addition, a DC offset circuit can be combined with a feedback filter to provide differential mode DC shift at the output of the preamplifier. The differential mode DC shift is determined by the difference in DC offsets provided by the first and second offset circuits. Thus, the so-called idle mode tones of an analog-to-digital converter in the form of a sigma-delta modulator can be controlled. The position of the idle mode tone is proportional to the differential mode DC shift (and a constant determined by one half of the sampler and sampling frequency of the sigma-delta modulator).

Preferably, the analog-to-digital converter comprises a sigma-delta modulator. A sigma-delta modulator provides a noise power spectrum that is flat for low frequencies (in an exemplary implementation) and distributed with increasing noise levels above the corner frequency but with a relatively low noise floor. Since strong over-sampling is applied, the corner frequency occurs beyond the audio band. Advantageously, the modulator provides a continuous output signal.

The sigma-delta modulator samples the differential signal provided by the preamplifier to provide a single-ended input signal for sigma-delta A / D conversion, and adjusts the DC voltage level so that the single-ended input signal is duplicated at the sampled DC voltage level. It may include a switch-capacitor sampler for sampling. This provides an easy use of the sigma-delta modulator because idle-mode tone control is performed in the modulator. Since the sampler (or control of the sampler) can already be used to sample the signal from the preamplifier, sampling of the DC level can be achieved to a degree that adds some complexity. Also, the preamplifier is not loaded with common-mode DC overhead, which in turn reduces the output AC swing.

Preferably, the sampler comprises a summing amplifier that is a combined portion of the sampler and sigma-delta modulator loop. Sigma-delta modulator loops are well known in the art, but for completeness, connections are provided to provide an integrated error signal to a quantizer that quantizes the signal to discrete levels, e.g., 2, 3, or 4 levels. Contains an integrator filter of the given order.

In addition, the summing amplifier may provide an integrated error feedback signal of the sigma-delta modulator through the first ground capacitor, and the DC voltage level is provided to the summing amplifier through the second series capacitor. Accordingly, idle mode tones may be provided by the ratio between the values of the first and second series capacitors. Idle mode tones are determined by the equation:

F _idle = (V _{DCoffset ∑ △} / V _{REF ∑ △} ) * (C _S1 / C _S2 ) * ½F _S

The F _idle is the location of the idle mode tones, impediment C _S1 and C _S2 is the value of the first and second capacitors, and F _S is the sampling frequency, V _{DCoffset Σ △} is the sampled DC voltage, V _{REF Σ △} is In sigma-delta modulators, it is the internal reference of the quantizer.

When the analog-to-digital converter includes a sigma-delta modulator and the DC offset voltage level input to the sigma-delta modulator is input to the preamplifier and the processed low frequency pulse is selected to provide idle mode tones above the audio band, Substantial reduction of nonlinear distortion in the microphone is achieved. The DC offset voltage level is provided by the preamplifier as a differential mode DC signal or by a sampler as described above. The duration of the pulse response of the combination of the microcircuit and the preamplifier is limited by the highpass function of the preamplifier, further reducing the sensitivity to the generation of idle mode tones.

In addition, a microphone is provided comprising an integrated circuit as described above and a condenser microphone element configured to provide a microphone signal in response to sound pressure at the microphone element. The condenser microphone element can be a microphone having an electric layer (ie, electric condenser microphone, ECM) or a DC biased condenser microphone.

Also provided with a microphone amplifier is a microphone including an integrated circuit as described above and a microphone electromechanical system (MEMS) for providing a microphone signal in response to sound pressure at the MEMS microphone element.

The present invention is explained in detail with reference to the following hibernation.

1 illustrates a digital microphone including a microphone element, a preamplifier with a high pass filter function, an anti-aliasing filter and an analog-to-digital converter.

Figure 2 shows a digital microphone including a microphone element, a preamplifier with combined band pass filter function and an analog-to-digital converter.

3 illustrates a microphone including a differential preamplifier and a phase-shifter with a filter function in a first configuration.

4 shows a microphone including a differential preamplifier with a filter function and a phase-shifter in a second configuration.

5 shows a first phase-shifter.

6 shows a second phase-shifter.

7 shows a four-port high pass feedback network providing a low pass filter function inside an amplifier.

8 illustrates a four-port band-blocking feedback network providing band pass filter functionality inside an amplifier.

9 details a preamplifier having a differential mode band pass filter function and a common-mode low pass filter function.

10 illustrates a preamplifier with differential mode band pass filter function and common-mode low pass filter function, and provided with differential DC shift.

Figure 11 shows the first frequency-gain transmission function of the preamplifier.

12 shows the second frequency-gain transmission function of the preamplifier.

Fig. 13 shows a differential preamplifier with a switch-capacitor connected thereto, and a sampler is coupled with a sigma-delta converter.

14 shows a first configuration of the digital microphone.

15 shows a second configuration of the digital microphone.

Figure 16 shows a single end preamplifier and analog-to-digital converter with filter function.

17 shows a schematic diagram of a microphone and microphone element having an integrated circuit.

18 shows a schematic diagram of a microphone and MEMS microphone element having an integrated circuit.

1 illustrates a digital microphone including a microphone element, a preamplifier with a high pass filter function, an anti-aliasing filter and an analog-to-digital converter. Microphone element C _m 105 comprises a first member in the form of a membrane or diaphragm that moves in response to sound pressure acting on the membrane. The membrane moves the microphone case holding the so-called backplate or movable membrane in relation to the second member. One of the members, typically the second member, is connected to ground, while the other member, generally the membrane, is biased through a biasing resistor R _mb connected to the DC bias voltage V _mb . Thus, charge is provided to the film or movable member of the microphone element 105, C _m . Because the amount of charge remains constant (for very low frequencies and higher frequencies), the electric microphone signal is provided by the film as the film moves in response to the sound pressure acting on the film. The microphone signal caused by the movements of the film is superimposed on the DC signal generated by biasing. Circuitry having a microphone element 105, a bias resistor 104 and a DC blocking capacitor is included in the microphone circuit 107.

The microphone signal is provided to the input phi of the preamplifier 101 via a DC blocking capacitor 106 which blocks the DC bias signal from reaching the input of the preamplifier.

The preamplifier 101 is characterized by having a high pass gain characteristic having a relatively low gain for frequencies below the audio band and a relatively high gain for frequencies in the audio band. Preferably, the gain characteristic drops to the first, second, third, fourth, or higher order below the audio band. Preferably, the low pass gain characteristic has a cutoff frequency of about 10 Hz. In addition, the amplifier is characterized by processing the low frequency microphone signal as a common-mode signal and processing the high frequency microphone signal as a differential mode signal. Thus, low frequency components are effectively suppressed. For an audio band with a lower corner frequency of 20 Hz, low frequencies are frequencies below about 50-20 Hz, and high frequencies are frequencies above about 10-30 Hz.

및

에서 차동 신호로서 제공되고, 안티-에일리어싱 필터(AAF, 102)를 통해 시그마-델타 A/D 컨버터(103)로 제공된다. 시그마-델타 컨버터는 D_O라 표시되는 오버 샘플링된 1-비트 출력 신호를 제공한다. 시그마-델타 컨버터는 약 2.4MHz 또는 그 이상의 샘플링 주파수에서 동작한다.The output of preamplifier 101 is output ports

And

Is provided as a differential signal at sigma-delta A / D converter 103 via an anti-aliasing filter (AAF) 102. The sigma-delta converter provides an oversampled 1-bit output signal labeled D _O. Sigma-delta converters operate at sampling frequencies of about 2.4 MHz or higher.

The digital microphone of FIG. 1 may be exposed to sounds occurring in an environment having a relatively large amplitude and a relatively sloped amplitude slope in the time domain. The sounds may result from the closing of the door of the vehicle, dropping the phone on the table, etc., and generating input signals with low frequency signals with large amplitude. When the microphone is connected to a power source, the electronic circuit of the microphone is represented by a stepped power pulse. This is referred to as a start-up or power-up pulse and generates low frequency (input) signals with large amplitudes.

The bias resistance Rmb and capacitance of the microphone element in the analog region form a high pass filter with an impulse response with a relatively short onset time but with a relatively long decay time. As a result, very low frequency pulses (considered almost DC) are held at the input of the preamplifier. If the impulse response does not change, some distortion of the microphone's output signal occurs in the digital domain.

In the analog domain, noise considerations require that the bias resistor Rmb be designed to have an ohmic value as high as a few hundred mecha ohms to giga or tera ohms. The typical capacitance value of a microphone combines with a large bias resistor to produce a low cutoff frequency of about 0.01 Hz. This is consistent with the reduction time of the impulse response by several minutes. Thus, large sound impulses generate a DC potential at the input of the preamplifier that is reduced but maintained for several minutes. In addition, since the preamplifier provides a gain of 0 dB or more, for example 6 dB, the gain will form a high DC level at the output of the amplifier unless it is for the high pass function of the preamplifier.

The high pass filter of the preamplifier provides a low cutoff frequency of about 10Hz. The filter of the amplifier allows the combined impulse response of the microelement and the preamplifier to decrease faster-typically faster than 0.5 seconds. Thus, the input of the sigma-delta converter is imposed with fast decreasing impulses rather than a slowly decreasing DC level.

The sigma-delta converter provides the output signal D _O in the digital domain. The output signal has a power spectrum that includes a flat noise floor. The noise level above the noise corner frequency gradually increases at higher frequencies. The nature of the sigma-delta converter makes the converter sensitive to the DC level of the aforementioned low frequency signals where the idle tones occur in the power spectrum of the digital output signal. The tones provide some distortion to the reproduction of the sound signal in the digital output signal. The idle mode tones should be controlled to provide good sound reproduction in the digital domain.

2 shows a digital microphone including a micro element, a preamplifier with combined band pass filter function, and an analog-to-digital converter. The microphone element 105 is included in the microphone circuit 107 and operates as described above.

Preamplifier 201 is comprised of a band pass filter that provides more than 0 dB of gain in the pass band. The pass band has a lower cutoff frequency of about 10 Hz and an upper cutoff frequency located less than one half of the sampling frequency of the sigma-delta converter 103. For a sampling frequency f _s of about 2.4 MHz, the upper cutoff frequency is 40-70 KHz, for example.

Thus, the preamplifier has a relatively low gain for frequencies below the audio band, a relatively high gain for frequencies within the audio band, and a relatively low gain for frequencies above the audio band. It is characterized by having a gain characteristic. Preferably, the frequency-gain characteristic is expressed in the first, second, third, fourth, or higher order in the range below and above the audible range. In addition, the preamplifier is characterized by processing a low frequency microphone signal as a common-mode signal and processing a high frequency microphone signal as a differential mode signal. Thus, low frequency components are effectively suppressed. Thus, the preamplifier implements a band pass filter. In the following, the preamplifier is designated as a differential preamplifier.

The function of the bandpass filter amplifier suppresses signals in the pass-band, i.e., frequencies below the audio band, to prevent overloading the input stage of the differential amplifier, and in subsequent sampling and digitization of the signal output from the bandpass filter amplifier. It is to suppress frequencies above about 1/2 of the Nyquest frequency to prevent aliasing problems. Thus, individual anti-aliasing filters can be avoided.

및

과 함께 도시된다. 전치 증폭기는 마이크로폰 회로(107)에 접속되어 입력 터미널 φ에서 마이크로폰 신호를 수신한다.FIG. 3 shows a microphone including a differential preamplifier with a filter function and a phase-shifter in a first configuration. Differential preamplifier 306 has input terminal φ and output terminals

And

Is shown with. The preamplifier is connected to the microphone circuit 107 to receive the microphone signal at the input terminal φ.

및 연산 증폭기(301)의 반전 입력 사이에 접속된다. 임피던스 Z2*(304)는 연산 증폭기(302)를 위한 피드백으로 접속된다. 임피 던스 Z1(305)는 각각 증폭기(301, 302)의 반전 및 비반전 입력 사이에 접속된다. 필터 피드백 네트워크는 차동 모드에서 차동 전치 증폭기의 고역 통과 필터 기능 및 대역 통과 필터 기능을 형성한다. 임피던스들(Z1, Z2, Z2*)은 용량성 또는 저항성 동작 또는 이들의 조합을 제공하는 칩 구현에 사용할 수 있는 구성요소들에 의해 구현된다.The preamplifier consists of a mechanical amplifier having a filter feedback network comprising impedances Z1, Z2, Z2 * 305, 303, 304 and a phase-shifter network 307. Impedance Z2 (303) is an output terminal

And an inverting input of the operational amplifier 301. Impedance Z2 * 304 is connected with feedback for the operational amplifier 302. Impedance Z1 305 is connected between the inverting and non-inverting inputs of amplifiers 301 and 302, respectively. The filter feedback network forms the high pass filter and band pass filter of the differential preamplifier in differential mode. Impedances Z1, Z2, Z2 * are implemented by components that can be used in a chip implementation to provide capacitive or resistive operation or a combination thereof.

Phase-shifter network PD (f) 307 has an input port designated by (a) and an output port designated by (b). The input port a is connected to the inverting input of the first operational amplifier 301 and the output port b is connected to the inverting input of the second operational amplifier 302. The phase-shifter network provides a frequency that is determined by a phase shift that shifts the phase by about ± 180 degrees at high frequencies and about 0 degrees at low frequencies. This ensures that the differential mode output signal operates as a common mode signal at low frequencies while operating as a real differential mode signal at high frequencies. The phase shift provides efficient high pass filtering because the output signal can be a common mode signal only at low frequencies.

This configuration of the preamplifier is advantageous when the input of the phase-shifter is connected to the input of the preamplifier through the inverting input of the operational amplifier 301.

However, in an optional configuration, the input of the phase-shifter is connected to a circuit node formed between the microphone block 107 and the input of the preamplifier (at the noninverting input of the operational amplifier 301).

It should be noted that the feedback filters may be implemented by active filters and / EH by an active DC servo device.

4 shows a microphone including a differential preamplifier with a filter function and a phase-shifter in a second configuration. The configuration is almost identical to that shown in FIG. 3, but the phase-shifter 307 is connected between the output terminal of the first operational amplifier 301 and the non-inverting input terminal of the second operational amplifier.

Thus, a phase-shifter 307 as shown in Figures 3 and 4, as described above, is cross coupled between the two common mode amplifiers of the differential preamplifier.

Amplifiers 301 and 302 are generally designated as amplifier sections because they may have several amplifier stages.

5 shows a first phase-shifter. Phase shifter 307 is implemented in combination with an operational amplifier to form a first order low pass filter that provides a phase shift of about 0 degrees at low frequencies and about 180 degrees at high frequencies.

및

이 낮은 주파수에서 공통 모드로 동작하고 높은 주파수에서 차동 모드로 동작하도록 한다. 따라서, 위상-쉬프터를 가지는 전치 증폭기는 낮은 주파수의 효율적인 차동 모드 억압을 제공한다. 바람직하게, 상기 차동 모드 고역 통과 필터의 컷오프 주파수는 약 10Hz에 위치하지만, 약 30 또는 50Hz 까지의 범위에 위치할 수 있다.Phase-shifter 307 is the outputs of the preamplifier

And

It operates in common mode at this low frequency and in differential mode at high frequencies. Thus, preamplifiers with phase-shifters provide low frequency efficient differential mode suppression. Preferably, the cutoff frequency of the differential mode high pass filter is located at about 10 Hz, but may be in the range up to about 30 or 50 Hz.

6 shows a second phase-shifter. In this embodiment, phase-shifter 401 is combined with an operational amplifier to form a band pass filter that provides a phase shift of about 0 degrees at low frequencies, about 180 degrees at intermediate frequencies, and about 0 degrees at higher frequencies. Is implemented. Intermediate frequencies are defined as frequencies that include frequencies up to the upper cutoff frequency of the audio band and the band pass filter. The higher cutoff frequency is generally designed to be provided by an operational amplifier and located around the pole. Frequencies above the upper cutoff frequency are indicated by higher frequencies.

및

이 낮은 주파수들 및 더 높은 주파수들에서 공통 모드로 동작하도록 한다. 중간 주파수들에서, 위상-쉬프터는 전치 증폭기가 차동 모드로 동작하도록 한다. 따라서, 위상 쉬프터를 가지는 전치 증폭기는 낮은 주파수들 및 높은 주파수들에서 효율적인 차동 모드 억압을 제공한다. 따라서, 효율적인 안티-에일리어싱 필터가 추가로 제공된다.Phase-shifter 401 outputs the preamplifiers

And

It operates in common mode at these lower and higher frequencies. At intermediate frequencies, the phase-shifter allows the preamplifier to operate in differential mode. Thus, a preamplifier with phase shifter provides efficient differential mode suppression at low and high frequencies. Thus, an efficient anti-aliasing filter is further provided.

The phase-shifter includes a capacitor 603 and a resistor 601 connected in parallel, forming a signal path between the input port (a) and the output port (b). The series connection of capacitor 602 and resistor 604 provides a signal path between output port b and ground.

The phase-shifter PD (f) ensures that the differential mode gain is very low at low and high frequencies, but not at intermediate frequencies, regardless of whether it is implemented as a primary, secondary or higher order phase-shifter. The phase-shifter provides very low gain by providing at least one zero and pole at OHz (DC) at frequencies above the audio band in the transmission function of the preamplifier filter.

7 shows a four-port high pass feedback network providing a low pass filter function to an amplifier. The feedback network includes ports (a), (b), (c) and (d). Feedback paths from (b) to (a) and from (d) to (c) are formed by capacitors C2 701 and C2 * 704, respectively.

The series connection of the capacitor and the resistor between the ports (a) and (c) is formed by C1 703 and R1 702. The series connection connects the inputs of the amplifier sections 301, 302.

8 illustrates a four-port band cutoff feedback network providing bandpass filter functionality to an amplifier. The feedback network includes ports (a), (b), (c) and (d). Feedback paths from (b) to (a) and from (d) to (c) are formed by a capacitor 802 connected in parallel to the resistor 801 and a capacitor 805 connected in parallel to the resistor 806.

The series connection of the capacitor and the resistor between the ports (a) and (c) is formed by C1 804 and R1 803. The series connection connects the inputs of the amplifier sections 301, 302.

However, it should be noted that other feedback filter configurations, including active filters and / or higher order filters, may be provided.

9 details a preamplifier having a differential mode band pass filter function and a common mode low pass filter function. Preamplifier 901 includes an AC feedback network consistent with that shown in FIG. 8 and a phase-shifter network consistent with that shown in FIG. 5 in a configuration such as that shown in FIG. The feedback network and phase-shifter form a differential mode highpass filter. In combination with this, the DC feedback network provides a common mode low pass filter function, which provides a clear common mode gain for DC and low frequencies. The preamplifier is configured such that the impedance of the DC feedback network prevails at DC (low frequencies), while the impedance of the AC feedback network prevails at high frequencies. Thus, the DC level at which the differential signal is provided can be controlled using very limited circuitry, and the desired AC filter function is realized.

The reference DC voltage level Vb is applied to the input of the preamplifier, and the common DC gain by the common mode forms the clear DC level at which the differential signal is provided.

The input phi of the preamplifier 901 is connected to the microphone circuit 107, and the controlled DC voltage level is applied to the input through the resistor Rb 902 by a voltage source (not shown). The DC voltage level is set according to the desired common mode DC output level and common mode DC gain of the preamplifier. To this end, the power supply to the microphone is typically between a single end, the nominal supply voltage level V _dd and ground. To provide a symmetrical limitation of the AC output swinging in differential mode, the common mode DC output level should be close to half of the nominal supply voltage. Thus, maximum differential mode AC voltage swing can be achieved. It is emphasized that the above limitation is the most restrictive to obtain a large AC gain. Also, since the noise in the digital microphone signal decreases as the preamplifier gain increases, a large preamplifier gain (and, in turn, a room for a large AC voltage swing) is desirable. Larger AC signal swings improve the signal-to-noise ratio because the noise from the sigma-delta converter is constant with respect to different amplitudes.

In general, the output DC level should be achieved by a DC preamplifier gain of 0 dB or more because the input stage of the preamplifier is saturated when exposed to relatively large input DC levels compared to the supply voltage.

If, for example, the nominal power supply voltage level is 1.5V, the common mode DC output level is 1/2 of 1.5V, or 0.75V. For input stages in the form of 1.5V supply voltage and PMOS differential pairs, the inputs must be able to handle input DC voltage levels up to about 0.4V. In order to set the DC level at the output, the DC gain should be at least about twice. A DC output level of 0.75V and a double DC gain are less than 0.4V and require a DC reference voltage Vb equal to 0.375V.

The preamplifier is implemented as a differential preamplifier formed from two operational amplifiers 903 and 904.

The DC gain of the differential preamplifier is realized by a DC feedback network located around each of the two operational amplifiers. The DC feedback network uses a voltage divider to sense the individual output signals. The voltage divider is implemented using resistors 906 and 907 for the operational amplifier 903 and resistors 909 and 908 for the operational amplifier 904.

The AC feedback network is configured as shown in FIG. However, because it is coupled with the DC network, the resistive feedback path of the AC network includes resistors 910, 906 and resistors 909, 913 for the amplifiers 903, 904, respectively. Capacitors 905 and 914 are connected in parallel with the resistive feedback paths. Capacitor 911 and resistor 912 are connected in series between the inverting inputs of amplifiers 903 and 904.

The phase-shifter is configured as shown in FIG. However, because it is coupled with the DC network, it receives an input from the circuit node formed by the voltage dividers 906 and 907.

At the DC input, the impedance of the DC feedback network is dominant. Thus, looking at the DC equivalent diagram, the voltage provided by the voltage dividers 906 and 907 is approximately equal to the voltage fed back to the inverting input of the amplifier 903. Thus, the DC gain of the amplifier 903 is determined by the voltage divider. For example, twice the DC gain is achieved by two 100 kiloohm resistors. Similarly, the DC gain of amplifier 904 is determined by resistors 909 and 908. The DC gain is selected to match the gain of amplifier 903.

Also, in order for the amplifier 904 to provide an output voltage at the same level as the amplifier 903, the resistor 912 of the phase-shifter is connected to a circuit node formed by the voltage dividers 906 and 907. Thus, the output level of the amplifier 904 follows the output level of the amplifier 903.

As an alternative embodiment, a DC offset or DC bias can be provided to the input stage of the amplifier 903 by, for example, shifting the source voltage level (and gate and drain voltage) of the differential pair PMOS devices at the input stage. The embodiment may be based on DC and AC feedback networks similar to those described above. In that case, V _b is connected to ground to set the DC bias level. It should be noted that a combination of the two configurations may be possible.

In AC, the impedance of the AC feedback network is dominant. Thus, looking at the AC equivalent diagram, the feedback filter and phase-shifter operate as described in connection with Figures 4, 5, and 8. However, a phase-shifter network comprising a resistor 512 and a capacitor 513 is configured to provide a gradual shifting phase of the signal input to the phase-shifter. This gives the preamplifier a differential mode signal at higher frequencies including the common mode output signal at lower frequencies including DC and frequencies above the audio and audio bands.

The phase shift between one end of the differential amplifier configured around the operational amplifier 903 and the other end configured around the operational amplifier 904 is implemented by a capacitor 916 and a resistor 915. Thus, phase shift is obtained by the phase shifter. Resistor 915 in series with capacitor 916 forms poles F _P2 and zero F _Z1 described below.

10 shows a preamplifier with a differential mode band pass filter function and a common mode low pass filter function, with differential DC shift provided. In the configuration of the preamplifier described in conjunction with FIG. 9, the common mode DC output is provided by a DC voltage reference provided at the input terminal of the preamplifier or at the input stage of one or both of the operational amplifiers 903, 904. In addition, the common mode DC output is determined by the DC voltage reference and the common mode DC gain of the preamplifier.

However, the preamplifier 1001 shown in FIG. 10 provides two common mode DC shifts using the active current sources 1001, 1002. In addition, differential mode DC shift is provided by the difference in currents provided by the two current sources. Since the sigma-delta converter is connected to detect the differential signal provided by the preamplifier 100, the idle mode tones can be controlled by the differential-mode DC shift provided by the preamplifier.

When provided by active current sources 1001 and 1002 regardless of whether the DC shift is a differential mode DC shift, a common mode DC shift, or both, the input terminal is for example a resistor 902. Must be set to the reference level by connecting the input to ground (ie V _b = 0 V, but in practical implementation V _b must be at least 100 mV). However, applying another DC reference in combination with active current sources is optional.

)에서 원하는 DC 전압을 제공할 것이다.For example, if the supply voltage V _dd is single ended to 1.5V and ground V _b is set to 0V, the common mode DC voltage with ½V _dd = 0.75V causes the resistor 906 to form a DC voltage of 0.75V around. It can be achieved by applying a DC current through. If resistor 906 has 100 kiloohms, a current of 7.5 mA is output (output terminal

Will provide the desired DC voltage.

)에서 DC 전압을 제공한다. 따라서, 요구되는 것과 같이 차동 모드 DC 출력은 15mV로 형성된다.In addition, if desired, with respect to the use of idle mode tones, to form a resistor 909 having a value substantially equal to the differential mode DC shift of 15 mV and the counter part 906, a current of 7.35 mA is applied. 0.735V Amplifier 904 Output (Output Terminals)

To provide the DC voltage. Thus, as required, the differential mode DC output is formed at 15mV.

In terms of AC, the phase-shifter, PD (f), is formed using a resistor 1003 and a capacitor 1004 as shown in FIG. 5 and inserted inside a preamplifier as shown in FIG. Thus, resistor 1003 is connected to the inverting terminal of operational amplifier 903 and the non-inverting input of operational amplifier 904. Capacitor 1004 is connected between the non-inverting input of amplifier 904 and ground.

Thus, the preamplifier is connected to provide a common mode DC level for maximization of the output AC voltage swing and a differential mode DC level for control of idle mode tones, where low frequency output signals are generated as common mode signals and At higher frequencies, the output signal occurs as a differential mode signal.

및 접지 사이에 접속된 전압 분배기에 의해 구현될 수 있다. 전압 분배기는 저항(1003)이 증폭기(903)의 반전 입력에 접속되는 대신에 접속되는 회로 노드에서 분배된 출력 전압을 제공한다.In an alternative embodiment, the differential mode DC shift is an output terminal.

And a voltage divider connected between ground. The voltage divider provides the output voltage distributed at the circuit node to which resistor 1003 is connected instead of being connected to the inverting input of amplifier 903.

Figure 11 shows the first frequency gain transmission function of the preamplifier. The frequency gain is shown as the preamplifier operates in common mode (A _CM = curve 1) and differential mode (A _DM = curve 2). The response is shown as a linear approximation on the logarithmic frequency axis and the logarithmic gain axis.

Curve 2 illustrates common mode operation. From DC to the position of pole F _{PI '} a flat response is provided. The flat response is typically provided with a gain of about 0 dB, but the amount of gain is determined by the embodiment of the selected DC offset and the desired DC level. The response proceeds to a low gain level beyond F _{P1 '} to F _Z1' . The low level is flat above F _{Z1 '} . The common mode response is provided by the phase-shifter shown in FIG. 5 when operating with a preamplifier.

Curve 1 illustrates the differential mode of operation. At least one pole in DC forms a positive slope of transmission function that continues up to the position of pole F _P1 . Thus, differential mode DC signals are effectively suppressed. From pole F _P1 to pole F _P2 , a flat response is provided. Preferably, the audio band is included in the frequency range of the flat response. _Above F _P2 and up to F _Z1 , the response proceeds to low gain stability. The purpose of gain stabilization is to suppress noise sources beyond the audio band, such as sound noise, electrical noise, and to reduce the gain effects of the resonance peaks of the microphone element. The gain stability level is determined so that noise sources and gain effects do not limit the output swing (gain) of the preamplifier. At pole F _P3 , the gain function starts to fall for higher frequencies. Pole F _P3 may be designed to be positioned around a pole provided by such operational amplifiers, or pole F _P3 may be provided by such operational amplifiers.

Series resistor C1 702; 804 in the feedback network provides pole zero pairs F _P2 , F _Z2 . The pole zero pairs are generally located at about 50-60 KHz and form gain stability from F _Z2 to F _P3 located around about 500 KHz, and the amplifier itself provides at least one pole, thus negative slope. .

The audio band is shown using the lower corner frequency F _AL and the upper corner frequency F _AU .

12 shows the second frequency-gain transmission function of the preamplifier. In other words, the frequency-gain is shown for the preamplifier to operate in common mode (A _CM = curve 1) and differential mode (A _DM = curve 2).

Curve 2 illustrates common mode operation. The common mode transmission function operates similarly to that shown in FIG. 11, but frequencies F _{Z2 ′} exceeding transmission function 0 are provided. The positive slope begins above _{FZ2 '} . This causes the preamplifier to start operating as a common mode amplifier for frequencies above F _{Z2 '} . Thus, the suppression of signal components in the upper cutoff band of the differential mode bandpass filter is further suppressed by a dominant common mode operation. The common mode response is provided by the phase shift shown in FIG. 6 when operating in conjunction with a preamplifier.

Curve 1 illustrates the differential mode of operation. The transmission function illustrates the actual bandpass filter function without gain stabilization provided by the series resistor R1 702; 803 in the feedback network of the preamplifier.

및

로 제공한다. 상기 출력들에 의해 제공되는 전치 증폭기로부터의 신호는 시그마-델타 컨버터(103)가 결합된 스위치-커패시터 검출기에 의해 서로 상이하게 샘플링된다. 스위치-커패시터 검출기는 선형 증폭기(1301) 주변에 형성된다. 차동 샘플링은 스위치들(S1-S4)에 의해 2개의 회로 구성들 사이에 접속된 입력 직렬 커패시터(1305) 및 피드백 커패시터(1304)에 이해 구현된다.FIG. 13 shows a differential preamplifier with a switch-capacitor sampler coupled to a sigma-delta converter. The differential preamplifier 201 receives a microphone signal at its input φ to output dual outputs.

And

To provide. The signals from the preamplifiers provided by the outputs are sampled differently from each other by a switch-capacitor detector to which the sigma-delta converter 103 is coupled. The switch-capacitor detector is formed around the linear amplifier 1301. Differential sampling is implemented in the input series capacitor 1305 and the feedback capacitor 1304 connected between the two circuit configurations by the switches S1-S4.

및

로 접속된다. 피드백 커패시터(1304)는 스위치 S3에 의해 피드백 경로로서 접속된다. 스위치 S4는 커패키서(1304) 및 스위치 S3의 직렬 접속과 병렬 접속된다.Input series capacitor 1305 is the outputs of the preamplifier by separate switches S1 and S2 at the input side.

And

Is connected. The feedback capacitor 1304 is connected as a feedback path by the switch S3. The switch S4 is connected in parallel with the series connection of the capacitor 1304 and the switch S3.

The switches S1-S4 can be controlled to open and close according to the manner shown in the lower right corner, i.e., the switches S1 and S3 work together, the switches S2 and S4 work together, but 180 The phase of the figure is shifted. The switches S1-S4 are controlled using the same clock frequency as the sampling frequency of the sigma-delta converter. Switch-capacitor sampling of differential signals is known to those skilled in the art and, although not described in detail, is shown to illustrate the interconnection of a differential preamplifier and a sigma-delta converter.

It should be noted that the amplifier 301 is connected by a capacitor 1303 to implement a summing amplifier of the sigma-delta feedback loop. Those skilled in the art generally know how sigma-delta modulators are configured. Those skilled in the art will appreciate that the summing amplifier compares the input signal with the feedback signal obtained from the quantizer providing the digital output signal D _O. The output of the summing amplifier is connected to an integrator (out of order) that provides an output signal to the quantizer. The feedback signal is provided to the summing amplifier 1301 by the capacitor 1303.

In addition to the switched capacitor sampling of the differential signal, one embodiment of DC shift is implemented. This embodiment of DC shift is a choice for the differential DC shift provided in the preamplifier and is configured to control the idle mode tones of the sigma-delta converter.

DC shift can be implemented as a single ended DC shift in the stage of the digital microphone. This can be implemented by sampling the DC voltage reference V _{DCoffset ? Δ} using a series capacitor 1306 connected alternately to the DC voltage reference and ground. The capacitor is alternately connected by switches S5, S6. The switching scheme of the switches S5, S6 is controlled by a logic network connected to the output of the quantizer of the sigma-delta converter.

In the above description, the anti-aliasing filter may be implemented by a higher cutoff frequency implemented by the band-pass filter of the preamplifier. It is desired to remove spectral components above the sampling frequency of the converter.

Thus, the summing amplifier may be provided with an integrated error feedback signal of the sigma-delta modulator through the first series capacitor, and a DC voltage level may be provided to the summing amplifier through the second series capacitor. Thus, idle mode tones can be controlled by the ratio between the values of the first and second series capacitors. The position of the idle mode tone is determined by the equation:

F _idle = (V _{DCoffset ∑ △} / V _{REF ∑ △} ) * (C ₁₃₀₆ / C ₁₃₀₄ ) * ½F _S

F _idle is the position of the idle mode tone, C ₁₃₀₆ and C ₁₃₀₄ are the values of wp 1 and the second capacitor, F _S is the sampling frequency, V DC _{offset ∑ △} is the sampled DC voltage, and V _{REF ∑ △} is In sigma-delta modulators, it is the internal reference of the quantizer.

When the analog-to-digital converter includes a sigma-delta modulator and the DC offset voltage level input to the sigma-delta modulator is input to the preamplifier and the processed low frequency pulse is selected to provide idle mode tones above the audio band, Substantial reduction of nonlinear distortion in the microphone is achieved. The DC offset voltage level is provided by the preamplifier as a differential mode DC signal or by a sampler as described above. The duration of the pulse response of the combination of the microcircuit and the preamplifier is limited by the highpass function of the preamplifier, further reducing the sensitivity to the generation of idle mode tones.

14 shows a first configuration of the digital microphone. For condenser microphone implementation, the digital microphone is sealed by a capsule 1401 surrounding the integrated circuit in the form of a chip 1402. Chip 1402 includes terminals Tc1, Tc2, Tc3, Tc4 for connecting the microphone element 1408, bias voltage, ground potential and power supply voltage, respectively. Terminal Tc6 provides the digital microphone output signal D _O from the A / D converter. Through terminal Tc5, the clock signal is provided to the A / D converter. The power supply voltage to the amplifier 1405 and the A / D converter can be provided via terminal Tc6 in the case where terminal Tc4 can be omitted.

For the condenser microphone capsule implementation, the microphone element 1408 becomes a condenser microphone that requires a DC bias supply to provide adequate charge to one of the microphone members. The DC bias is provided through a resistor 1403. DC blocking capacitor 1404 prevents the DC bias level from reaching the input stage of preamplifier 905. In an alternative embodiment, the microphone 1408 is an electric condenser microphone (ECM). Thus, the microphone element 1408 is directly connected to the input and bias resistor of the preamplifier 1405 and no DC blocking capacitor is required.

For micro electromechanical system (MEMS) implementation, the digital microphone is implemented with a MEMS device that includes a microphone electrical circuit portion and a micro mechanical portion that implements the microphone element 1408. For circuit perspective, the microphone element changes the circuit position with the DC blocking capacitor. The micro electrical circuit portion or chip unit comprises a preamplifier 1405 and an A / D converter 1407 with EK band pass filter function.

15 shows a second configuration of the digital microphone. Note that the following description applies to the implementation of a digital microphone by a chip and a condenser microphone. However, based on the description provided in connection with FIG. 14, those skilled in the art can perform a MEMS implementation.

Digital microphone 1501 includes an integrated circuit 1502 having a DC voltage regulator 1503 providing a regulated voltage to amplifier 1509 and sigma-delta converter 1011. The microphone bias voltage is provided by an on-chip voltage up converter 1504 that receives an off-chip oscillation signal having a voltage magnitude, and in response to the oscillation signal, the up converter provides an output oscillation signal having a large voltage magnitude. The output signal is low pass filtered by low pass filter 1505 and is provided to microphone element 1508 by series resistor 1506. Capacitor 1507 blocks the DC bias voltage from reaching the input of preamplifier 1509 with the transmission function described above. The output of preamplifier 1509 is provided by sigma-delta converter 1511.

The voltage up converter or voltage pump UPC 1004 may be in the form of a so-called Dickson-converter. The voltage pump is preferably operated by an oscillator that provides a square wave oscillator signal to the voltage pump. For example, other signals with low harmonic components, such as sine waves or filtered square waves, can be used to obtain lower noise. In an alternative embodiment, the oscillator is inserted into chip 1502.

It is shown that the up converter and the sigma-delta converter share the same oscillator / clock signal as provided through terminal Tc4. It should be noted that the signal must be distributed to obtain different oscillation / clock signal frequencies in the UPC and sigma-delta converters.

16 shows a single ended preamplifier and analog-to-digital converter. In this embodiment, the microphone circuit 107 provides the signal to a single ended preamplifier 1602. The output of preamplifier 1602 is provided to an analog-to-digital converter 103 in the form of a sigma-delta modulator.

Preamplifier 1602 includes an amplifier section 160. Advantageously, said amplifier section is an operational amplifier having a differential input. The amplifier section receives a signal from the microphone circuit 107 at a non-inverting input (+), the feedback filter 1603 receives an output signal from the amplifier section 1601, and the inverting input (−) of the amplifier section 1601. ) Provides a feedback signal.

The frequency-gain characteristic of the feedback filter 1603 has a low pass characteristic that in turn implements the high pass filter characteristic of the preamplifier using the feedback filter. Preferably, the pass band (low frequency) of the feedback filter provides a substantially flat gain response to DC and a flat response to higher frequencies for higher frequencies above the gain transition band. The above is explained by the feedback filter.

However, the frequency gain transfer function can be configured to match the transfer function of the differential amplifier with the necessary changes.

17 shows a schematic diagram of a microphone having a microphone element and an integrated circuit. The microphone is shown as a cartridge having a microphone member that includes a microphone film and an integrated circuit.

18 shows a schematic diagram of a microphone having an integrated circuit and a MEMS microphone element. The microphone 1902 includes a MEMS microphone member 1903 coupled to the first substrate and a preamplifier circuit 1901 coupled to the second substrate. The preamplifier circuit comprises one of the other embodiments described above, i.e. a preamplifier comprising a feedback circuit and for example a voltage pump and / or a feedback circuit, the preamplifier being a differential amplifier or a single end amplifier. .

It should be noted that the MEMS microphone member 1903 and microphone preamplifier 1901 may be integrated into a single semiconductor substrate.

In general, it should be noted that the preamplifier may be implemented as another type of amplifier such as a single end amplifier, a differential amplifier or a differential differential amplifier. In the case where a preamplifier with several inputs and outputs is used, different implementations of the multiple feedback filter paths may perform the desired frequency transfer function.

The condenser microphone consists of a very light diaphragm and a back plate to which a polarity voltage is applied. Thus, a constant charge is provided for the frequencies involved. The principle of operation is that the sound waves impinging on the diaphragm cause the capacitance between the diaphragm and the backplate to change resonance. This in turn induces an AC voltage on the backplane.

The electric condenser microphone ECM operates in a similar manner except that it has a permanent charge voltage embedded in the electrical material to provide a polarity voltage. The microphone can be implemented in three ways, most commonly when the diaphragm is an electrical material, in which case one side is a metallic material. This is known in the form of flakes or diaphragms. The electrical material does not optimally form the diaphragm, and when higher performance is required, the diaphragm is formed of another material and the electrical material is applied to the back plate. This is known as the back type. The most recent variant is called the front type. Here, the electrical material is applied to the metal diaphragm connected inside the front cover of the microphone and to the input of the preamplifier.

The audio band may be defined as any band within the general definition of the audio band. The general definition can be 20 Hz to 20 KHz. Exemplary lower cutoff frequencies for the audio band can be 20 Hz, 50 Hz, 80 Hz, 100 Hz, 150 Hz, 200 Hz, 250 Hz. Exemplary upper corner frequencies may be 3KHz, 5KHz, 8KHz, 10KHz, 18KHz, 20KHz. Substantially flat gain response variations mean that they are within about ± 1 dB, ± 3 dB, ± 4 dB, and ± 6 dB. However, other variation values can be used to define the term "substantially flat."

Earlier differential preamp configurations have been disclosed. The configurations include different input / output terminal configurations, such as a two terminal configuration. However, it should be noted that three, four, or more terminals may be provided for input / output of signals to the microphone and preamplifier. In particular, it should be noted that individual terminals may be provided for the supply voltage (at the first terminal) and the preamplifier output (at the second terminal). In the case of a differential preamplifier output, two terminals for the output signals can be provided in addition to the terminal of the power supply. A separate terminal is provided for grounding. The ground is shared by the power and output signals, which is common but not always.

Claims (20)

An integrated circuit configured to process microphone signals, First input φ and second input φ * and output (

Amplifier section 301 and the output (

;

,

); And A preamplifier with a feedback filter network (Z1; Z1, Z1 *, Z2) connected between said second input ([phi] ') and an analog connected to receive an anti-aliased filtered input signal and provide a digital output signal. Includes a digital converter, The first input φ to the amplifier section 301 has an input impedance, the input impedance being substantially separated from the feedback network with respect to the input impedance by the input impedance of the amplifier section, A preamplifier is an integrated circuit having a frequency-gain transmission function that suppresses low frequencies. The method of claim 1, The preamplifier 306 is connected to the differential output signal by the first and second amplifier sections 301 and 302.

,

), The preamplifier 306 has a differential mode transmission function including a band pass characteristic (A _DM ), The preamplifier 306 forms filter feedback paths (ab; cd) that connect outputs to the individual inverting inputs of the amplifier section 301, 302, and filter interconnection paths that interconnect the inverting inputs. and an feedback filter network (303, 304, 305) forming (ac). 3. Integrated circuit according to claim 1 or 2, characterized in that the lower cutoff frequency (F _P1 ) of the filter implemented by the preamplifier (306) is located below the lower corner frequency of the audio band. The method according to any one of claims 1 to 3, wherein the pre-amplifier 306 is the analog to the upper position so that the cut-off frequency lower than half the sampling frequency (F _S) of the digital converter (F _P3; F _And a differential mode transmission function (A _DM ) comprising a band pass characteristic with _P2 ). 5. The preamplifier 306 according to any one of claims 1 to 4, wherein the preamplifier 306 comprises a band pass characteristic having a nominal pass band (F _P1 -F _P2 ) and a gain stable band (F _Z2 -F _P3 ). Has a differential mode transmission function (A _DM ), wherein the nominal pass band extends to audio band frequencies, and the gain stable band extends beyond the audio band to frequencies up to an upper cutoff frequency F _P3 . Integrated circuit, characterized in that. 6. The integrated circuit of claim 1, wherein the preamplifier (306) has a common mode transfer function (A _CM ) that includes low pass characteristics. 7. The preamplifier according to any one of claims 1 to 6, wherein the preamplifier has a common mode transmission function (A _CM ) comprising a cut-band characteristic (F _{Z1 '} -; F _Z1' -F _{Z2 '} ), The flat gain response is provided for low frequencies (DC-F _P1 ). 8. The preamplifier of claim 1, wherein the preamplifier comprises a common mode transfer function (A _CM ) and a differential mode transfer function (A _DM ), the frequencies of which the common mode gain (A _CM ) is low. Dominant at (DC-F _{P1 ′} ), while the differential mode gain (A _DM ) is configured to prevail at audio band frequencies (F _AL -F _AU ). 9. The method of claim 1, wherein the common mode gain A _CM is additionally dominant at frequencies above the upper cutoff frequencies F _P2 , F _P3 of the bandpass characteristic. Integrated circuit. 10. The phase shifter 307 is cross-connected between the output of the first amplifier section 301 and the input of the second amplifier section 302. integrated circuit. 11. Integrated circuit according to any one of the preceding claims, characterized in that the phase-shifter (307) is connected between the individual inputs (-) of the individual amplifier sections (301, 302). 12. The circuit of claim 1, wherein the preamplifier is a DC offset circuit coupled with the feedback filter (Z1; Z1, Z1 *, Z2) to provide a DC shift at the output of the preamplifier. 907,908; 1001,1002. 13. The apparatus of any of claims 1 to 12, wherein the feedback filter is coupled and includes DC offset circuits (907, 908; 1001, 1002) configured to provide differential mode DC shift at the output of the preamplifier. Integrated circuit. 14. An integrated circuit as claimed in any preceding claim, wherein said analog-to-digital converter comprises a sigma-delta modulator (1032). 15. The differential signal of claim 14, wherein the sigma-delta modulator 1032 is a differential signal provided by the preamplifier 201 to provide a single-ended input signal for sigma-delta A / D conversion.

,

And a switch-capacitor sampler (1307) for sampling the DC voltage level (V _{Ref ΣΔ} ) so that the single-ended input signal overlaps the sampled DC voltage level. 16. The integrated circuit of claim 15, wherein the sampler comprises a summing amplifier (1301) that is a combined portion of the sampler (1307) and the sigma-delta modulator loop. 17. The summing amplifier 1301 is provided with a coupling error feedback signal of the sigma-delta modulator via a first series capacitor 1303, and the DC voltage level is passed through a second series capacitor 1306. And an integrated circuit (1301). 18. The apparatus of claim 1, wherein the analog-to-digital converter comprises a sigma-delta modulator, the DC offset voltage level input to the sigma-delta modulator is input to a preamplifier 201. And wherein the low frequency pulse being processed is selected to provide idle mode tones above the audio band. 19. A condenser microphone element 105 configured to provide a microphone signal to an input [phi] of microphone preamplifier 201 in response to sound pressure at the microphone element and integrated circuit as described in any of claims 1-18. Microphones (108; 202). 20. A microphone comprising an integrated circuit as described in any one of claims 1 to 19 and a MEMS microphone element configured to provide a microphone signal to the microphone preamplifier in response to sound pressure at the MEMS microphone element.

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