KR20040076837A - Demultiplexer - Google Patents

Abstract

PURPOSE: A branching filter for reducing size and executing out-of-band-pass attenuation is provided to reduce the size and enhance the space efficiency by using a phase shifter including a lumped element. CONSTITUTION: A branching filter includes a common terminal(C), a plurality of surface acoustic wave filters(BF1,BF2), and a phase shifter(6). Each of the surface acoustic wave filter is formed with a surface acoustic wave element and includes signal terminals(t1,t2,t3,t4) connected to the common terminal. In addition, each of the surface acoustic wave filter has different bandwidths. The phase shifter is formed by a lumped element between the surface acoustic wave filters and the common terminal.

Description

Splitter {DEMULTIPLEXER}

The present invention relates to a splitter using a plurality of surface acoustic wave filters composed of a plurality of surface acoustic wave elements formed on a piezoelectric plate.

Background Art A surface acoustic wave filter is known in which a number of surface acoustic wave elements are formed on a piezoelectric substrate as a band pass filter for a high frequency in a mobile communication device.

In such a surface acoustic wave filter, a ladder arm is formed between an input terminal and an output terminal, a plurality of parallel arms are formed between the serial arm and the reference potential terminal, and a surface acoustic wave resonator is appropriately disposed between the serial arm and the parallel arm. Surface acoustic wave filters constituting a circuit are known. In addition, a technique for configuring a splitter using a plurality of ladder surface acoustic wave filters having different passband frequencies is known.

As a conventional technique, what is described, for example in Unexamined-Japanese-Patent No. 3246906 is known.

23 is an equivalent circuit diagram showing a conventional splitter, and FIG. 24 is a graph showing frequency characteristics in the vicinity of a pass band in the splitter of FIG. FIG. 25 is a graph showing frequency characteristics in the high frequency region of the splitter of FIG.

As shown in Fig. 23, Japanese Patent No. 3246906 discloses the first end of the first surface acoustic wave filter BF1 on the common terminal C side, and the first end of the second type surface acoustic wave filter BF2 on the common terminal C side. A technique of using a delay line 9 between a common terminal C and a second surface acoustic wave filter BF2 is described. This publication also discloses a technique using an inductance element in place of the phase path 9.

According to such a technique, as shown in FIG. 24, the splitter can be configured simply without significantly deteriorating the original characteristics of the filter.

In recent years, the demand for miniaturization and thinning of the splitter itself and the improvement of the characteristics of the high frequency range are increasing.

Particularly, in the splitter using less than 1 GHz, miniaturization is difficult due to the limitations in the dimensions and the configuration of the surface acoustic wave device.

In addition, the delay line has a wavelength shortening effect due to the dielectric constant of the dielectric substrate used. For example, in the 800 MHz system, for example, the delay line length using a glass epoxy substrate having a dielectric constant of about 4 is about 40 mm and the dielectric constant is 7 In the case of using a ceramic substrate of about 35mm, it is also necessary to store in a volume smaller than 5mm x 5mm while paying attention to crosstalk and phase shortening between delay lines.

Therefore, when the splitter is configured using such a delay line, the shape of the splitter is determined by the configuration dimension of the delay line, and the above-mentioned requirements for compactness and thinning cannot be satisfied.

In recent years, the direct conversion of the RF circuit portion of mobile communication terminals has increased the demand for attenuation of twice, three, and four times the high frequency of the reception filter. However, in the ideal phase of the delayed line, which is a distribution integer line, since it is a simple distribution integer line, its effect as a filter is extremely low and does not contribute effectively to the improvement of the attenuation in the high frequency region.

Therefore, an object of the present invention is to provide a splitter capable of achieving miniaturization and obtaining large attenuation in a frequency band other than the passband.

1 is an equivalent circuit diagram showing a splitter which is an embodiment of the present invention.

FIG. 2 is a graph showing the input impedance characteristics of the filter used in the splitter of FIG. 1. FIG.

3 is a graph showing the input impedance characteristics of a filter versus the wideband of the filter.

4 is a graph showing input impedance characteristics of a longitudinal double mode type surface acoustic wave filter.

5 is an exploded view showing a specific structure of the splitter of FIG.

6 is an explanatory diagram showing a modification of the inductance element in the concrete structure of the splitter.

Fig. 7 is an explanatory diagram showing a modification of the capacitance element in the concrete structure of the splitter.

Fig. 8 is a graph showing the passage characteristic of the abnormal phase by the lumped constant element in comparison with the delay line.

Fig. 9 is a graph showing the phase characteristics of the phase shifter by the lumped constant element in comparison with the delay line.

Fig. 10 is a Smith chart showing the impedance characteristic of the abnormal phase by the lumped constant element in comparison with the delay line.

FIG. 11 is a graph showing frequency characteristics in the vicinity of a pass band in the divider of FIG. 1; FIG.

Fig. 12 is a graph showing the frequency characteristics in the high frequency region of the splitter of Fig. 1;

FIG. 13 is a Smith chart showing an impedance change of a second surface acoustic wave filter in the splitter of FIG. 1; FIG.

Fig. 14 is an equivalent circuit diagram showing a splitter which is another embodiment of the present invention.

FIG. 15 is a graph showing frequency characteristics in the vicinity of a passband in the denominator of FIG. 14; FIG.

FIG. 16 is a graph showing the frequency characteristics in the high frequency region of the splitter of FIG.

Fig. 17 is an equivalent circuit diagram showing a splitter which is another embodiment of the present invention.

Fig. 18 is an equivalent circuit diagram showing a splitter which is another embodiment of the present invention.

FIG. 19 is a graph showing frequency characteristics near a passband in the denominator of FIG. 17; FIG.

20 is a graph showing the frequency characteristics of the harmonic region in the splitter of FIG.

21 is a graph showing frequency characteristics in the vicinity of a pass band in the divider of FIG.

FIG. 22 is a graph showing frequency characteristics in the high frequency region of the splitter of FIG. 18; FIG.

Fig. 23 is an equivalent circuit diagram showing a conventional splitter.

FIG. 24 is a graph showing frequency characteristics in a passband in the splitter of FIG.

FIG. 25 is a graph showing frequency characteristics in the high frequency region of the splitter of FIG.

※ Explanation of symbols about main part of drawing ※

1: first signal terminal 2: second signal terminal

3: surface acoustic wave resonator 3P: parallel arm resonator

3S: Series Arm Resonator 4: Series Arm

5: parallel arm 6: abnormal phase

7: inductance element 8: capacitance element

9: phase shifter BF1: transmitter side filter (first surface acoustic wave filter)

BF2: Receive-side filter (2nd surface acoustic wave filter)

C: common terminal G: reference potential terminal

J: fork

t1, t2, t3, t4: signal terminal

In order to solve the above problems, the splitter according to the present invention comprises a common terminal, a surface acoustic wave element, and a signal terminal connected to the common terminal, and a plurality of surface acoustic wave filters having different pass bands, and a common terminal. An ideal phase by a lumped constant element disposed between a common surface and a surface acoustic wave filter having an input impedance affecting matching between the impedance in the pass band of the predetermined surface acoustic wave filter seen from the side and the impedance seen from the signal terminal side. Characterized in having a.

According to the present invention, it is advantageous to miniaturize the abnormal phase itself by the concentrated water purifying element, and to improve the layout between elements by reducing the crosstalk and phase shortening by approaching the delay line, which is concerned when using the delay line. This makes it possible to significantly reduce the size of the splitter using delay lines.

In addition, since the signal components in the frequency bands other than the pass band are attenuated by the abnormality of the lumped constant element, it is possible to achieve miniaturization and to obtain a large attenuation outside the pass band, thereby greatly improving the performance of the splitter. It can be improved.

Hereinafter, an embodiment of the present invention will be described in more detail with reference to the drawings. In the accompanying drawings, the same members are denoted by the same reference numerals and redundant descriptions are omitted. In addition, embodiment of this invention is an especially useful form to which this invention is implemented, Comprising: This invention is not limited to the Example.

The splitter of this embodiment is formed with a predetermined element on a piezoelectric plate made of, for example, LiTaO ₃ . That is, in Fig. 1, the common terminal C serving as the input / output of the signal and one signal terminal t1 are connected to the common terminal C through the branch point J, while the other signal terminal t2 is formed. The transmitting filter (first surface acoustic wave filter) BF1 and one signal terminal t3 connected to the one signal terminal 1 are connected to the common terminal C through the branch point J, The signal terminal t4 is provided with a receiving filter (second elastic surface wave filter) BF2 connected to the second signal terminal 2.

The transmitting filter BF1 and the receiving filter BF2 have frequency characteristics in which one passband becomes the other stopband, that is, different frequency bands and different frequency characteristics. The BF2 side has a higher passband than the transmission side filter BF1.

In this embodiment, however, the receiving side filter BF2 has a higher passband than the transmitting side filter BF1, but may be reversed. In addition, the piezoelectric plate is not limited to the above-mentioned one, but various piezoelectric plates such as ceramics such as piezoelectric plates made of LiNbO ₃ can be used.

Between the common terminal C and the receiving filter BF2, an idealizer 6 by a lumped constant element that attenuates signal components in a frequency band other than the pass band of the receiving filter BF2 is disposed.

Fig. 2 shows the impedance characteristics of the transmission filter BF1 seen from the common terminal C side. The impedance of the transmitting filter BF1 in the pass band of the receiving filter BF2 is an input impedance in the pass band of the second elastic surface wave BF2 and is an impedance seen from the signal input terminal t1 side. It is sufficiently high compared to 50Ω, and even if both are connected to the common terminal C, the transmission filter on the second surface acoustic wave filter BF2 matched to the impedance of the entire circuit in the pass band of the second surface acoustic wave filter BF2. (BF1) has no effect. In other words, the branching function is realized.

The impedance characteristic of the receiving filter BF2 seen from the common terminal C side is similarly shown in FIG. The impedance of the receiving filter BF2 in the pass band of the transmitting filter BF1 is as small as about 15 Ω. If both are connected to the common terminal, the impedance of the first elastic surface wave BF1 of the receiving filter BF2 is reduced. Impedance affects the impedance characteristic of the transmission filter BF1. That is, the offset function is impaired. Thus, some impedance converting means is required.

Thus, the branching function in the case where a plurality of filters are connected to the common terminal C is determined by the input impedance of each filter seen from the common terminal C side, and is not limited only to the configuration of the filter. As an example, FIG. 3 shows impedance characteristics of a wide band of the transmission filter BF1 and the reception filter BF2 seen from the common terminal C side. In the case of illustration, the input impedance rises toward the low frequency side, and an abnormal state is not necessary if the position of the pass band of the transmission filter BF1 is in the low frequency band.

The same applies to the case of using a rough double mode type surface acoustic wave filter. 4 is an impedance characteristic of the longitudinally coupled surface acoustic wave filter seen from the common terminal side. In this case, similarly to the above, the influence of the self-impedance characteristic on the impedance of the pass band of the other filter is determined by the position of the pass band of the other filter connected to the common terminal, and the presence or absence of the branching function and the necessity of the abnormal phase are determined. Similarly, the same principle applies to other modal surface acoustic wave filters, transversal surface acoustic wave filters, notch filters, and the like.

Next, the concrete structure of the splitter which splits the transmission side filter and the reception side filter connected to this common terminal C into small size and high performance without using a delay line is demonstrated using FIG.

As shown in Fig. 5, the splitter of the present embodiment is constituted by, for example, a substrate having a four-layered layer structure, and each substrate is bonded to each other by positioning. The transmission side filter BF1, the reception side filter BF2, the 1st signal terminal 1, the 2nd signal terminal 2, and the reference potential terminal G are formed in the board | substrate 11 located in the uppermost layer.

In addition, the transmission filter BF1 and the reception filter BF2 may employ a flip chip type mounted on the substrate 11 by face down bonding or a wire mounting type mounted on the substrate 11 by bonding wires. However, it is preferable to adopt the flip chip type in the lighting that the frequency characteristic does not change due to the point where high precision mounting is possible and there is no inductance of the bonding wire.

In the substrate 12 of the second layer, the substrate 13 of the third layer, and the substrate 14 of the fourth layer, which is the lowest layer, an ideal device 6 made of a lumped constant element composed of a π-type low pass filter (LPF) is provided. It is composed. In other words, an inductance element 7 formed of a wiring pattern having a serpentine shape is formed on the substrate 12, and both ends thereof are electrically connected to the common terminal C and the receiving filter BF2 of the substrate 11, respectively. have. In the substrate 13, the electrode patterns 8a of the two capacitance elements 8 connected to both ends of the inductance element 7 are provided with the electrode patterns 8a and a predetermined gap in the substrate 14. Opposite electrode patterns 8b are formed, respectively.

In addition, the patterns of the inductance element 7 and the capacitance element 8 are not limited to those shown in FIG. The inductance element 7 can be formed in the shape of a spiral as shown in FIG. 6 and the shape of the comb teeth shown in FIG. 7, for the capacitance element 8, for example.

In addition, the splitter may not be configured three-dimensionally in the stacked structure as shown, but may be configured two-dimensionally on a single plane.

In addition to the following description, in the splitter according to the present embodiment, the phase shifter 6 by the lumped constant element is one-packaged including the transmission filter BF1, the reception filter BF2, and the like. It may be an external type.

The ideal phase 6 by the lumped constant element is a π-type low pass filter which is represented by a π-type equivalent circuit when represented by a circuit diagram. In other words, the low pass filter, which is the phase shifter 6 by the lumped constant element, includes an inductance element 7 arranged on a transmission line extending from the common terminal C, and between the two sides of the inductance element 7 and the reference potential terminal G. Two capacitance elements 8 are disposed respectively. In addition to the following description, the ideal device 6 by the lumped constant element is composed of three elements, but may be composed of a desired number of elements such as five elements and seven elements.

Here, the lumped constant element can be significantly miniaturized compared to the delay line by the distribution constant. In the prior art, the splitter using a surface acoustic wave device is often constituted by a delay line which is a distribution constant element.

The characteristic impedance of the delay line is designed around 50Ω, which is the input impedance of the surface acoustic wave device to be connected or the driving impedance and the load impedance of the device. Therefore, the line width of the delay line needs to be 40㎛ ~ 120㎛, for example. In order to avoid deterioration of the characteristic impedance due to the coupling of delay lines, a large mounting area is inevitably necessary since the layout is laid out on the mounting board to achieve a desired phase while providing a gap between the lines.

The conventional splitter using the delay line is used in the 800MHz band and 2GHz band, and its mounting area is about 5mm x 5mm. However, considering the miniaturization, it was not possible to sufficiently satisfy the miniaturization in the delay line in advance.

On the other hand, in general, the abnormal phase is very low frequency, for example, in the HF or VHF band using a delay line is very rare. This is because the length of the delay line exceeds 1m and it is difficult to form on the circuit board. Use is limited if dimensional constraints are allowed. In these frequency bands, an ideal phase by a nearly concentrated constant element is used.

Therefore, the present inventors thought that the same principle can be used to solve the problem of miniaturization of the splitter. Although the configuration of the ideal phase using the lumped constant element in the microwave region is not an example, it is similarly required to be significantly smaller for the phase abnormality used for the surface acoustic wave element of one hundred thousandth of the electromagnetic wave wavelength.

However, in the conventional technique using the delay line, it is difficult to miniaturize this further. Therefore, the lumped constant element is used in the construction of the deviator of the splitter by the surface acoustic wave device.

The inductor, which is a lumped constant element, can obtain inductance as large as the line width, and the inductance of the inductance is increased by its own inductance as the gap between lines is narrow. Therefore, the delay line whose width is limited by the characteristic impedance is used. It is advantageous to miniaturization. In addition, the capacitor can be increased in capacity by narrowing the opposing conductor gap, which is advantageous for miniaturization. In addition, the capacitor can be easily realized by the laminated structure using the multilayer board. Therefore, the abnormality caused by the lumped water purification element was considered to be advantageous in miniaturization even when the number of elements is increased.

Further, in the above-mentioned claim 4 of Japanese Patent No. 3246906, a technique for constructing a delay line with an inductance L, which is a lumped constant element, is shown, but inductance alone is insufficient. The impedance characteristics of the connected filter in any inductor range are inducted over the entire band including not only out-of-band but also the pass band. This is because, in the case of a filter sufficiently matched, any part of the matching characteristics deteriorates. Therefore, as described in Japanese Patent No. 3246906, it is not preferable to form a branch wave in a circuit by only an inductance element in series, and it is more preferable to have a reactance element for matching.

In view of this, the present invention is arranged so that the impedance of the phase shifter at a desired frequency is sufficiently matched with the load / driving impedance (e.g., series inductance and parallel capacitor) and the phase shifting angle is realized at a desired frequency. Configured. In addition, in the frequency band to be suppressed outside the pass band, desired attenuation characteristics are obtained by the effect of the π or T type impedance element filter simultaneously realized.

In the configuration of the abnormal phase 6 to be inserted, the input / output impedance at a desired frequency of the abnormal phase is given by Zlump, the ideal rotation angle γ, and the number of elements. In the case of a simpler three-element configuration, if the desired angular frequency is ω,

Zlump = (L / C) ^1/2

γ = jω (LXC) ^1/2

Is given. The abnormality caused by the lumped circuit element is composed of four types of the T type low pass filter of FIG. 14, the π type high pass filter (HPF) of FIG. 1, and the T type high pass filter of FIG. 14, 17 and 18 will be described later.

All four device values are given above, and the impedance and abnormal rotation angle of the desired frequency are the same, and the transmission characteristics at different frequency bands are different. In the case of three elements of five elements, half of the value is given to γ, which is calculated in the same manner.

Fig. 8 compares the passage characteristics of the ideal phase of the four types of configurations according to the prior art and the present invention in the apparatus for the 2 GHz band. In the prior art, it can be seen that the abnormal phase caused by the configuration of the low pass filter and the high pass filter of the present invention attenuates the signal at or below a desired frequency band while transitioning to a low loss up to 10 GHz.

Fig. 9 compares the phase characteristics of the phase shifter according to the four types of configurations according to the prior art and the present invention in an apparatus for the 2 GHz band. It can be seen that the low pass filter gives an equal phase to the delay line at a desired frequency, and the high pass filter gives an equal phase at a different frequency. Zlump and γ can be arbitrarily given by the input / output impedance of the device and the load / drive impedance of the device by setting L and C.

Fig. 10 is a Smith chart of impedance characteristics of four abnormal phases according to the present invention. The prior art shows the transmission characteristics of a delay line matched to the load / drive impedance over a wide band, whereas the ideal phase of the present invention differs greatly in the reflection coefficient at other than the desired frequency.

As described above, the phase shifter according to the present invention is characterized by giving different transmission characteristics, different phase characteristics, and different impedance characteristics except the desired frequency, unlike the delay line. Therefore, it is possible to change the transmission characteristics in other frequency bands while satisfying the impedance characteristics and phase characteristics of the ideal phase at the desired frequency.

The frequency characteristics in the vicinity of the pass band for the splitter having the above structure are shown in FIG. 11 and the frequency characteristics in the high frequency region in FIG. Fig. 13 shows a Smith chart of the impedance characteristics of the receiving filter BF2 which is the receiving filter.

As shown in Fig. 11, the transmitting filter BF1 and the receiving filter BF2 have different passbands, and the receiving filter BF2 has a higher passband than the transmitting filter BF1. As shown in Fig. 12, the receiver side filter BF2 effectively operates the phase shifter 6 by the lumped constant element described above, so that the signal component of the high frequency region is higher than in the case of using the delay line (BF1 ', BF2'). It can be seen that this is greatly suppressed. In addition, as shown in FIG. 13, the impedance of the receiving side filter is clockwise in spite of the insertion of the concentrated constant element and the high frequency suppression characteristics are obtained. 11, it can be seen that the transmission characteristics of the pass band of the BF1 also give equivalent transmission characteristics.

By using such a splitter, the attenuation characteristic in the high frequency region of the receiving filter can be improved, and this miniaturization of the splitter is possible.

Here, the T-type low pass filter shown in the T-type equivalent circuit shown in FIG. 14 may be used for the idealizer 6 using the lumped constant element. That is, the low pass filter, which is the ideal phase 6 shown in Fig. 14, has two inductance elements 7 arranged in a transmission line extending from the common terminal C, the midpoints and the reference potential terminals of these inductance elements 7. It is provided with the capacitance element 8 arrange | positioned between (G).

The frequency characteristics in the vicinity of the pass band of the lumped constant element 6 shown in FIG. 14 are shown in FIG. 15, and the frequency characteristics in the high frequency region are shown in FIG. As shown in Fig. 15, the receiving side filter BF2 has a passband higher than the transmitting side filter BF1, and as shown in Fig. 16, a delay line is used for the receiving side filter BF2 (BF1 ', BF2). Compared with '), it can be seen that the signal component of the high frequency region is greatly suppressed.

Furthermore, the ideal phase 6 includes a high pass such as a π-type high pass filter represented by, for example, a π-type equivalent circuit shown in FIG. 17, or a T-type high pass filter represented by a T-type equivalent circuit as shown in FIG. Filters can also be used.

That is, the π-type high pass filter, which is the ideal phase 6 shown in Fig. 17, has a capacitance element 8 arranged on a transmission line extending from the common terminal C, and both sides of the capacitance element 8 and a reference. Two inductance elements 7 disposed between the potential terminals G are provided.

The T-type high pass filter, which is the ideal phase 6 shown in Fig. 18, includes two capacitance elements 8 arranged on a transmission line extending from the common terminal C, and the capacitance elements 8 of these capacitance elements 8; An inductance element 7 disposed between the midpoint and the reference potential terminal G is provided.

FIG. 19 shows the frequency characteristics in the vicinity of the pass band of the phase shifter 6 shown in FIG. 17, and FIG. 20 shows the frequency characteristics in the high frequency region. FIG. 21 shows the frequency characteristics in the vicinity of the pass band of the phase shifter 6 shown in FIG. 18, and FIG. 22 shows the frequency characteristics in the high frequency region.

As shown in Figs. 19 and 21, the receiving filter BF2 has a passband that is higher than the transmitting filter BF1, respectively. As shown in Figs. 20 and 22, the receiving filter BF2 has a low frequency in the frequency band other than the pass band (in this case, since the outlier 6 is a high pass filter) compared with the case where the delay line is used (BF1 ', BF2'). It can be seen that the signal component (in the region) is greatly suppressed.

In addition, the grounded inductance element 7 can provide resistance to a surge voltage. In addition, the DC component can be cut off by the capacitance element 8 inserted in series.

As described above, the ideal device using the lumped constant element as described above can realize any input / output impedance and an arbitrary amount of phase in a desired frequency band, so that they can be directly applied instead of the delay line in the conventional splitter.

In addition, in the present embodiment, the case of two filters has been described, but it is not limited to two splitters and can be applied to a multisplit by any number of filter combinations.

In the case of a multi-component filter, when the input impedance seen from the common terminal side affects the impedance of the pass band of another filter to which the common terminal side is connected, an ideal phase (6) by the lumped constant element of the present application between the common terminal and this filter Insert

By the phase shifter 6, the pass band of this filter can give a desired impedance characteristic. In addition, the impedance of this filter in the pass band of another filter to be connected is increased by the abnormal phase, and the impedance of this filter can reduce the influence on the impedance in the pass band of the other filter.

As can be seen from the above description, according to the present invention, the following effects can be obtained.

That is, in the splitter of the present application, it is possible to reduce the size of the splitter because it uses an ideal phase of the lumped constant element, which is significantly superior in space efficiency.

In the passband of the filter, signal components can be attenuated in frequency bands other than the passband while realizing desired impedance characteristics and phase characteristics as the splitter, thereby minimizing the splitter and improving performance.

In addition, surge resistance and DC cut function in the case of using a high pass filter can be added, thereby achieving multi-functionality.

Claims (13)

Common terminal, A plurality of surface acoustic wave filters formed of surface acoustic wave elements and having signal terminals connected to common terminals and having different pass bands; A concentration disposed between the common surface and the surface acoustic wave filter having an input impedance which affects matching between the impedance in the pass band of the predetermined surface acoustic wave filter seen from the common terminal side and the impedance seen from the signal terminal side The ideal group formed by the water purification element, Separator characterized in that provided. The method of claim 1, The surface acoustic wave device A first elastic surface wave filter formed of one surface acoustic wave element and having a signal terminal connected to the common terminal and having a predetermined pass band; A second surface acoustic wave filter formed of one surface acoustic wave element and having a signal terminal connected to the common terminal and having a pass band different from that of the first surface acoustic wave filter. A splitter comprising a. The method of claim 2, The first surface acoustic wave filter is formed of a surface acoustic wave resonator and is formed of a series arm resonator disposed on a serial arm and a surface arm resonator arranged on a parallel arm. Composed of a series arm resonator, The second surface acoustic wave filter is formed of a surface acoustic wave resonator and is formed of a series arm resonator disposed on a serial arm and a surface arm resonator arranged on a parallel arm. A splitter comprising a parallel arm resonator. The method according to claim 1 or 2, And the ideal phase is a low pass filter. The method of claim 4, wherein And the low pass filter is a T type low pass filter represented by a T type equivalent circuit. The method of claim 5, And the low pass filter comprises two inductance elements arranged on a transmission line extending from the common terminal, and a capacitance element disposed between the midpoint of the inductance elements and the reference potential terminal. The method of claim 4, wherein And the low pass filter is a? Type low pass filter represented by a? Type equivalent circuit. The method of claim 7, wherein And the low pass filter comprises an inductance element arranged on a transmission line extending from the common terminal, and two capacitance elements respectively disposed between both sides of the inductance element and a reference potential terminal. The method according to claim 1 or 2, And the phase shifter is a high pass filter. The method of claim 9, And the high pass filter is a T type high pass filter represented by a T type equivalent circuit. The method of claim 10, The high pass filter includes two capacitance elements arranged on a transmission line extending from the common terminal, and an inductance element disposed between the midpoint of the capacitance elements and the reference potential terminal. Cultivator. The method of claim 9, And the high pass filter is a? Type high pass filter represented by a? Type equivalent circuit. The method of claim 12, The high pass filter includes a capacitance element arranged on a transmission line extending from the common terminal, and a splitter comprising two inductance elements disposed between both sides of the capacitance element and a reference potential terminal. .