KR20020035626A - Wireless terminal - Google Patents

Abstract

The wireless terminal includes a ground conductor 1102 and a transceiver coupled to the antenna feed, with the antenna feed directly connected to the ground conductor 1102. In one embodiment, the ground conductor is conductive case 1102. Coupling may be through a parallel plate capacitor 504 formed of the surface 1108 of the case 1102 and the plate 506. Case 1102 serves as an effective broadband radiator, which eliminates the need for a separate antenna. In a variation of this embodiment, the slot 1112 is provided to increase the resistance of the case 1102, as seen by the transceiver, to increase the radiating bandwidth of the terminal.

Description

Wireless terminal {WIRELESS TERMINAL}

Generally, a wireless terminal, such as a mobile telephone handset, is an external antenna, such as a helix or meander line antenna in normal mode, or a Planar Inverted-F Antenna (PIFA) or similar Merge with an internal antenna, such as an antenna.

Since such antennas are small (in terms of wavelength), they are narrowband due to the fundamental limitations of small antennas. In general, however, cellular radio communication systems have a fractional bandwidth of at least 10%. For example, achieving such bandwidth from PIFA requires a significant volume because there is a direct relationship between the bandwidth and the volume of the patch antenna, but such volume, as the current trend is towards smaller handsets, It is not readily available. Thus, due to the above limitations, it is not possible to achieve effective broadband radiation from small antennas at current wireless terminals.

A further problem with known antenna arrangements for wireless terminals is that they are generally unbalanced and therefore are firmly coupled to the terminal case. As a result, a significant amount of radiation is emitted from the terminal itself rather than the antenna.

The present invention relates to a wireless terminal, for example a mobile telephone handset.

1 shows a model of an asymmetric dipole antenna, showing a combination of an antenna and a wireless terminal.

2 is a graph illustrating the separability of impedance components of an asymmetric dipole.

3 is an equivalent circuit diagram of a combination of a handset and an antenna.

4 is an equivalent circuit diagram of a capacitively back-coupled handset.

5 is a perspective view of a basic capacitive bag coupled handset.

FIG. 6 is a graph of simulated return loss (S ₁₁ ) versus frequency (f) in MHz for the handset of FIG. 5; FIG.

FIG. 7 is a Smith chart showing the simulated impedance of the handset of FIG. 5 over a frequency range (1000-2800 MHz). FIG.

FIG. 8 illustrates a simulated resistor of the handset of FIG. 5. FIG.

9 is a perspective view of a narrow capacitive back coupled handset.

10 is a graph showing simulated resistance of the handset of FIG.

11 is a perspective view of a slotted capacitive back coupled handset.

FIG. 12 is a graph of simulated return loss S ₁₁ in dB versus frequency f in MHz for the handset of FIG.

13 is a Smith chart showing the simulated impedance of the handset of FIG. 11 over a frequency range (1000-2800 MHz).

14 is a plan view of a capacitive bag coupled prototype.

FIG. 15 is a graph of measured return loss (S ₁₁ ) in dB versus frequency (f) in MHz as measured for the prototype of FIG. 14.

FIG. 16 is a Smith chart showing the measured impedance of the prototype of FIG. 14 over a frequency range (800-3000 MHz).

17 is a plan view of a capacitive back coupled prototype using inductive elements.

FIG. 18 is a graph of measured return loss (S ₁₁ ) in dB versus frequency (f) in MHz as measured for the prototype of FIG. 17. FIG.

FIG. 19 is a Smith chart showing the measured impedance of the prototype of FIG. 17 over a frequency range (800-3000 MHz). FIG.

It is an object of the present invention to provide a wireless terminal having effective radiation characteristics over a wide bandwidth.

According to the present invention, there is provided a wireless terminal comprising a transceiver connected to an antenna feed, and a ground conductor, wherein the antenna feed is connected to a ground conductor.

The present invention recognizes that the impedance of the antenna and wireless handset is similar to that of a detachable asymmetric dipole, which does not exist in the prior art, and that the antenna impedance is non-radiating coupling. It is based on the further recognition that the device can be replaced.

Embodiments of the present invention will now be described by way of example with reference to the accompanying drawings.

In the drawings, like reference numerals have been used to indicate corresponding features.

1 shows a model of the impedance seen by a transceiver in transmit mode in a wireless handset at an antenna feed point. Impedance is modeled as an asymmetric dipole, where the first arm 102 represents the impedance of the antenna, the second arm 104 represents the impedance of the handset, and both arms are driven by the source 106. . As shown in the figure, the impedance of such a device is substantially equal to the sum of the impedances of each arm 102, 104 driven separately from the virtual ground 108. This model can equally well be used for reception by replacing the source 106 with an impedance that represents the transceiver's impedance, although this is slightly more difficult to simulate.

The validity of this model is the well-known Numerical Electromagnetics Code (NEC) values for the first arm 102, 40 mm long and 1 mm in diameter, and the second arm 104, 80 mm in length and 1 mm in diameter. Electromagnetic code) to check by simulation. FIG. 2 shows the results for the real and imaginary parts of the impedance R + jX of the devices Ref R and Ref X combined with the results obtained by individually simulating the impedances and summing the results. It can be seen that the results of the simulations are very similar. When impedance is difficult to accurately simulate, the only significant deviation is in the half-wave resonant region.

As can be seen at the feed of the antenna, an equivalent circuit for the combination of the antenna and the handset is shown in FIG. R ₁ and jX ₁ represent the impedance of the antenna, while R ₂ and jX ₂ represent the impedance of the handset. From this equivalent circuit, it can be inferred that the power ratio radiated by the antenna P ₁ and the handset P ₂ is given as follows.

If the size of the antenna is reduced, its radiation resistance R ₁ will also decrease. When the antenna is infinitely small, the radiation resistance R ₁ will drop to zero and all radiation will come from the handset. This condition will be advantageous if the handset impedance is suitable for the source driving the arm 106 and if the capacitive reactance of the infinitesimal antenna can be minimized by increasing the capacitive back coupling of the handset. Can be.

Through this modification, the equivalent circuit is transformed into the circuit shown in FIG. Therefore, the antenna is replaced with a physically very small back coupling capacitor, which is designed to have a large capacitance for maximum coupling and minimum reactance. The residual reactance of the back coupling capacitor can be tuned into a simple matching circuit. Due to the precise design of the handset, the resulting bandwidth can be much wider than a conventional antenna and handset combination, because the handset has a low Q radiation element (simulation shows that typical Q is about 1). While conventional antennas generally have a Q of about 50.

A basic embodiment of a capacitive back coupled handset is shown in FIG. 5. Handset 502 generally has a size of 10 × 40 × 100 mm of a modern cellular handset. A parallel plate capacitor 504 with a size of 2 × 10 × 10 mm is a 10 × 10 mm plate 506 above 2 mm of the top edge 508 of the handset 502 at a location typically occupied by a much larger antenna. ) Is formed by mounting. The resulting capacitance is about 0.5pF, which is between the capacitance (increased by reducing the gap between the handset 502 and the plate 504) and the coupling efficiency (depending on the distance between the handset 502 and the plate 504). Represents a compromise. The capacitor is fed through a support 510 that is insulated from the handset case 502.

After matching, the return loss S ₁₁ of this embodiment is simulated using a High Frequency Structure Simulator (HFSS), available from Ansoft Corporation, which results in 1000MHz and It is shown in Figure 6 for a frequency f between 2800 MHz. Two conventional inductor " L " networks are used to match at 1900 MHz. The resulting 7dB return loss (approximately 90% of the copied input power) has a bandwidth of approximately 60MHz, or 3%, which is useful but need not be as large as possible. A Smith chart showing the simulated impedance of this embodiment over the same frequency range is shown in FIG. 7.

The low bandwidth is because the handset 502 provides an impedance of approximately 3-j90 Hz at 1900 MHz. 8 shows resistance variation over the same frequency range as before, simulated using HFSS. This can be improved by redesigning the case to increase the resistance.

One way this can be done is to reduce the width of the handset 502 because the resistance increases in much the same way as the dipole when the radius of the dipole is reduced. 9 shows a second embodiment with a narrow capacitive back coupled handset 902. The handset 902 has a size of 10 × 10 × 100 mm, while the size of the plate 506, the top surface 908 of the handset 902, and the capacitor 504 formed from the support 510 is the size of the previous implementation. It does not change from ancient times. The simulation is again performed to determine the variation of the resistance of this embodiment, the result of which is shown in FIG. 10. This clearly demonstrates that the use of a narrow handset provides a wider bandwidth where the resistance is higher than the resistance of the basic configuration. The length of the handset can be optimized to provide a wide bandwidth centered on a particular frequency by shifting the resonant frequency of the structure. For handsets of fixed length, horizontal slots (ie, slots across the width of the handset) can be used to electrically short or extend the handset.

An alternative way to increase the resistance of the case is to insert a vertical slot (ie, the length of the handset, or a slot parallel to the major axis). FIG. 11 shows a third embodiment with a slotted capacitive back coupled handset 1102 having a capacitor 504 and a slot 1112 with a depth of 33 mm in the case. The size of the support 510 and the capacitor 504 formed from the plate 506 and the top surface 1108 of the handset 1102 do not change from the previous embodiment. The provision of slot 1112, as can be seen by the transceiver, significantly increases the resistance of the case in the region of 1900 MHz and allows the low-Q case to match 50 Hz without significant loss in bandwidth.

The return loss S _{11 of} this embodiment is simulated again using HFSS, and the result is for frequencies f between 1000 and 2800 MHz using two inductor matching networks similar to those used in the base embodiment. 12 is shown. The resulting bandwidth at 7dB return loss is greatly increased at approximately 350MHz, or 18%, which reaches the bandwidth needed to simultaneously cover the UMTS and DCS 1800 bands. A Smith chart showing the simulated impedance of this embodiment over the same frequency range is shown in FIG. 13.

Prototypes are built to demonstrate the practical application of the simulation results described above. 14 is a top view of a prototype, which includes a copper ground plane 1402 having a size of 40 × 100 mm on a 0.8 mm thick FR4 circuit board (with a dielectric constant of 4.1 measured). A slot 1412 of 3 × 29.5 mm is provided in the ground plane, and a plate 506 of 10 × 10 mm is located 2 mm above the corner of the ground plane 1402. The co-extensive portion of the ground plane 1402, and the plate, form a parallel plate capacitor as in the embodiment described above. The capacitor is fed through a coaxial cable 1404 and a vertical pin 510 attached to the rear surface of the circuit board.

The return loss S _{11 of} this embodiment is measured without a match, and then the match is added to the simulation. Added matches are 3.5 nH series inductors and 4 nH shunt inductors, similar to those used in the simulations described above. The result is shown in FIG. 15 for the frequency f between 800 and 3000 MHz. The resulting bandwidth at 7 dB return loss is approximately 350 MHz, or 22%, centered at 1600 MHz, which is approximately the non-bandwidth needed to cover the UMTS and DCS 1800 bands simultaneously. A Smith chart showing the impedance of this embodiment over the same frequency range is shown in FIG.

The above described embodiment is based on capacitive coupling. However, any other sacrificial (not copied) coupling element can be used, for example, instead of inductive coupling. In addition, the coupling element can be modified to assist in impedance matching. For example, capacitive coupling can be achieved through inductive elements that have the advantage of not requiring any additional matching components.

As an example of this technique, additional prototypes are fabricated, which are shown in plan view in FIG. This prototype is similar to that shown in FIG. 14, with the difference that the plate 506 is slightly offset from the corner of the ground plane 1402 and is no longer fully metallized, instead the feed pin 510 Is provided with a spiral track 1706 connected at one end. The length of the track 1706 is chosen to resonate at the required frequency, in this embodiment approximately 1600 MHz. The track 1706 is fed through a stripline 1704 on the back of the circuit board.

The return loss S _{11 of} this embodiment is measured without matching. The result is shown in FIG. 18 for a frequency f between 800 and 3000 MHz. The resulting bandwidth at 7dB return loss is approximately 135MHz, or 9% centered at 1580MHz, and it is believed that this bandwidth can be significantly improved by further optimization and matching. A Smith chart showing the impedance of this embodiment over the same frequency range is shown in FIG. 19.

In the above embodiment, the conductive handset case is a radiation element. However, other grounding conductors at the wireless terminal may perform similar functions. Examples include conductors, such as ground planes, used in the metallization region of EMC shielding and printed circuit boards (PCBs).

By reading this specification, other variations will be apparent to those skilled in the art. Such modifications may include other features that are already known in the design, manufacture, and use of wireless terminals and components of the wireless terminal, and that may be used instead of or in addition to the features already described herein. Although the claims are described herein for a particular combination of features, the scope of the disclosure herein also includes any new feature or any new combination of features disclosed explicitly or implicitly herein, or a generalization thereof. It is to be understood that the present invention includes whether or not it relates to the same invention currently claimed in any claim and whether or not to alleviate some or all of the same technical problems solved by the present invention. Applicants should therefore note that new claims may be formed of such features and / or combinations of features during the performance of this specification or any further application derived herein.

In the specification and claims, the elements described in the singular do not exclude the presence of a plurality of such elements. Moreover, the word "comprising" does not exclude the presence of other elements or steps than those written.

As mentioned above, the present invention finds use in wireless terminals such as mobile telephone handsets.

Claims (9)

A wireless terminal comprising a transceiver coupled to an antenna feed, and a ground conductor, And the antenna feed portion is connected to the ground conductor. The wireless terminal of claim 1, wherein the antenna feed portion is connected to the ground conductor via a capacitor. 3. The wireless terminal of claim 2, wherein the capacitor is a parallel plate capacitor formed of a portion of the ground conductor and a conductive plate. The wireless terminal of claim 1, wherein the antenna feed portion is connected to the ground conductor by a capacitance between an inductive element and the ground conductor. 5. A wireless terminal according to any one of the preceding claims, wherein a slot is provided in the ground conductor. 6. The wireless terminal of claim 5, wherein the slot is parallel to the major axis of the terminal. The wireless terminal of claim 1, wherein the ground conductor is a handset case. 7. A wireless terminal according to any one of claims 1 to 6, wherein the ground conductor is a ground plane of a printed circuit board. 9. A wireless terminal according to any one of claims 1 to 8, wherein a matching network is provided between the transceiver and the antenna feed.