KR20020034830A - Swept performance monitor for measuring and correcting rf power amplifier distortion - Google Patents

Abstract

PURPOSE: A swept performance monitor for measuring and correcting RF power amplifier distortion is provided to additionally increase gain to offset the fact that the signal level extracted from the output amplifier is very low and to prevent producing IMDs in the swept receiver's mixer. CONSTITUTION: The switchless architecture of the swept performance monitor includes the Wilkinson splitter used to coupled the output of the swept oscillator(45) to swept input and output receivers(20,30), respectively, replaced by a resistive Y splitter(150), which has a flatter frequency response than a Wilkinson splitter, and thus complies with the desire to maintain the output level of the VCO constant as it is swept over frequency. The resistive Y splitter(150) further includes a first resistor arm(151) coupled to the output of oscillator(45). A second resistor arm(152) is coupled through buffer amplifier(55) to the second input(23) of the input receiver's mixer(22). A third resistor arm(153) is coupled through buffer amplifier(57) to the second input(33) of the output receiver's mixer(32).

Description

Sweeping performance monitoring device for distortion measurement and correction of RF power amplifiers {SWEPT PERFORMANCE MONITOR FOR MEASURING AND CORRECTING RF POWER AMPLIFIER DISTORTION}

This application is a Continuation-In-Part of pending US patent application Ser. No. 09 / 479,723, filed Jan. 7, 2000, hereafter referred to as the '723 application. Claiming the benefit of pending preliminary US application Ser. No. 60 / 175,279, filed with, each of which is assigned to the assignee, the present application and its contents contained herein.

In general, the present invention relates to a radio frequency (RF) communication system, in particular a "low" carrier relative to a "low" carrier to intermod ratio (C / I) relative to the input path. An RF power amplifier distortion correction mechanism for controlling a pre-distortion circuit controlled by an adaptive digital signal processor installed in an RF amplifier having a carrier. This invention may detect the distortion energy output at the output of the RF amplifier by sweeping the input and output receivers and using a sweeping oscillator to locate and separate the RF carrier components at the amplifier output. Once detected, the distortion energy may be controllably removed by its predistortion.

As described in the pending '723 application mentioned above, a telecommunications service provider follows very stringent bandwidth constraints, including the requirement that the excess of energy outside the licensed channel or band of interest is rapidly attenuated (e.g., about 50 dB). Are specified in the specifications and specifications of the Federal Communications Commission (FCC). Such limitations can easily overcome traditional modulation forms such as FM, but it is difficult to achieve digitally based modulation formats such as M-ary modulation simultaneously.

Using such modulation techniques to sufficiently attenuate the sidebands to meet industry or management-based standards requires very linear signal processing systems and components. Although relatively linear components are obtained at a reasonable cost for the relatively low bandwidth (baseband) of the telephone network, linearizing components such as power amplifiers at such RF frequencies can be very expensive.

The basic difficulty in linearizing an RF power amplifier is the fact that the power amplifier is essentially a nonlinear device and produces unwanted intermodulation distortion (IMD). This IMD appears as the spurious signals of the amplified RF output signal that are separate and distinct from the RF input signal. Another indication of IMD is spectral regrowth or spreading of the compact spectrum into the spectrum, which does not appear in the RF input signal. With this distortion, the phase-amplitude of the amplified output signal is separated from the phase-amplitude of the input signal, and this distortion may be thought of as temporary (and unwanted) source amplifier modulation of the RF input signal.

A blind technique for implementing linear RF power amplifiers is to design the amplifier as a large, high power device, but at low power levels (ie, a small percentage of the amplifier's rated output power) where the transfer function of the RF amplifier is relatively linear. Operate the amplifier. Disadvantages of this solution are excessive disadvantages-high cost and large size RF devices.

Other prior arts that overcome this disadvantage include feedback correction techniques, feedforward corrections, and predistortion corrections.

At this time, the feedback correction technique uses bipolar envelope correction (as described in US Pat. No. 5,742,201) and Cartesian feedback when the distortion component at the RF amplifier output is used to directly modulate the input signal to the amplifier in real time. It includes. This feedback technique has the advantage of self-convergence, just like other feedback techniques in other design areas. However, systems using negative feedback are stable on limited bandwidth, preventing applications in wideband environments such as multi-carrier or W-CDMA. However, in this respect, the feedforward and predistortion correction are not limited. In the feedforward approach, the error (distortion) in the output signal of the RF amplifier is extracted, amplified to an appropriate level, and then (in ideally) the RF amplifier is re-injected with the opposite phase into the amplifier's output path. Eliminate distortion effectively.

With predistortion correction, the signal is modulated upstream of the RF input signal path of the RF amplifier. This ideal predistortion signal is characterized by the inversion of the distortion expected at the output of the high power RF amplifier, effectively eliminating distortion when subjected to the distortion transfer function of the RF amplifier.

The predistortion or feedforward may be adapted to extract error signal components at the output of the RF amplifier and adjust its control signal (s) to continuously minimize distortion at the amplifier's output in accordance with the extracted error behavior of the RF amplifier. have.

One conventional mechanism for extracting the error signal component involves injecting a pilot (tone) signal into the signal flow path through an amplifier and measuring the amplifier's response. A fundamental disadvantage to the use of pilot tones is the need for dedicated pilot generating circuitry and the difficulty of placing the pilot tones within the signal bandwidth of the amplifier. Another approach uses a high intercept receiver to detect low level distortion in the case of high power carriers that add real complexity and cost.

According to the invention described in the '723 application, in the case of a multi-frequency input signal, the RF power amplifier distortion is accurately measured by tuning the RF input / output receiver using a sweeping local oscillator. If the distortion is corrected by an adaptive predistortion circuit installed in the input path to the RF amplifier with a relatively " low " carrier to intermode ratio (C / I), the sweep local oscillator configuration is shown in FIG. Corresponding to FIG. 1 of the application). The relatively low C / I ratio means that the RF amplifier cannot effectively distinguish the RF carrier level from the level of the intermodulation result. As a non-limiting example, the low C / I ratio term may be considered to apply to RF amplifiers with intermodulation results above -50 dBC.

In the configuration of Fig. 1, the RF input signal RF _in to be amplified is connected to the input port 11 of the signal input path to the RF power amplifier 10, so that its distortion characteristic is controllably blanked distortion energy detector ( 100). In order to control the RF input signal RF _in for the carrier energy, its RF input port 11 is controllably tuned or firstly coupled to the first input 21 of the mixer 22 of the sweeping input receiver 20. And a digital control predistorter 14 provided in the signal input path to the RF power amplifier 10.

The predistorter 14 at this time may operate to adjust the amplitude and phase of the RF input signal to the RF amplifier 10 dynamically and may include a vector modulator driven by a complex polynomial work function. The predistorter 14 also supplies the weight coefficients W ₀ , W ₁ , W ₂ ,. Supplied on the multi-link 15 by the parameters for updating the digital signal processor (DSP) 16 and the performance of controlling them. ..W _N is connected to receive. The DSP executes sixteen one or more error optimization algorithms (e.g., squared or least average squared) to controllably adjust the distortion introduced into the RF signal input path through its predistortion section 14.

The output of the RF power amplifier 10 is connected to the RF output port RF _out and is the first input of the mixer 32 in the control receiver tuned or sweeped output receiver 30 via a second directional coupler 17. (31). The output of the directional coupler 17 represents the original amplified RF input signal and any intermodulation (spectral regrowth) distortion result IMD introduced by the RF amplifier.

Each input and output receiver 20, 30 is controlled by a digital sweep-control signal generated by the DSP 16. To this end, the digital sweep-control signal line 17 is connected to a digital-to-analog converter (DAC) 41, filtered by a low pass filter 43, and then a voltage controlled oscillator (VCO). Generates an analog output sweep voltage connected to " 45 ". The output of the VCO 45 at this time is connected to the input port 51 of the Wilkinson splitter 50.

The Wilkinson splitter 50 has a first output port 52 connected to the second input 23 of the mixer 22 via a buffer amplifier 55 and a mixer 32 via a buffer amplifier 57. It has a second output port 53 connected to a second input 33. The IF output 25 of the mixer 22 is filtered by a broadband bandpass filter 61 and connected via a buffer amplifier 63 to a carrier power detector 65 represented as a diode with a cathode coupled to ground. . The carrier power detector 65 at this time is connected to the threshold detector 67, and its output is connected to the blanking detection input 18 of the DSP 16, and the first in the output receiver 30. And control ports 71 and 81 of the second high isolation switch 70 and 80, respectively.

If there is no output of the carrier power detector 65 that exceeds a predetermined threshold associated with the RF carrier signal, the output of the threshold detector 67 is in the first logic state. However, when the carrier power detector 65 detects power exceeding a predetermined threshold, the output of the threshold detector 67 changes to the second logic state. The change in the state of the output of the threshold detector 67 controls the blanking signal input 18 to the DSP 16 to controllably empty the output receiver 30 through measurement of the RF amplifier distortion.

For this reason, the IF output 35 of the mixer 32 is connected to the 1st input port 72 of the 1st isolation switch 70 as shown, and the 2nd input port 73 has an impedance interruption | blocking. It is. The isolation switch 80 has an output port 84 connected via a IF buffer amplifier 85 to a diode configuration (distortion) power detector 91 having a cathode grounded by a capacitor coupling, and is provided by the RF amplifier 10. It measures the distortion power within the generated output receiver bandwidth.

This distorted power detection diode 91 has its output (cathode) connected to an analog-to-digital converter (ADC) 95 via a low pass filter 93, and this digitized output is connected via a link 97 to the DSP ( 16 is connected to the distortion detection input 19. The digitized output of this distortion power detection diode 91 is integrated by the DSP 16 using one or more error minimization algorithms to control the variable attenuation and phase shift components in the predistortion section 14. And is processed.

In response to the operation of the controllable distorted energy measuring unit 100, the signal path through the output receiver 30 passes through the first and second isolation switches 70 and 80 to the distorted power detection diode 91. Is typically connected to. The output of the power detection diode 91 is sampled, digitized and connected to the distortion input 19 of the DSP 16. When the DSP 16 sweeps the control voltage input to the VCO 45, the tuning frequencies for each of the input and output receivers 20, 30 are generally stamped. During this frequency sweep, the power detected by the carrier power detector diode 65 of the input receiver 20 is applied to a threshold detector 67 in which the threshold value between the carrier wave and the distortion is different.

Unless the threshold detector 67 is exceeded, it means that the output of the receiver 30 is the distortion power generated by the RF power amplifier 10. This distortion power is digitized, connected to the processor 16, and integrated over the entire sweep to control the predistortion correction unit 14. However, when the output of the carrier power detector 65 exceeds the threshold of the threshold detector 67-indicating that the output receiver is tuned in close proximity to the carrier energy-the output of the threshold detector 67 is in a state. To change. This change in status signal empties the DSP 16 and blocks its signal path through the isolation switches 70 and 80, so that the distortion correction operation performed by the DSP 16 is not affected by the carrier wave.

Selective carrier based blanking of the distortion measurement receiver circuit effectively prevents saturation of the output of the IF amplifier 85, allowing the use of lower IP3 components. The bandwidth of the input receiver 20 required by the bandpass filter 61 can be made slightly wider than the bandwidth of the output receiver 3, so as to adequately provide a guard band for the switching operation.

In the configuration of FIG. 1, the high isolation switches 70, 80 serve to improve the dynamic range of the receiver in two ways. First, they prevent the IF component from being overloaded by the carrier because the carrier is sweeping through the IF passband. This undesirable overload drives the amplifier and detector to saturate, and these elements require a certain period of time to saturate and resume normal active mode operation. Next, the isolation switch prevents the bandpass filter circuit from being excited by a sudden transient, so that the carrier is sweeped into the IF passband. Because bandpass filters are relatively narrow and high-resonant filters, these step transients cause the filter to 'ring' with a relatively long delay.

Both of these effects may cause errors in ACPR measurements in the portion of the spectrum immediately adjacent to the desired carrier. Unfortunately, this is the area in the spectrum where most of the IMD produced by the RF power amplifier is concentrated and minimization of the IMD is highly desirable. The effect of this error is that the optimization of the ACPR is unbalanced between the two sidebands containing the carrier. One sideband may be optimized for better ACPR than required at the expense of performance in the other sideband.

This effect is most undesirable if all energy with one or more carriers can pass entirely within the bandwidth of the IF filter (ie, the ratio of carrier C bandwidth to IF bandwidth is low). This is the maximized ring in the bandpass filter. Also, if a larger receive dynamic range is desired, it is not desirable in a multi-carrier amplifier, so some carriers have all of their energy limited to a relatively small bandwidth, while others have significant bandwidth (e.g. IS-95 CDMA). They have their energy to 'spread'.

The effect of that error is more undesirable when a high C / I ratio (greater than about 50 dB) of the amplifier system is specified and requires a high dynamic range. Finally, this effect is undesirable when ultra fast VCO sweep ratios are needed and will not be practical or beneficial to slow down sweep ratios.

There are cases where only a small number of wideband signals (eg 5) are present and can accept slow sweep frequencies. According to the present invention, in this case, the above-mentioned advantage of using a high isolation switch in the signal flow path of the output receiver of the performance measurement and correction configuration of the aforementioned '723 application is not required. In these cases, the configuration of the '723 application can be simplified by replacing the controllably blanked isolation switch of the sweep output receiver with a buffer amplifier-filter stage. This buffer amplifier-filter stage provides additional gain to offset the very low signal level extracted from the RF amplifier and prevents the generation of IMD in the mixer of its sweeping receiver.

As described below, in the 'switchless' distortion measurement and correction configuration of the present invention, the Wilkinson splitter used to couple the sweep oscillator to each mixer of the input / output receiver is a resistive Y splitter having a platter frequency response than the Wilkinson splitter. And the output level of the VCO remains constant when turned to frequency. In addition, the input and output signal paths through the mixers of the input receivers are connected to improve reverse isolation through one or more dependent buffer amplifier stages and to prevent the VCO signal from being reversed into the input path to the RF amplifier. In addition, the parameters of the amplifier stage and the bandpass filter of the input receiver are selected to prevent damaging any device.

In addition, the DSP uses the sampled input power information to control the threshold setting of the threshold detector. Applying a threshold to the sampled input power gives the DSP the ability to apply carrier detection thresholds at different power levels. In addition, the high isolation switch of the sweep output receiver is removed and the IF output of the mixer of the output receiver is connected via a cascade of amplifier gain stages and a bandpass filter.

The bandpass filter of the output receiver can be the same as the bandpass filter in the sweep input receiver. This helps to make the bandwidth of the output receiver slightly narrower than the input receiver and to minimize the number of other components used in the circuit. In both input and output receivers, the gain of the receiver amplification stage and the losses through the bandpass filter are chosen to avoid damaging the power detection diode and any other components.

1 is a diagram illustrating a RF power amplifier distortion measurement and predistortion correction configuration in accordance with the invention described in US patent application Ser. No. 09 / 479,723, FIG.

2 is a diagrammatic representation of a variant of FIG. 1 for realizing a switchless RF power amplifier distortion measurement and correction configuration of the present invention.

* Description of the symbols for the main parts of the drawings *

10: RF amplifier 11: input path

13 directional coupler 14 predistortion correction circuit

16: DSP20: input receiver

22: input mixer 30: output receiver

32: Mixer 41: DAC

43, 93: low pass filter 45: VCO

57, 63: buffer amplifier 61: broadband bandpass filter

65 carrier power detector 67 threshold detector

85: IF buffer amplifier 95: ADC

100: distortion energy detector 121, 125: buffer amplifier stage

150: resistive Y splitter 175, 177, 178: bandpass filter

Before describing in detail the novel and improved 'switchless' RF power amplifier distortion measurement and correction mechanism of the present invention, it should be noted that the modified configuration of the present invention controls the operation of such circuits and components. Inherent in certain configurations of conventional RF communication circuitry associated with processing components and ancillary control circuitry. As a result, the construction of such circuit components and methods of interfacing with other communication system equipment are, for the most part, described in easily understood block diagrams showing only the details belonging to the present invention, thereby benefiting from the description herein. The disclosure with details that will be readily apparent to those skilled in the art will not be ambiguous. Thus, first, the block diagram description shows the main components of the RF amplifier distortion measurement and correction system divided into simple functional groups, so that the present invention can be more easily understood.

Referring now to FIG. 2, there is schematically shown a switchless version of the RF power amplifier distortion measurement and correction configuration of FIG. As in the circuit of FIG. 1, the amplifier distortion is caused by an adaptive predistortion circuit 14 installed in the input path 11 to the RF amplifier 10 having a relatively " low " carrier to intermode ratio (C / I). Corrected.

As pointed out above, due to the relatively low C / I ratio, the RF amplifier cannot be effectively distinguished from the RF modulation level from that of the intermodulation result, and is considered to be applicable to an RF amplifier having an intermodulation result above -50 dBC. It may mean an amplifier.

The switchless configuration of FIG. 2 differs from the distortion measurement and correction configuration of FIG. 1 in the following. First, the Wilkinson splitter 50, which is connected to the output of the sweep oscillator 45 and used to sweep the input and output receivers 20 and 30, respectively, has a resistive Y splitter 150 having a platter frequency response than the Wilkinson splitter. In turn, according to the requirement to keep the output level of the VCO 45 constant when it is turned to frequency. This resistive Y splitter 150 has a first resistance arm 151 connected to the output of the oscillator 45. The second resistance arm 152 is connected to the second input 23 of the mixer 22 of the input receiver via the buffer amplifier 55. The third resistance arm 153 is connected via a buffer amplifier 57 to the second input 33 of the mixer 32 of the output receiver.

According to a second aspect of the invention, the input signal path through the combiner 13 to the input mixer 22 is connected via one or more dependent buffer amplifier stages 121 to improve the reverse isolation so that the VCO signal is directional. It serves to prevent back connection into the input path to the RF amplifier 10 through the combiner 13. The gain of the amplifier stages 121, 125, 63 of the input receiver and the losses through the combiner 13, mixer 22 and bandpass filter 61 are selected to strategically prevent damage to any component.

According to a third aspect of the invention, in addition, the output of the carrier power detector 65 coupled to the threshold detector 67 is digitized by the ADC 165, which is digitized by the DSP 16. Is connected to the input power detection input 166. The DSP 16 uses this sampled input power information to generate a threshold setting and is connected to the threshold detector 67 via the DAC 168 by the threshold control output port 167. The adaptive threshold based on the sampled input power gives the DSP the ability to apply the carrier detection threshold to different power levels.

The high isolation switches 70 and 80 of the postmark output receiver 30 according to the fourth variant of the invention are removed. For this reason, the IF output 35 of the mixer 32 of the output receiver is connected to the amplifier gain stage of the slave structure via a narrower band pass filter. As in the non-limiting example, the mixer output 35 may be connected to the slave amplifiers 176 and 177 through an amplifier stage 135 and one band bandpass filter 175. The output of the amplifier stage 177 is connected to the IF buffer amplifier 85 through the band bandpass filter 178. These additional gain stages are used to prevent the IMD from occurring in the mixer 32 since the signal level extracted from the output combiner 17 is very low. The band pass filters 175 and 178 can be the same as the band pass filter 61 in the post-input receiver. This makes the bandwidth of the output receiver slightly narrower than the input receiver, helping to minimize the number of other components used in this circuit. As in this input receiver, the gain of the receiver amplification stage and the loss through the bandpass filter are selected so that the power detection diode 91 and any other components (e.g., the detection stage of the slave amplifier, filter and IF section are hard saturated). to prevent damage). For example, its saturated output power performance of amplifier stage 85 may be about 12-13 dBm, while detector 91 may safely absorb 17 dBm of input.

In operation, the signal path to the distortion low power detection diode 91 through the bandpass filter stage of the buffer amplifier and output receiver 30 is sampled, digitized and connected to the distortion input 19 of the DSP 16. As in the circuit of FIG. 1, when the DSP 16 sweeps the control voltage input to the VCO 45, the tuning frequencies for each input and output receiver 20, 30 are commonly stamped. During this frequency sweep, the power detected by the carrier power detector diode 65 of the input receiver 20 is applied to a threshold detector 67 where the controlled threshold setting serves to distinguish between carrier and distortion. . As long as the threshold of the set threshold detector 67 is not exceeded, it can be inferred that the output of the receiver 30 represents the distortion power generated by the RF power amplifier 10. This distortion power display is digitized and connected to the processor 16 which uses this information as a principle to control the predistortion correction circuit 14 in the input path to the amplifier 10.

If the output of the carrier power detector 65 exceeds the threshold of the threshold detector 67 (indicating that the output receiver is tuned near the carrier energy), the output of the threshold detector 67 is a second logic. Change to state. This change in status signal is supplied to the blanking input 18 of the DSP 16 to prevent the distortion correction operation performed by the DSP 16 from being affected by the carrier wave.

In the configuration of FIG. 2, the bandwidth of the carrier (as a non-limiting example, may be about 8 MHz) is relatively 'big' relative to the IF passband width (which may be about 250 KHz). This RF amplifier 10 includes a single carrier, single mode (W-CDMA only) amplifier, whose C / I specification is relatively low (eg about 40 dB at the carrier edge). Also, the sweep ratio of the VCO 45 is relatively slow compared to the attenuation of the bandpass filter (relative to a group delay of about 1 msec) (e.g. 20 msec for sweeping from the carrier edge for a few MHz scan position). Required).

As can be seen from the above description of the switchless distortion measurement and correction configuration of the present invention, the unnecessary complexity of using a high isolation switch in the signal flow path of the output receiver in certain applications of the present invention described in the above-referenced '723 application, The controllable blanking isolation switch of the sweep output receiver is effectively removed by switching to a buffer amplifier-passband filter stage. These buffer-filter stages provide additional gain to offset the fact that the signal level extracted from the output amplifier is very low and prevents IMD from occurring in the mixer of the sweep receiver, as well as the same bandpass filter in the sweep input receiver. You can also implement This makes the output receiver's bandwidth slightly narrower than the input receiver and serves to minimize the number of other components used in this circuit. At the input and output receivers, the gain of the buffer amplifier stage and the losses through the bandpass filter are chosen to avoid damaging any other components. Like the configuration of the '723 application, the switchless sweeping oscillator configuration of the present invention enables the RF power amplifier distortion to be accurately measured even in the presence of a multi-frequency input signal.

While certain embodiments have been shown and described in accordance with the present invention, the foregoing is not limited to this, and various changes and modifications as known to those skilled in the art can be made and are not intended to be limited to the details shown and described herein. It is intended to include all such changes and modifications as would be apparent to the intellectuals.

Claims (19)

An RF input port to which an RF input signal is applied; An RF output port for obtaining an amplified RF output signal, An RF power amplifier and an RF distortion connected between the input port and the output port and operable to controllably adjust one or more parameters of the RF signal processing path to compensate for the distortion introduced by the RF power amplifier Including an RF signal processing path having a correction unit, The RF distortion correction unit is connected to obtain information representing the distortion introduced by the RF power amplifier at a predetermined bandwidth, but eliminates the effect of the RF carrier frequency of the RF input signal and includes the amplified RF output signal. A frequency stamped output receiver connected to control the applied energy, and an RF signal processing path connected on a continuous output signal path for supplying information indicative of the distortion introduced by the RF power amplifier at the predetermined bandwidth. RF power amplifier device. The method of claim 1, And an adjustable threshold detector operable to controllably block operation of the RF distortion corrector in accordance with the RF output signal comprising an RF carrier energy exceeding a controllable selectable threshold. Power amplifier device. The method of claim 2, And the RF distortion correcting unit includes an RF predistortion correcting unit for controllably transmitting distortion upstream of the RF input signal of the RF amplifier to remove the distortion given by the RF amplifier. . The method of claim 3, wherein The RF distortion corrector further comprises a frequency sweeping input receiver connected to control the energy contained in the RF input signal, wherein the controllable selectable threshold of the adjustable threshold detector is included in the RF input signal. RF power amplifier device, characterized in that determined according to the energy. The method of claim 1, The RF signal processing path has a sweeping input receiver connected to measure carrier power at the input port, an operating bandwidth less than the operating bandwidth of the sweeping input receiver, and is connected to measure distortion energy at the output port. And a sweeping local oscillator operative to sweep the operating frequencies of the respective input and output receivers in common. The method of claim 5, And an adjustable threshold detector operable to controllably block operation of the RF distortion corrector in accordance with the RF output signal comprising an RF carrier energy exceeding a controllable selectable threshold. Power amplifier device. The method of claim 6, And the controllable selectable threshold of the adjustable threshold detector is determined in accordance with the energy contained in the RF input signal. The method of claim 7, wherein The RF distortion correction unit is connected to an input path of the amplifier and is operable to controllably transfer distortion upstream of the RF input signal of the RF amplifier, the RF according to the distortion energy measured by the sweep output receiver. And a predistortion correction unit for removing the distortion given by the amplifier. The method of claim 5, The sweep output receiver is coupled to an RF predistortion correction processor through a first buffer amplifier and narrowband filter circuitry such that the RF output signal includes RF carrier energy whose operation exceeds the adjustable selectable threshold. RF control amplifier device characterized in that the controllable block by the adjustable threshold detector according to. The method of claim 9, And a signal path from the input port through the post-input receiver to the RF predistortion correction processor is connected through a second buffer amplifier and narrowband filter circuitry. The method of claim 10, And the first and second buffer amplifiers and narrowband filter circuitry use the same narrowband filter circuit component. Varying the operating frequency of the connected output receiver through controlling the energy contained in the amplified RF output signal in a continuous output signal path to obtain information indicative of the distortion introduced by the RF power amplifier at a predetermined bandwidth, (A) extracting therefrom the information indicative of the distortion introduced by the RF power amplifier at the predetermined bandwidth; Controllably adjusting one or more parameters of the RF signal processing path to compensate for the distortion introduced by the RF power amplifier according to the information obtained in step (a) and to remove the RF carrier frequency and performing (b) to connect the RF input signal to the input port and to obtain the amplified RF output signal from the output port. The method of claim 12, Said step (a) controllably blocking derivation of said information indicative of said distortion introduced by said RF power amplifier in accordance with said RF output signal comprising an RF carrier energy above a controllably selectable threshold; Method further comprising a. The method of claim 13, The step (b) is characterized in that for controlling the operation of the RF pre-distortion correction unit installed upstream of the RF input signal of the RF amplifier, to remove the distortion given by the RF amplifier. The method of claim 13, The step (a) is characterized in that for controlling the energy contained in the RF input signal through a frequency sweep input receiver, and setting the controllably selectable threshold value according to the energy included in the RF input signal. Way. The method of claim 13, The step (a) comprises a sweeping input receiver connected to measure carrier power at the input port, an operating bandwidth less than the operating bandwidth of the sweeping input receiver, and connected to measure distortion energy at the output port. And obtaining the information through a postmarked output receiver and controlling the operation of a local oscillator to commonly sweep the operating frequencies of the respective input and output receivers. The method of claim 16, The sweep output receiver is connected to an RF predistortion correction processor through a first buffer amplifier and a narrowband filter circuit arrangement, the operation when performing step (b) exceeding the controllably selectable threshold. And controllably cut off by the adjustable threshold detector in accordance with the RF output signal including RF carrier energy. The method of claim 17, And a signal path from said input port through said sweep input receiver to said RF predistortion correction processor is via a second buffer amplifier and narrowband filter circuit arrangement. The method of claim 18, Wherein said first and second buffer amplifiers and narrowband filter circuitry use the same narrowband filter circuitry.