KR20010022783A - Method and regulator for electrical reluctance machines - Google Patents

Abstract

The present invention relates to a process of estimating the instantaneous resistance of a phase winding in a magnetoresistive machine. The process includes: receiving a signal U _dm indicative of a voltage U _W across at least one phase winding line; Receiving a signal i _Wm indicative of a current i _W passing through the phase winding; Estimating the magnetic flux according to the voltage signal and the current signal; Estimating the instantaneous resistance R _W in the phase winding line according to the phase relationship between the current signal i _{W m} and the magnetic flux signal Ψ _est . The invention also relates to a method for controlling an electric machine and a drive system for carrying out the method.

Description

METHOD AND REGULATOR FOR ELECTRICAL RELUCTANCE MACHINES

The electric machine has two mutually movable parts, typically called "stator" and "rotor", respectively. The most common type of motor has a rotor, which is a hanging member that can rotate inside the stator. The motor is equipped with a coil to which a current can be supplied, thereby generating a magnetic flow. The stator and the rotor form a magnetic circuit through which the magnetic field generated by the coil flows.

As the mutual position between the rotor and the stator changes, the magnetic resistance of the magnetic circuit changes.

In order to drive a magnetoresistive motor with multiple windings, a current is connected to the winding which, to some extent, depends on the position of the rotor with respect to the stator.

Known methods of providing such current control include sensing the position of the rotor using respective position sensors coupled to the rotor, whereby the position sensors generate an output signal according to the position of the rotor.

Another known method of controlling phase current takes advantage of the fact that the phase inductance varies with the position of the rotor relative to the stator. Patent application PCT / SE87 / 00442 discloses a method for determining the rotor position of a magnetoresistive machine, starting from the following equation.

(U-Ri) = d / dt (Li),

Where i is the current through the phase winding, U is the voltage across the phase winding connected in series, the transistor valve and the current sensor resistor, and R is the resistance of the phase winding, the active transistor and the current measurement resistor. A defined constant corresponding to the sum.

The present invention relates to a control method of an electric machine, a drive system and a method of determining the mutual position of moving parts in the electric machine.

1A shows a schematic principle of an electric machine having two mutually movable parts and winding wires.

1B shows the magnetic flow in the machine according to FIG. 1A.

2A shows that in the machine according to FIG. 1A the inductance of the windings varies with the mutual position between the movable parts.

Fig. 2B is a diagram showing the positional dependence of the torque obtained by operating the winding winding.

3 shows a control device with valve bridges connected to the upper winding of the motor according to FIG. 1A.

4 is an equivalent circuit diagram of one of the valve bridges of FIG.

FIG. 5 shows an estimator according to one embodiment of the present invention for estimating the magnetic flow for the upper winding of the motor according to FIG. 1A. FIG.

FIG. 6 shows an example of a period for estimating magnetic flow that may be generated by the estimating apparatus according to FIG. 5; FIG.

FIG. 7 shows a second embodiment of an estimating apparatus for estimating magnetic flow for a phase winding in the motor according to FIG. 1A; FIG.

8 shows another embodiment of the estimating apparatus.

FIG. 9 is a block diagram showing an embodiment of the control device shown in FIG. 3; FIG.

10 is a block diagram of a portion of the control device shown in FIG. 3, in accordance with an embodiment of the invention.

11 is a flowchart illustrating an embodiment of a process of calculating an estimated value of a winding resistance.

The present invention relates to the problem of improving torque adjustment in magnetoresistive machines.

The present invention also relates to the problem of improving torque adjustment of an electric machine without using any sensor that finds the position of the rotor using an electric motor shaft.

More specifically, the present invention relates to the problem of having improved torque adjustment at both high and low rotational speeds without having each rotor position sensor.

Moreover, the present invention relates to the problem of providing torque control of a magnetoresistive motor by estimating phase winding values, such as magnetoresistance values or magnetic flow, to determine the position of the rotor. In addition, the present invention relates to torque control of a motor using the estimated phase winding value, wherein the phase winding values are estimated with improved accuracy, and the dependence on the rotational speed of the motor and the temperature of the motor winding is reduced or Or completely removed.

The magnetoresistive machine includes two mutually movable parts and at least one phase winding with resistance and inductance depending on the mutual position of the parts. The device for adjusting the magnetoresistive machine includes a controllable valve, which can be connected in series with the winding wire and can be adjusted between the original open state and the conduction state. The above mentioned problems are solved by a valve control method comprising the following steps:

a) measuring the current passing through the winding;

b) measuring the voltage across the phase winding;

c) generating a signal value according to the measured current value and the measured voltage value according to an equation comprising an adjustable parameter;

d) determining a relationship value between the signal value and the measured current value;

e) adjusting the parameter value according to the relationship value.

The present invention takes advantage of the fact that there is a magnetic flow in the phase with the winding current for a machine of the type described above, estimates the magnetic flow according to the measured current value and also controls the electrical machine with increased precision.

In one embodiment, a signal value is generated corresponding to the magnetic flow Ψ according to the following equation:

Ψ = U _LW dτ

here

_{U LW = K 3 * U d} + i W · K 1 + K 2

The adjustable parameter value K ₁ relates to the resistance of the phase winding, the parameter K ₂ corresponds to the voltage drop at the valve, and the parameter K ₃ also depends on the actual operating state of the machine.

The solution described above is that the estimate automatically adjusts the control parameter K ₁ so that, for example, as the temperature result changes, the motor's windings significantly agree with the actual magnetic flow even when the series resistance R _W is changed. It provides an advantage. If the voltage U _d is determined in both the winding and the valve, the parameter K ₁ is also determined in consideration of the resistance change that may occur in the valve.

For the sake of simplicity, the invention is described below in connection with a rotating machine. However, the present invention is not limited to a rotor, but the described part also relates to a linear machine capable of moving a moving part made of, for example, a soft magnetic material in a straight line along a linear linear stator having a plurality of stator windings. The same can be applied to other machines.

1A is a schematic diagram illustrating an embodiment of a magnetoresistive motor 10 having a stator 20 and a rotor 30 that can rotate inside the stator.

The stator 20 has three separate windings W _A , W _B and W _C , respectively.

In one embodiment, the rotor is made of soft magnetic material that includes a plurality of projections 40, as shown in FIG. 1A. Soft magnetic materials are ferromagnetic materials that, once magnetized, can be very easily demagnetized, ie require small coercive power to remove the magnetism generated when the material is magnetized. In one variation of the invention, the rotor comprises soft magnetic iron. In addition, in one embodiment, the stator includes a soft magnetic material, such as soft magnetic iron.

When the rotor rotates the central axis, its position changes, which is explained using the angular position [theta] in Fig. 1A. As shown in FIG. 1, the virtual cooperative system has its origin at the center axis of the motor and has two mutually perpendicular axes (x, y). Next, the position of the rotor can be formed according to the angular position θ of the rotor protrusion 40 with respect to the x-axis.

Driving a current through a winding, such as winding W _A , generates a magnetic flow that flows from the stator through the rotor and back to the stator, thereby generating a magnetic circuit. 1B shows a three-phase magneto-resistive motor when driving the current through the winding W _A when the protrusion of the rotor is toward the stator portion in which the magnetic field-generating windings W _A are placed. An example of the magnetic circuit is shown. It should be noted that FIG. 1B merely illustrates the principle of magnetic flow and should not be understood as having to drive a current through a winding when the rotor is in the position shown in FIG. 1B.

By adjusting the current to the windings of the motor, the current must be supplied when the rotor is in position with respect to the stator to make the best use of the torque of the motor.

1A and 1B illustrate a machine with three windings, the rotor having eight protrusions and the stator having twelve protrusions. However, in a preferred embodiment, the rotor has four protrusions, so-called salient poles, and the stator has six protrusions.

2A shows how the inductance changes in the winding W _A depending on the position θ of the rotor.

2B shows the torque that can be obtained at the rotor position θ by operating the winding winding. By comparing the curves, i.e., each solid line of Figs. 2A and 2B showing an inductance (L _WA) and the winding torque (T _WA) obtained by operating the (W _A) of the winding (W _A), the inductance is a positive differential It can be seen that by operating the winding when it has, a positive torque is obtained from the winding W _A.

One embodiment of the control device

3 shows a control device 60 connected to three phase windings W _A , W _B and W _C. The controller 60 includes a power supply 70 for supplying the connection 80 with a DC-voltage having an amplitude of + U _d . 3, the phase winding W _A includes a current sensor S _WA , a power transistor T1 _A , and a power transistor T2 _A between the ground connection 90 and the plus pole 80 of a power supply. It is coupled through a circuit comprising a.

The diode D1 _A is connected to the ground connection 90 through an anode and to the emitter of the transistor T2 _A through a cathode.

The diode D _2A connects the cathode through the current sensor S _D2A to the collector of the power transistor T _2A , and the anode connects to the collector of the power transistor T1 _A. The current sensor S _D2A supplies a signal to the control unit 100 so that the control unit obtains information about whether the current flows through the diode D2 _A.

The other phase windings W _B and W _C respectively couple to current sensors, power transistors, and diodes in the same manner.

The control unit 100 including the microprocessor is adapted to control the switching on and off of the transistor valve. The control unit 100 has six outputs connected to the base of six transistor valves through their respective amplifiers 110. Transistor 110 acts as a valve controller.

In one embodiment, the current sensor S _WA is a Hall sensor that supplies the measured current value i _WAm to the control unit 100. In the same way, the current values measured for the windings W _B and W _C are also supplied to the control unit 100.

The sensor unit 120 is connected to sense the voltage U _d between the anode 80 and the ground connection 90. The sensor unit 120 supplies the measured voltage value U _dm to the control unit 100. In one embodiment, the sensor unit 120 includes a voltage driver having resistors R _X and R _Y connected between the positive voltage connection 80 and the ground connection 90. As shown in FIG. 3, the output of the sensor unit 120 is connected to the point 140 between the resistors R _X and R _Y such that the output signal U _{dm of the} sensor is proportional to the driving voltage U _d . Connect to

FIG. 4 is an equivalent circuit diagram showing the valve bridge for the winding W _A of FIG. 3.

If a current flows through a valve, such as valve T2 _A of FIG. 3, a voltage drop occurs across the valve. The voltage drop across the valve T2 _A is referred to as reference U _T2 in FIG. 4. The valve T2 is shown in FIG. 4 as an equivalent diagram demonstrating that the voltage drop U _T2 depends in part on the internal series resistance R _v and some on the fixed voltage drop U _v . The same applies to the valve T1 _A.

The winding W _A has an impedance which is inductive in nature but also includes a resistive component. In FIG. 4, this is explained by the pure inductance L _W connected in series with the resistor R _W.

The Hall sensor S _WA has a very low impedance that can be neglected in this connection. However, it should be noted that in the analysis below, the internal impedance of the sensor can be treated in the same way as the resistance (R _W ), which, as demonstrated below, allows the estimator to obtain the resistance of the windings and the possible resistance of the current sensor. It means rewarding two things.

The torque T _WA obtained by operating the winding W _A is a function of the current through the winding and the position of the rotor θ:

T _WA = f ₁ (i _WA , θ) (1)

The rotor position θ can be calculated from the magnetic flow Ψ and the phase current i _WA flowing through the corresponding magnetic circuit:

θ = f ₂ (i _WA , Ψ _A ) (2)

This means that the torque T _WA can be expressed as a function of the magnetic flow (Ψ _A ) and the phase current (i _WA ) flowing through the corresponding magnetic circuit:

T _WA = f ₃ (i _WA , Ψ _A ) (3)

As described above, the current i _WA passing through the phase winding is controlled to obtain a desired torque T _W from the active phase winding as shown in FIG. 2B. The torque obtained is a function, in part on the winding current i _W , and part on the magnetic flow. The magnetic flow Ψ in turn depends on the inductance L _W and the current i _W. Thus, T _W depends on i _WA and L _WA :

T _WA = f ₄ (i _WA , L _WA ) (4)

The change in time (dΨ / dt) of the magnetic flow coincides with the instantaneous voltage drop U _LW on the pure inductance L _WA of FIG. 4:

dΨ / dt = U _LW (5)

From Equation (5), we integrate with time to obtain the magnetic flow (Ψ):

Ψ = U _LW dτ (6)

Equation (3) means that if the magnetic flow Ψ can be determined, the desired torque T _W can be generated. Equation (6) means that the magnetic flow Ψ can be determined by determining the voltage U _LW .

In one embodiment of the invention, the magnetic flow Ψ is determined according to equation (6), ie by time integration of the voltage U _LW on the "pure" inductance L _W.

However, since L _W is also an inductance in the winding having the internal resistive voltage drop U _RW , the voltage U _LW cannot be measured. Thus, in one embodiment of the invention, thereby generating a voltage voltage (U ^* _LW) corresponding to (U _LW) via the estimation process. This is explained in more detail below.

4 shows an equivalent diagram for the winding W _A. From this, the following equation is demonstrated:

U _LW = U _W -i _{WAR WA} (7)

From equations (5), (6), and (7),

Ψ = (U _W - i · R _{WA WA)} dτ (8)

Referring to FIG. 4, it can be seen that three optional operating states may occur depending on which and / or which valves T2 and T1 are closed or opened.

I. First operating state

In the first operating state I, both the valve T2 and the valve T1 are closed. The voltage U _LW is then the portion of the voltage across the phase winding that causes magnetic flow, ie resistive loss in the winding. In the first operating state, the voltage U _LW becomes as follows:

U _LW = U _d -i _W (R _W + 2Rv) -2Uv (9)

II. Second operating state

In the second operating state II, both the active valves T1 and T2 are opened so that they interrupt the current, so that current flows in the winding from the ground connection 90 to the positive connection 80. In this case, the voltage U _LW becomes

_{U LW = -U d - i W} · R W - (U D1 + U D2) (10)

III. Third operating state

In the third operating state III, only one of the active valves T1 or T2 is conducted. Assuming the losses are the same for diodes D1 and D2 and for active valves T1 and T2, the voltage U _LW on the "actual inductance" L _W of the phase winding is as follows:

U _LW = i _W (R _W + Rv)-(Uv + U _D1 ) (11)

Use equations (9), (10) and (11) shown above to calculate the voltage (U _LW ), and use equation (6) above to calculate the magnetic flow (Ψ). By using this information and measuring the actual winding current i _W , the torque T _W obtained as a result of operating the winding W can be calculated very accurately according to equation (3).

The control unit determines the actual operating state of the winding W _A using the input signal from the diode current sensor S _D2A and the output signal from the amplifiers 110 _1A and 110 _1B . Each of these three signals has one of the logic values "CONDUCTS" or "DOES NOT CONDUCT". By combining these three signals, the actual operating state can be known.

First embodiment of the estimating apparatus

Starting from the above equation (6) for the magnetic flow (Ψ), knowing that if the current is zero, the magnetic flow generated from the winding of the magnetoresistance is also zero, and the present inventors designed the estimating apparatus 200 shown in FIG. .

FIG. 5 shows an estimating apparatus according to an embodiment of the present invention for estimating the value of the magnetic flow Ψ for the winding W _A starting from the equations (9) to (11) and (6). .

The three equations ((9), (10), and (11)) can be summarized in terms of three parameters (K ₁ , K ₂ , and K ₃ ):

_{U LW = K 3 * U d} + i W · K 1 + K 2 (12)

Here, the parameter values depend on the following operating states:

The following applies to the operating state (I):

K _{1, I} =-(R _W + 2Rv)

K _{2, I} = -2Uv

K _{3, I} = 1

In operating state (II) the following applies:

K _{1, II} =-(R _W )

K _{2, II} =-(U _D1 + U _D2 )

K _{3, II} = -1

In operating state (III) the following applies:

K _{1, III} =-(R _W + Rv)

K _{2, III} =-(U _V + U _D1 )

K _{3, III} = 0

Thus, parameter K ₁ is the actual total loss resistance in the current circuit at each time point, and parameter K ₂ is the voltage drop independent of the actual current in the current circuit. The parameter K ₃ is a variable having an absolute value of 1 and has a positive or negative sign according to an actual operating state.

The estimator 200 includes an input 202 for receiving a logic value from the sensor S _D2A and an input for receiving a state (see FIGS. 3 and 4) for the active valves T _1A and T _2A (see FIG. 3 and FIG. 4). 204 and 206). The inputs 202, 204, and 206 are coupled to a state analyzer 208 that combines input signals to determine an actual operating state and supplies an operating state signal to the output 209.

The estimator 200 also includes an input 210 for the measured value U _dm corresponding to the voltage U _d , and an input 220 for inputting the actual current value i _W.

The input 210 is connected to an addition point 230 via a multiplication unit 232. Multiplication unit 232 has a control input coupled to an operating state signal from analyzer 208. The multiplication unit 232 multiplies the voltage value U _d by the parameter K ₃ , and as a result, supplies the value U _d , -U _d , or 0 to the addition point 230 according to the actual operating state. .

Input 220 is coupled to an input on zero detector 240 and an input on multiplication point 250. The output of the zero detector 240 is coupled to the trigger input 260 of the sample-and-hold circuit 270. The sample hold circuit 270 has an output 280 for a flow error value Ψ _e . Output 280 is connected to parameter adjuster 290. The parameter adjuster 290 generates a variable value C at the output connected to the multiplication unit 300 and the multiplication unit 310.

The output signal of the multiplication unit 300 is the parameter value K1, and the output signal of the multiplication unit 310 is the parameter value K2. Multiplication unit 300 includes a memory unit with parameters G1 _I , G1 _II and G1 _III and has a control input coupled to an operating state signal from analyzer 208. Depending on the operating state, one of G1 _I , G1 _II or G1 _III is multiplied by the variable C, and the result is supplied to the multiplication point 250. The multiplication unit 310 operates in the same way as the multiplication unit 300, but has three parameters G2 _I , G2 _II and G2 _{III in} the memory unit.

The multiplication unit 300 is connected to the multiplication point 250, and the multiplication unit 310 is connected to the addition point 230. The output of multiplication point 250 is connected to the input of addition point 230. The output of the addition point 230 is connected to the integrator 320. An integrator 320 that generates an estimated magnetic flow value Ψ _est is connected to the output 330 of the estimator 200. The output of the integrator is also connected to the sample input 340 of the sample hold circuit 270.

Estimator 200 operates as follows:

An actual value corresponding to the voltage U _d is supplied to the input 210, _and an actual value corresponding to the current i _W is supplied to the input 220. The memory of the multiplication unit 300 has a parameter G1, which is chosen such that the product C * G1 is an approximation of the parameter K ₁ in equation (12):

In case of operating state I, C * G _{1, I} : = K ₁ =-(R _W + 2Rv)

The multiplication unit 310 has a parameter value G2 such that the product C * G2 is an approximation of the voltage drop independent of the current in the active valves T1 and T2:

In case of operating state I, C * G _{2, I} : = K ₂ = -2Uv

Thus, the voltage U _LW estimate is obtained at the output of the addition point 230. By supplying this estimated value to the integrator 320, an estimated value of the magnetic flow Ψ at the output 330 is obtained.

When the current value i _W of the input 220 becomes zero, the zero-detector 240 detects this and supplies a trigger signal to the trigger input 260 of the sample hold circuit. Upon receiving the trigger signal, the sample hold circuit 270 finds the actual magnetic flow estimate. As described above, if the current value is 0, the value should be 0, so the sample value is a representation Ψ _e for the flow estimation error. This error value is supplied to a parameter adjuster 290 which adjusts the variable C (and thus the parameter values K ₁ or K ₂ ) in the direction in which the flow estimation error value is closer to zero. If the estimation error Ψ _e is positive, it is necessary to increase the estimated resistance value C * G1 to reduce the flow estimation error value Ψ _e . If the estimation error Ψ _e has a negative value, the estimated resistance value C * G1 needs to decrease. By repeating this process, the estimation of the resistance value is improved to minimize or eliminate the flow estimation error value Ψ _e .

FIG. 6 illustrates a time change with respect to the estimated value Ψ ^* of the magnetic flow generated by the estimating apparatus 200 according to FIG. 5.

In FIG. 6, it is assumed that the voltage U _d is constant and the current value i _W changes as shown in the figure. The thick line in FIG. 6 shows the estimated magnetic flow when C * G1 coincides with the sum K ₁ of the actual loss resistance, and in FIG. 6, the voltage generated by the addition point 230 in FIG. U _LW ) is shown as an estimate of U ^* _LW .

As can be seen from Fig. 6, when C * G1 = K1, the current value becomes zero and the estimated value of the magnetic flow becomes zero. As described above with reference to FIG. 5, the flow estimation error value Ψ _e becomes zero at that time.

6 also shows the flow estimation value when the absolute value of the product C * G1 is smaller than the absolute value of K ₁ . As shown in FIG. 6, if the absolute value of the product C * G1 is less than the absolute value of K ₁ , the flow estimation value has a positive value Ψ _e when the current i _W passes through zero. do. This means that the flow estimation error Ψ _e output from the sample hold circuit 270 in FIG. 5 can be found in FIG. 6.

The figure also shows the estimated value Ψ ^* for the magnetic flow when the absolute value of the product C * G1 is greater than the absolute value of the actual value K ₁ of the loss resistance.

Referring to FIG. 4, an equivalent view of the active valves T2 and T1 consists of a resistor Rv and a voltage source exhibiting a constant voltage drop Uv in series with the switch. Such equivalent drawings are usually suitable for active valves such as bipolar transistors, thyristors, IGBTs, and metal oxide semiconductor field effect transistors (MOSFETs). In the first three, it is true that the voltage drop (Uv) has a non-zero positive value, but in the MOSFET it is true that the voltage drop (Uv) actually coincides with zero.

In the manual valves D1 and D2 as shown in Fig. 4B, it is true that the original constant voltage drop occurs when the valve is driven forward, that is, when the valve conducts current. Examples of such valves are diodes such as, for example, Schottky-diode. The voltage drop U _D has a positive value with a difference of about 0 to about 0.6 volts.

If the voltage drops Uv and U _D are smaller than the voltage drops [i _W (R _W + Rv]), the unit 310 of FIG. 5 may not be considered by neglecting the Uv and U _D , and thus the unit 300 may be considered. Only the parameter G ₁ can increase the output signal of the parameter adjuster 290.

Second embodiment of the estimating apparatus

A second embodiment of the estimating apparatus for estimating magnetic flow Ψ is shown in FIG. The estimation apparatus 400 shown in FIG. 7 includes an input 210 for a value corresponding to the voltage U _d , an input 220 for a value corresponding to the current i _W , and a magnetic flow estimation. An output 330 to output, an addition point 230 for generating a value corresponding to the estimated value U ^* _LW of the voltage U _LW across the pure inductance, and a magnetic field according to the voltage estimate U ^* _LW . Corresponding to the estimating apparatus shown in FIG. 5 in that it has an integrator 320 for generating flow estimation.

The estimator 400 is different from the estimator shown in FIG. 5 by including a phase error detector 410, where one input of the phase error detector is connected to an input 220 and the other input 430 is an integrator. Connected to the output of 320. The phase error detector has an output for the signal representing the phase relationship between the current signal and the magnetic flow estimation signal. The output signal from the phase error detector 410 may include three logic values according to one embodiment of the present invention; That is, it may have one of Raise, Lower, and Freeze. This can be realized, for example, via a phase error detector supplying a signal having three level values. The output signal of the phase error detector is supplied to the counting port on the counter 440.

Counter 440 also has a counter input 450 that receives a pulse signal with an original fixed pulse frequency from oscillator 460. Counter 440 supplies digital count values through output 470. This coefficient value is an estimated value of the winding resistance R _WA .

The coefficient value output 470 is connected to an adder 472 which also receives the actual parameter value 2Rv, 0 or Rv depending on the actual operating state. The parameter unit 474 has a switch 476 that outputs the memory positions of the valve resistance values for the three operating states and the actual valve parameter value depending on the status signal on the input 478. The state signal is supplied from the state analyzer 208 as described above.

The output signal of the adder 472 corresponds to the parameter K1 in equation (12). The output signal is supplied to an input 480 of a digital-analog converter 490. The D / A converter 490 has a reference voltage input 500 connected to an input 220 that receives an analog value i _W.

The estimating apparatus 400 operates as follows. As described above, in the magnetoresistive motor, it is true that the magnetic flow and the current simultaneously have zero values, which means that the current and the magnetic flow in the magnetoresistive motor are in phase with each other in nature. The phase error detector 410 is used to detect the phase relationship between the actual current i _W and the estimated magnetic flow Ψ _est . The count value generated by the counter 440 corresponds to the winding resistance R _W.

Since the analog signal i _Wm corresponding to the analog current value i _W is supplied to the reference input of the D / A converter, the D / A converter 490 is a function of the current value i _W and the count value K ₁ . Generate a product. Therefore, the output signal of the D / A converter is an analog signal whose amplitude corresponds to the current value multiplied by the parameter K ₁ in equation (12).

The counter 440 together with the oscillator 460 forms an integral circuit, so that the output signal on the output 470 of the counter is a time integration result. In this embodiment, the parameter value K ₁ is continuously adjusted in accordance with the phase relationship, thereby minimizing the phase error, i.e., in the phase state with the actual measured current i _W supplied to the input 220 of the estimating apparatus. Adjust the magnetic flow estimate so that it is. This means that the estimator automatically adjusts the control parameter K ₁ so that the motor winding changes its series resistance R _W and the transistors T1 or T2 change their series resistance as a result of a temperature change, for example. In this case, the magnetic flow estimates also have corresponding gains.

Third embodiment of estimator

The third embodiment of the estimating apparatus corresponds to the estimating apparatus 400 of FIG. 7, but removes the adder 472 and the parameter unit 474, ie, the output 470 of the counter and the input 480 of the D / A converter. To combine directly. Thus, this embodiment of the estimating apparatus is advantageously simply configured. This embodiment involves an approximation ignoring the resistance Rv of the valves. It should be noted from Equation (9) that if the valve resistance 2Rv is much smaller than the winding resistance R _W , an accurate estimate of the magnetic flow is obtained even in the third embodiment. At a predetermined current value and a predetermined magnetic flux value Ψ, the position of the rotor can be determined by analyzing dΨ / dt, that is, U _LW .

However, in the preferred embodiment, the position is determined by combining the magnetic flux estimates for at least two windings.

Fourth Embodiment of Estimation Apparatus

According to a fourth embodiment of the estimating apparatus shown in FIG. 8, the rotor position of the motor with three windings W _A , W _B and W _C is determined. The estimation apparatus, the magnetic flux (Ψ _A) estimate (Ψ _Aest) to obtain a magnetic flux estimation unit (E _A) and a magnetic flux (Ψ _B) estimate (Ψ _Best) another magnetic flux estimation unit to obtain the (E _B of ) And a third flux estimating unit E _C that obtains an estimated value Ψ _Cest of the flux Ψ _C. The position θ can be determined by combining the instantaneous values Ψ _A , Ψ _B , Ψ _C with the magnetic flux values and the instantaneous current values i _WA , i _WB , i _WC .

In some motor driving states, sufficient control accuracy is obtained if the resistance loss Rv at the valve is ignored and the voltage drop at the valve U _D and diode is also ignored. In such a case, as shown in FIG. 8, it is possible to advantageously simply design each magnetic flux estimating unit by ignoring losses Rv and U _D. In Fig. 8, the embodiment signals representing the voltages U _W across the windings are transferred directly to the estimation units. These voltage signals may be obtained using the multiplication unit 232 as described in connection with FIG. 5 or 7 or by measuring. In one embodiment, the measurement of winding voltage U _W obtains a useful measurement by measuring the voltage across the winding and integrating the measured voltage signal with time using a time constant.

Embodiment of the control unit

FIG. 9 is a block diagram showing an embodiment of the control unit 100 shown in FIG. As mentioned above, the control unit 100 has inputs for the winding current and for the voltage U _dm . These signals are fed to the estimators E _C , E _B , E _{A which} operate to measure the instantaneous magnetic flux and the parameters K1, R _W. The winding current value and the measured magnetic flux are fed to a device 520 which determines the position θ therewith. The position value is transmitted to the unit 530 which generates the reference current values I _Aref , I _Bref, I _Cref . The control unit 100 also includes an input 510 for receiving the torque reference value, ie an input for setting the desired motor torque. The unit 530 generates current reference values according to the torque reference value and the position value. The regulator 540 supplies a control pulse to the valve in accordance with the current reference signals I _Aref , I _Bref, I _Cref and the measured current values I _WA , I _WB , I _WC .

10 illustrates a control in accordance with an embodiment including a microprocessor 550 coupled to an input / output port 560, a memory 570, and a programmable logic circuit (PLD) 580. A block diagram of a unit 100 portion. Signals representing the winding currents are passed to a series of multiplication units 490. A bus coupled to the processor 550 supplies the multiply units with updated instantaneous estimates of the respective winding resistors R _WA , R _WB , R _WC . The output of each multiplication unit 490 is a voltage signal representing the product of instantaneous current and winding resistance. Each voltage signal is transferred to a VCO that transforms it into an oscillation signal. The oscillation signals are transmitted to the PLD 580. The PLD 580 receives an oscillation signal representing the voltage U _d and calculates the above-described voltage U _LW from this. PLD 580 also generates a magnetic flux estimate by integrating voltage U _LW over time. Compared with the current measurement signal, real-time magnetic flux estimation error occurs. The PLD 580 delivers the resulting flux error Ψ _err to the processor 550 in real time over the bus 590. The processor operates in accordance with a computer program stored in memory 570. The program includes a routine for estimating resistors R _WA , R _WB , R _WC , which are updated and passed to multiplication unit 490. Multiplication units may operate as described with respect to FIG. The PLD may be operable to calculate the real-time magnetic flux estimation error value Ψ _Aerr by fixing the instantaneous value Ψ _A for the magnetic flux estimation whenever the winding current iA is zero. Therefore, the processor receives the flux error values Ψ _Aerr , Ψ _Berr , Ψ _Cerr obtained when the current is zero for each winding. The processor calculates the correct resistance value according to the above.

A process of generating an estimated value of the winding resistance R _W as implemented by combining a processor and a computer program is described in FIG. 11. The above process is performed only in one winding wire, so that resistance in the one winding wire is obtained. In a preferred embodiment for machine control, the process is carried out for a number of windings. The process may begin with the resistance estimate as a pre-set value (step S600). The magnetoresistive machine is started and when the machine is operated, the voltages U _WA , U _WB , U _WC across each winding are measured or calculated from the voltage U _d (step S610) and the winding current i _WA , i _WB , i _WC ) are measured (step S620), and also integrated as described above in step S630 to obtain a magnetic flux estimate. The real-time estimated value of the magnetic flux is compared with the current measurement value to generate an error signal (step S640). Adjust the resistance estimate according to the error signal as described below.

In one embodiment, when executing a program routine stored in memory 570, microprocessor 550 receives magnetic flux error signals from PLD 580 and is instructed to adjust the resistance estimate according to the sign and value of the error signal. . If the estimation error Ψ _e is positive, the program instructs the processor to increase the estimated resistance value. If the estimated error Ψ _e has a negative value, the program instructs the processor to decrease the measured resistance value. PLD 580 actually generates a new magnetic flux value according to the adjusted resistance estimate. A new error signal is generated according to the magnetic flux value when the current in the winding is 0, and the error signal is supplied to the processor 550. This process is repeated to improve the estimation of the resistance value and to minimize or eliminate the flow estimate error value Ψ _e .

In another embodiment, the PLD also _{delivers the} measured current signals i _WAm , i _WBm , i _WCm and digital signals corresponding to U _d via the bus 590 to the processor 550. When executing the program routine stored in the memory 570, the microprocessor 550 receives the signal U _dm representing the voltage U _WA across the associated winding wire W _A , and also the associated winding wire ( Is _commanded to receive a signal i _WAm representing the current i _WA passing through W _A ). Thus, the program in the memory 570 instructs the microprocessor to estimate the magnetic flux according to the voltage signal and the current signal. This may be obtained by performing numerical integration. The program instructs the processor to generate an error signal by comparing the instantaneous current measurement signal with the measured magnetic flux. The microprocessor 550 is then instructed to adjust the resistance estimate in accordance with the sign and value of the error signal as described above.

In one embodiment, in cooperation with the program, the processor also calculates the position of the rotor to generate control signals for the valve.

The program may be embedded in the memory 570 by supplying it through the I / O pod 560. Alternatively, the circuit shown in FIG. 10 includes a socket that can be safely accessed by a recording medium such as a nonvolatile memory (PROM). The recording medium includes machine readable instructions for instructing the processor as described above.

The present invention is advantageous in that it is possible to improve the current state of the controller for magnetoresistance of the prior art by embedding a program for performing the resistance estimation process. Thus, improved magnetic flux estimates are obtained, and torque adjustment is obtained at both high and low rotational speeds.

Temperature measurement

The actual resistance of the winding has a temperature dependence on the electrically conductive material used in the winding. Thus, the measurement of the winding resistance can be used to represent the instantaneous average temperature in the winding. Knowing the winding resistance at the predetermined temperature and the temperature coefficient of the winding material, the temperature can be calculated from the instantaneous resistance value R _W.

Speed-Torque Independent Adjustment

By adjusting the active valves T1 _A , T2 _A , T1 _B , T2 _B , T1 _C , T2 _C (see Fig. 3) using any of the above-mentioned estimation devices, the torque and rotational speed of the Accurate adjustment can reduce energy loss because the valve can be operated to allow current to flow through the windings W _A , W _B , W _C when the torque yield is greatest. Further, the adjustment according to the invention means that the machine can be controlled with improved accuracy independent of the original rotational speed. It can be seen from the following that the dependence on the rotational speed is reduced:

As described above with respect to FIG. 2A, driving the magnetoresistive machine changes the inductance L _W depending on the position θ. Magnetic flow (Ψ) changes with inductance and current. If the magnetic circuit is not saturated, the relationship is as follows:

Ψ = L _W (θ) * i _W (13)

Differentiating with respect to time gives:

dΨ / dt = L _W (θ) * di _W / dt + dL _W (θ) / dθ * dθ / dt * i _W (14)

From the above formulas (5) and (7), we can know:

U _W = D _Ψ / dt + R _W * i _W (15)

Substituting equation (14) into equation (15) gives:

U _W = L _W (θ) * di _W / dt + dL _W (θ) / dθ * dθ / dt * i _W + R _W * i _W (16)

Where dθ / dt is the rotor angular frequency, ie the speed of rotation of the machine.

The second term [dL _W ([theta]) / d [theta] * d [theta] / dt * i _W ] of Equation (16) causes the torque of the machine, and the third term (R _W * i _W ) is a complete loss. Since the second term is proportional to the rotation speed, it can be seen that a large amount of winding voltage U _W induces a power loss at a low rotation speed.

Similarly, the winding resistance (R _W) the specified constant (R _C) that, assuming and adjusting the reluctance machine, the actual winding resistance (R _W) is defined constant (R _C) corresponding to the sum of the resistance in the phase winding It can be seen from equation (16) that the deviation is greatest at low rotational speeds. Since the winding resistance is temperature dependent, the actual winding resistance R _W changes while the machine is running. The estimating apparatus according to the present invention has the advantage that the machine adjustment can be adapted to the actual winding resistance of the machine, so that the machine can be adjusted with excellent accuracy at all rotational speeds and a wide temperature range.

Claims (21)

Apparatus 100 for controlling a magnetoresistive machine having at least one phase winding W _A , W _B , W _C having two mutually movable parts and an inductance L _W depending on the mutual position of the parts In this case, the adjusting device 100 includes controllable valves T1 and T2 which can be adjusted between the original open state and the conduction state, whereby the valve is connected in series with the winding wire, and the winding winding is connected to the resistor R _W. The method for controlling the valve has: a) measuring a current i _W passing through the phase winding; b) measuring a voltage (U _d ; U _W ) across the winding wire and the valve; c) measured current value and measured voltage, according to an equation comprising an adjustable parameter K1, R ^* _W depending on the resistance R _{W of the} winding, multiplied by the measured current value i _Wm . _Generating a signal value U _LWest , Ψ _est depending on the value; d) determining a relationship value between the signal value U _LWest , Ψ _est and the measured current value i _W ; e) adjusting the parameter value K1 according to the relationship value; And f) adjusting the valves (T1, T2) according to the signal values (U _LWest , Ψ _est ). The method according to claim 1, wherein step b) is modified in that it is replaced by a step of estimating a constant voltage U _d across the upper winding and the valve. The method of claim 1 or 2, wherein: At least steps a) and c) after adjusting the parameter value K1 such that the generated signal value U _LWest , Ψ _est is an estimated value U _LW , Ψ depending on the current value and inductance L _W. Repeating; _Adjusting the valves (T1, T2) according to the signal values (U _LWest , Ψ _est ) to obtain the desired torques (T _WA , T _WB , T _WC ). In a magnetoresistive machine having first and second parts which are parts which are movable relative to one another, the first part 30 comprises a soft magnetic material and the second part 20 depends on the mutual position of the parts. And at least one phase winding (W _A , W _B , W _C ) having an inductance (L _W ), the method of determining the instantaneous mutual position of the parts: a) measuring a current i _W passing through the phase winding; b) measuring the voltage U _d across the phase winding; c) in dependence on the measured current value and the measured voltage value according to the equation comprising an adjustable parameter K1 dependent on the resistance R _{W of the} phase winding, multiplied by the measured current value i _Wm . _{Generating a} signal value U _LWest , Ψ _est ; d) determining a difference or relationship between the signal value U _LWest , Ψ _est and the measured current value i _W ; e) adjusting the parameter value K1 according to the difference value or the relationship value; f) determining the instantaneous mutual position of the parts according to the signal value U _LWest , Ψ _est . The method of claim 4, wherein After adjusting the parameter value K1 such that the generated signal values U _LWest , Ψ _est are magnitude estimates U _LW , Ψ depending on the current value and inductance L _W , at least steps a) and c Repeating c); And Determining the instantaneous mutual position of the parts according to the signal value (U _LWest , Ψ _est ). 6. The method according to claim 4 or 5, further comprising repeating at least step d) after adjusting the parameter value (K1). 7. A method according to claim 4, 5 or 6, characterized in that adjusting the parameter value K1 affects the actually generated signal value U _LWest , Ψ _est so that the absolute value of the difference value is minimized. How to. 8. The method according to any one of claims 4 to 7, wherein step e) increases the parameter value K1 if the difference value has a first sign (+) and the parameter value if the difference value has a second sign (-). Decreasing the value K1. The method according to any one of claims 1 to 8, wherein step d) determines a phase difference between the signal value U _LWest , _est and the measured current value i _W , or the measured current value i _W ) determining the amplitude of the signal value U _LWest , Ψ _est at the correct amplitude level of _W ). The first portion includes soft magnetic material and the second portion includes at least one winding wire having inductance L _W depending on the mutual position of the portions, the first portion and the second being movable portions with respect to each other. In a drive system for adjusting an electric machine having parts: Controllable valves (T1, T2) which can be adjusted between the original open state and the conduction state; A first winding (80) and a second wiring (90) having a resistance winding (R _W , Rv) connected in series with the valve in series with the connections; A control unit 100 for controlling the valve during mutual movement between parts of the electric motor; Means for measuring a current i _W passing through the phase winding; Means for measuring a voltage U _d across the phase winding; Signal value dependent on the measured current value and the measured voltage value, according to the equation comprising an adjustable parameter K1 dependent on the resistance R _{W of the} phase winding, multiplied by the measured current value i _Wm . Means for generating (U _LWest , Ψ _est ); Means for determining a relationship between the signal value (U _LWest , Ψ _est ) and the measured current value (i _W ); Means for adjusting a parameter value K1 depending on the relationship value; And The control unit comprises means for controlling valves (T1, T2) depending on the signal values (U _LWest , Ψ _est ) such that a desired torque (T _W ) is obtained. In a magnetoresistive machine having first and second parts which are movable parts with respect to each other, the second part comprises at least one having an inductance L _W and a resistance R _W depending on the mutual position of the parts. In a method comprising determining a winding line (W _A , W _B , W _C ), the instantaneous mutual position of the parts: a) generating a signal i _WAm corresponding to the current i _WA passing through the phase winding W _A ; b) generating a signal corresponding to the voltage U _WA across the phase winding W _A ; c) generating 440 a parameter value K1 _A that actually corresponds to the resistance loss R _WA in the winding wire W _A ; d) generating an amplitude signal U _LWAest , Ψ _Aest depending on the current signal, the voltage signal and the parameter value K1 _A ; e) adjusting the parameter value K1 _A according to the relationship between the current value i _Wm and the amplitude signal U _LWAest , Ψ _Aest ; e) determining a position value θ depending on the amplitude signal U _LWest , Ψ _est . The method of claim 11 wherein: Generating a second current signal i _WBm corresponding to the current i _B passing through the second phase winding W _B ; Generating a second voltage signal corresponding to the voltage U _WB across the second phase winding W _B ; Generating (440) a second parameter value K1 _B that actually corresponds to a loss resistance R _WB in the second phase winding W _B ; _Generating an amplitude signal (U _LWBest , Ψ _Best ) depending on the second current signal, the second voltage signal, and the second parameter value K1 _B ; _Adjusting a second parameter value K1 _B depending on the relationship between the second current signal i _WBm and the second amplitude signal U _LWBest , Ψ _Best ; And determining a position value (θ) dependent on the first amplitude signal (U _LWAest , Ψ _Aest ) and the second amplitude signal (U _LWBest , Ψ _Best ). The first and second portions are movable relative to each other, and the second portion 20 has at least one phase winding W _A , W _B , W _C having a resistance R _W and an inductance L _W. In the estimation device for determining the mutual position of the first portion 30 and the second portion 20 in the magnetoresistance machine, a) an input 220 for receiving a signal i _Wm indicative of the current i _W passing through the phase winding; b) an input 210 for receiving a signal U _dm indicative of the voltage U _W across the winding wire; c) means (410, 440, 460) for generating a parameter value (R ^* _W , K1) that actually corresponds to the resistance loss (R _W ) at the winding; d) means _500,320 for generating an amplitude signal Ψ _est in accordance with the current signal i _Wm , the voltage signal U _dm and the parameter values R ^* _W , K1; And e) means (500, 320) for generating a position value [theta] depending on the signal value (U _LWest , Ψ _est ); f) The parameter value generator controls means 410, 240, 270 for adjusting the parameter values R ^* _W , K1 according to the phase relationship between the current signal i _Wm and the amplitude signal U _LWest , Ψ _est . Estimation apparatus comprising a. In estimating the instantaneous resistance of the winding wire in the magnetoresistance: Receiving a signal U _dm indicative of a voltage U _W across at least one phase winding line; Receiving a signal i _Wm indicative of a current i _W passing through the phase winding; Estimating magnetic flux according to the voltage signal and the current signal; Estimating the instantaneous resistance (R _W ) in the phase winding according to the phase relationship between the current signal (i _Wm ) and the magnetic flux signal (Ψ _est ). 15. The process of claim 14, further comprising repeating magnetic flux estimation using the estimated instantaneous resistance (R _W ). 16. The method according to claim 14 or 15, further comprising the step of determining the mutual position between the first part and the second part of the magnetoresistive machine according to the current signal i _Wm and the magnetic flux estimation value Ψ _est . Characterized in that the process. The process according to claim 14, 15 or 16, further comprising the step of determining the current reference values i _refA , i _refB , i _refC according to the torque reference value and the magnetic flux estimation value Ψ _est . . In a device for estimating the instantaneous resistance of a winding wire in a magnetoresistive machine: An input for receiving a signal U _dm indicative of a voltage U _W across at least one phase winding line; An input for receiving a signal i _Wm indicative of a current i _W passing through the phase winding; A microprocessor; And A memory having a computer program instructing the microprocessor to perform the process of estimating the instantaneous resistance; 18. A device, characterized in that a microprocessor is coupled to the memory and the signal inputs such that the device performs a procedure according to any of claims 14 to 17 while executing a program. An input for receiving a signal indicative of a phase winding current; An input for receiving a signal U _dm indicative of a voltage U _W across at least one phase winding line; In a computer program product for use with a device for estimating the instantaneous resistance of a winding in a magnetoresistive machine, including a microprocessor and a memory: A recording medium; Means recorded on a recording medium for instructing the microprocessor to receive a signal U _dm indicative of a voltage U _W across at least one phase winding line; Means recorded on the recording medium instructing the microprocessor to receive a signal i _Wm indicative of a current i _W passing through the at least one winding line; Means recorded on the recording medium for instructing the microprocessor to estimate the magnetic flux according to the voltage signal and the current signal; _Means for instructing the microprocessor to estimate the instantaneous resistance (R _W ) of the phase winding in accordance with the phase relationship between the current signal (i _Wm ) and the magnetic flux signal (U _LWest , Ψ _est ). Computer program products made. An input for receiving a signal indicative of a phase winding current; An input for receiving a signal U _dm indicative of a voltage U _W across at least one phase winding line; A computer program for use with a device for generating a magnetic flux estimation error signal (Ψ _error ) in real time, and a device for estimating the instantaneous resistance of a winding wire in a magnetoresistive machine, including a microprocessor 550 and a memory 570. In the computer program product: A recording medium; Means recorded on the recording medium for instructing the microprocessor to receive a signal indicative of a magnetic flux estimation error (Ψ _error ) for at least one phase winding line; Repeatedly corrected in accordance with the magnetic flux error signal, in that it comprises at least one resistance estimate to produce a ^{_{(R * WA, R * WB}} , R * WC, CG 1) that instructs the microprocessor, the means for recording on a recording medium Computer program product characterized. In a device 230, 320, 580 for generating a magnetic flux estimation error signal Ψ _{error in} real time for use in a device for estimating the instantaneous resistance of a phase winding in a magnetoresistive machine: An input 220 for receiving a signal i _Wm representing a current i _W passing through the phase winding; An input 210 for receiving a signal U _dm indicative of the voltage U _W across the phase winding; Means (410, 440, 460) for generating a parameter value (R ^* _W , K1) that actually corresponds to the resistance loss (R _W ) at the winding; Means 550, 320, 490, 580 for generating an amplitude signal Ψ _est in accordance with the current signal i _Wm , the voltage signal U _dm and the parameter values R ^* _W , K1; And Means (580) for generating an error signal (Ψ _error ) in accordance with the phase relationship between the current signal (i _Wm ) and the amplitude signal (U _LWest , Ψ _est ).