KR102477037B1 - Method and apparatus for channel estimation in a wireless communication system - Google Patents

Abstract

A method and apparatus for performing channel estimation by a terminal in a wireless communication system. According to the present invention, a plurality of pilot signals for channel estimation are received from a base station through a plurality of beams, time synchronization of at least one pilot signal among the plurality of pilot signals is obtained, and the at least one pilot signal is obtained. Aligning in time, calculating a propagation delay of each of the aligned at least one pilot signal, and performing channel estimation based on the at least one pilot signal for which the calculated propagation delay is compensated. device can be provided.

Description

Method and apparatus for channel estimation in a wireless communication system

The present invention relates to channel estimation in a wireless communication system, and more particularly, to a method and apparatus for estimating a channel by receiving a pilot signal in a wireless communication system.

Mobile communication systems have been developed to provide voice services while ensuring user activity. However, the mobile communication system has expanded its scope to data services as well as voice. Currently, the explosive increase in traffic causes a shortage of resources and users demand higher-speed services, so a more advanced mobile communication system is required. have.

The requirements of the next-generation mobile communication system are to support explosive data traffic, drastic increase in transmission rate per user, significantly increased number of connected devices, very low end-to-end latency, and high energy efficiency. should be able to To this end, Dual Connectivity, Massive MIMO (Massive Multiple Input Multiple Output), In-band Full Duplex, Non-Orthogonal Multiple Access (NOMA), Super Wideband Wideband) support, various technologies such as device networking (Device Networking) are being studied.

An object of the present invention is to provide a method and apparatus for estimating a channel by receiving a pilot signal.

In addition, an object of the present invention is to provide a method and apparatus for estimating a channel through a pilot signal in a process of searching for an asynchronous cell in a mmWave environment.

In addition, an object of the present invention is to provide a method for receiving a pilot signal when a pilot signal for channel estimation is transmitted through multiple directional beams.

In addition, an object of the present invention is to provide a method for compensating for path delays of pilot signals transmitted through various directional beams.

Another object of the present invention is to provide a method for estimating a channel using a plurality of pilot signals with path delays compensated for.

Another object of the present invention is to provide a method for estimating a channel by removing signal characteristics from a plurality of received pilot signals.

Another object of the present invention is to provide a method for transmitting data through different beams in order to reduce data loss due to obstacles on a communication path in a hybrid MIMO environment using mmWave.

The technical problems to be achieved in this specification are not limited to the technical problems mentioned above, and other technical problems not mentioned will be clearly understood by those skilled in the art from the description below. You will be able to.

Specifically, a method for performing channel estimation by a terminal according to an embodiment of the present invention includes receiving a plurality of pilot signals for channel estimation through a plurality of beams from a base station; obtaining time synchronization of at least one pilot signal among the plurality of pilot signals; aligning the at least one pilot signal in time; calculating a propagation delay of each of the aligned at least one pilot signal; and performing channel estimation based on the at least one pilot signal for which the calculated propagation delay is compensated for.

In the present invention, the aligning may include calculating a delay between the at least one pilot signal; and aligning the at least one pilot signal in time according to a transmission order in the base station based on the calculated delay.

Also, in the present invention, the delay includes the number of OFDM symbols and transmission delay between the at least one pilot signal.

In addition, the present invention comprises the steps of comparing the plurality of pilot signals and a threshold value; and determining the at least one pilot signal greater than the threshold among the plurality of pilot signals.

Also, in the present invention, performing channel estimation may include calculating a channel state value of each of the at least one pilot signal; and selecting the highest channel state value based on the calculated channel state value.

In the present invention, the calculating of the channel state value may include obtaining a first channel state value based on a first signal characteristic value of each of the at least one pilot signal; and obtaining a second channel state value from the first channel state value using a specific algorithm.

In the present invention, the first signal characteristic is a value representing the digital characteristic of the pilot signal transmitted from the base station, and the first channel state value is obtained by removing the first characteristic value from each of the at least one pilot signal. is obtained

Also, in the present invention, the specific algorithm is an orthogonal matching pursuit (OMP) algorithm.

In addition, the present invention further includes receiving configuration information of the at least one pilot signal from the base station.

In addition, the present invention, a radio unit for transmitting and receiving radio signals with the outside (Radio Frequency Unit); and a processor functionally coupled to the communication unit, wherein the processor receives a plurality of pilot signals for channel estimation through a plurality of beams from a base station, and transmits at least one pilot signal among the plurality of pilot signals. Acquiring time synchronization, aligning the at least one pilot signal in time, calculating a propagation delay of each of the aligned at least one pilot signal, and assigning the calculated propagation delay to the compensated at least one pilot signal. Provides a terminal that performs channel estimation based on

The present invention has an effect of estimating a channel through a pilot signal in a process of searching for an asynchronous cell in a mmWave environment.

In addition, the present invention has an effect of estimating a channel through a plurality of pilot signals transmitted through different directional beams.

In addition, the present invention has an effect of designing beamforming based on channels estimated using pilot signals transmitted through different directional beams.

In addition, when a plurality of pilot signals for channel estimation are transmitted in multiple directional beams, the present invention estimates a channel through a received pilot signal by compensating for a path delay of each of the plurality of transmitted pilot signals, thereby omni-directional beams. It has the same effect as estimating a channel through pilot signals transmitted through .

In addition, the present invention has an effect of effectively performing channel estimation by estimating a channel by removing specific signal characteristics from a plurality of received pilot signals.

In addition, the present invention has an effect of reducing data loss due to obstacles on a communication path by transmitting data through different beams.

The effects obtainable in the present specification are not limited to the effects mentioned above, and other effects not mentioned can be clearly understood by those skilled in the art from the description below. will be.

BRIEF DESCRIPTION OF THE DRAWINGS The accompanying drawings, which are included as part of the detailed description to aid understanding of the present invention, provide examples of the present invention and, together with the detailed description, describe the technical features of the present invention.
1 is a diagram illustrating an example of an evolved packet system (EPS) related to an LTE system to which the present invention can be applied.
2 is a block diagram illustrating an example of a radio protocol architecture to which technical features of the present invention may be applied.
3 is a diagram for explaining physical channels used in a 3GPP LTE/LTE-A system to which the present invention can be applied and a general signal transmission method using them.
4 shows the structure of a radio frame in a radio communication system to which the present invention can be applied.
5 is a diagram illustrating a resource grid for one downlink slot in a wireless communication system to which the present invention can be applied.
6 shows a structure of a downlink subframe in a wireless communication system to which the present invention can be applied.
7 shows a structure of an uplink subframe in a wireless communication system to which the present invention can be applied.
8 shows an example of a radio frame structure for transmission of a synchronization signal in a wireless communication system to which the present invention can be applied.
9 shows an example of a structure of a secondary synchronization signal in a wireless communication system to which the present invention can be applied.
10 and 11 are diagrams illustrating an example of a method of receiving a pilot signal for channel estimation using a directional beam to which the present invention can be applied.
12 is a diagram illustrating an example of a structure of a frame and a pilot signal proposed in the present invention.
13 is a flowchart illustrating an example of a method for estimating a channel using a pilot signal proposed in the present invention.
14 is a diagram illustrating an example of a method for obtaining time synchronization of a pilot signal proposed in the present invention.
15 to 17 are diagrams illustrating an example for receiving a pilot signal by compensating for time synchronization according to a path delay of a pilot signal proposed in the present invention.
18 is a diagram illustrating an example of an IDFT (Inverse Discrete Fourier Transform) value of a pilot signal.
19 is a diagram illustrating an example of a method for estimating a channel using a pilot signal to which the present invention can be applied.
20 to 22 are diagrams illustrating an example of a method for estimating a channel using a pilot signal proposed in the present invention.
23 is a diagram illustrating an example of a case in which a communication failure occurs due to an obstacle in a MIMO system using a directional beam.
24 is a diagram illustrating an example of a double space time space diversity (DSTTD) system of multi-input multi-output (MIMO) to which the present invention can be applied.
25 is a diagram illustrating an example of a DSTTD system to which an antenna shuffling technique to which the present invention can be applied is applied.
26 and 27 are diagrams illustrating an example of a DSTTD system to which a beam shuffling technique proposed in the present invention is applied.
28 is a flowchart illustrating an example of a method for estimating a channel using a pilot signal proposed in the present invention.
29 is a diagram illustrating an example of an internal block diagram of a wireless device to which the present invention can be applied.

Hereinafter, preferred embodiments according to the present invention will be described in detail with reference to the accompanying drawings. The detailed description set forth below in conjunction with the accompanying drawings is intended to describe exemplary embodiments of the present invention and is not intended to represent the only embodiments in which the present invention may be practiced. The following detailed description includes specific details for the purpose of providing a thorough understanding of the present invention. However, one skilled in the art recognizes that the present invention may be practiced without these specific details.

In some cases, in order to avoid obscuring the concept of the present invention, well-known structures and devices may be omitted or may be shown in block diagram form centering on core functions of each structure and device.

In this specification, a base station has a meaning as a terminal node of a network that directly communicates with a terminal. A specific operation described as being performed by a base station in this document may be performed by an upper node of the base station in some cases. That is, it is obvious that various operations performed for communication with a terminal in a network composed of a plurality of network nodes including a base station may be performed by the base station or other network nodes other than the base station. 'Base Station (BS)' may be replaced by terms such as fixed station, Node B, evolved-NodeB (eNB), base transceiver system (BTS), and access point (AP). . In addition, a 'terminal' may be fixed or mobile, and may include user equipment (UE), mobile station (MS), user terminal (UT), mobile subscriber station (MSS), subscriber station (SS), and AMS ( Advanced Mobile Station), wireless terminal (WT), machine-type communication (MTC) device, machine-to-machine (M2M) device, device-to-device (D2D) device, and the like.

Hereinafter, downlink (DL) means communication from a base station to a terminal, and uplink (UL) means communication from a terminal to a base station. In downlink, a transmitter may be part of a base station and a receiver may be part of a terminal. In uplink, a transmitter may be a part of a terminal and a receiver may be a part of a base station.

Specific terms used in the following description are provided to aid understanding of the present invention, and the use of these specific terms may be changed into other forms without departing from the technical spirit of the present invention.

The following technologies are CDMA (code division multiple access), FDMA (frequency division multiple access), TDMA (time division multiple access), OFDMA (orthogonal frequency division multiple access), SC-FDMA (single carrier frequency division multiple access), NOMA It can be used in various wireless access systems such as (non-orthogonal multiple access). CDMA may be implemented as a radio technology such as universal terrestrial radio access (UTRA) or CDMA2000. TDMA may be implemented with a radio technology such as global system for mobile communications (GSM)/general packet radio service (GPRS)/enhanced data rates for GSM evolution (EDGE). OFDMA may be implemented with radio technologies such as IEEE 802.11 (Wi-Fi), IEEE 802.16 (WiMAX), IEEE 802-20, and evolved UTRA (E-UTRA). UTRA is part of the universal mobile telecommunications system (UMTS). 3rd generation partnership project (3GPP) long term evolution (LTE) is a part of evolved UMTS (E-UMTS) that uses E-UTRA and adopts OFDMA in downlink and SC-FDMA in uplink. LTE-A (advanced) is an evolution of 3GPP LTE.

Embodiments of the present invention may be supported by standard documents disclosed in at least one of IEEE 802, 3GPP, and 3GPP2, which are wireless access systems. That is, steps or parts not described to clearly reveal the technical spirit of the present invention among the embodiments of the present invention may be supported by the above documents. In addition, all terms disclosed in this document can be explained by the standard document.

For clarity of explanation, 3GPP LTE/LTE-A is mainly described, but the technical features of the present invention are not limited thereto. Hereinafter, preferred embodiments according to the present invention will be described in detail with reference to the accompanying drawings. The detailed description set forth below in conjunction with the accompanying drawings is intended to describe exemplary embodiments of the present invention and is not intended to represent the only embodiments in which the present invention may be practiced. The following detailed description includes specific details for the purpose of providing a thorough understanding of the present invention. However, one skilled in the art recognizes that the present invention may be practiced without these specific details.

In some cases, in order to avoid obscuring the concept of the present invention, well-known structures and devices may be omitted or may be shown in block diagram form centering on core functions of each structure and device.

Messages, frames, signals, fields, and devices described in this specification are for explaining the present invention, and are not limited to their respective names, and may be replaced with other messages, frames, signals, fields, and devices that perform the same function. .

In this specification, a base station has a meaning as a terminal node of a network that directly communicates with a terminal. A specific operation described as being performed by a base station in this document may be performed by an upper node of the base station in some cases. That is, it is obvious that various operations performed for communication with a terminal in a network composed of a plurality of network nodes including a base station may be performed by the base station or other network nodes other than the base station. A 'base station (BS)' includes a fixed station, Node B, evolved-NodeB (eNB), base transceiver system (BTS), access point (AP), macro eNB (MeNB), SeNB ( Secondary eNB) and the like.

In addition, a 'terminal' may be fixed or mobile, and may include user equipment (UE), mobile station (MS), user terminal (UT), mobile subscriber station (MSS), subscriber station (SS), and AMS ( Advanced Mobile Station), wireless terminal (WT), machine-type communication (MTC) device, machine-to-machine (M2M) device, device-to-device (D2D) device, and the like.

Hereinafter, downlink (DL) means communication from a base station to a terminal, and uplink (UL) means communication from a terminal to a base station. In downlink, a transmitter may be part of a base station and a receiver may be part of a terminal. In uplink, a transmitter may be a part of a terminal and a receiver may be a part of a base station.

Specific terms used in the following description are provided to aid understanding of the present invention, and the use of these specific terms may be changed into other forms without departing from the technical spirit of the present invention.

The following technologies are CDMA (code division multiple access), FDMA (frequency division multiple access), TDMA (time division multiple access), OFDMA (orthogonal frequency division multiple access), SC-FDMA (single carrier frequency division multiple access), NOMA It can be used in various wireless access systems such as (non-orthogonal multiple access). CDMA may be implemented as a radio technology such as universal terrestrial radio access (UTRA) or CDMA2000. TDMA may be implemented with a radio technology such as global system for mobile communications (GSM)/general packet radio service (GPRS)/enhanced data rates for GSM evolution (EDGE). OFDMA may be implemented with radio technologies such as IEEE 802.11 (Wi-Fi), IEEE 802.16 (WiMAX), IEEE 802-20, and evolved UTRA (E-UTRA). UTRA is part of the universal mobile telecommunications system (UMTS). 3rd generation partnership project (3GPP) long term evolution (LTE) is a part of evolved UMTS (E-UMTS) that uses E-UTRA and adopts OFDMA in downlink and SC-FDMA in uplink. LTE-A (advanced) is an evolution of 3GPP LTE.

1 is a diagram illustrating an example of an evolved packet system (EPS) related to an LTE system to which the present invention can be applied.

The LTE system aims to provide seamless IP connectivity (Internet Protocol connectivity) between a user equipment (UE) and a pack data network (PDN) without interfering with end-user applications while the user is on the move. . The LTE system completes the evolution of radio access through an Evolved Universal Terrestrial Radio Access Network (E-UTRAN) that defines a radio protocol architecture between a user terminal and a base station, which uses an Evolved Packet Core (EPC) network. It is also achieved through evolution in the non-wireless aspect by System Architecture Evolution (SAE), which includes LTE and SAE include Evolved Packet System (EPS).

EPS uses the concept of EPS bearers to route IP traffic from a gateway to a user terminal within a PDN. A bearer is an IP packet flow having a specific Quality of Service (QoS) between the gateway and a user terminal. E-UTRAN and EPC together establish or release bearers requested by applications.

The EPC is also called a CN (core network) and controls the UE and manages bearer setup.

As shown in FIG. 1, the node (logical or physical node) of the EPC of the SAE is a Mobility Management Entity (MME) 30, a PDN-GW or a PDN gateway (P-GW) 50, an S-GW ( Serving Gateway (40), PCRF (Policy and Charging Rules Function) (60), HSS (Home subscriber Server) (70), and the like.

The MME 30 is a control node that processes signaling between the UE 10 and the CN. A protocol exchanged between the UE 10 and the CN is known as NAS (Non-Access Stratum) protocol. An example of the functions supported by the MME 30 are functions related to bearer management manipulated by the session management layer in the NAS protocol, including setting, management, and release of bearers, network and Including establishment of connection and security between UEs 10, it is manipulated by a connection layer or a mobility management layer in the NAS protocol layer.

In the present invention, the MME 30 is an entity implemented with functions necessary for processing authentication and context information for a terminal, and has been described as one embodiment. Accordingly, other devices as well as the MME 30 may perform the corresponding function.

The S-GW 40 serves as a local mobility anchor for data bearers when the UE 10 moves between base stations 20 (eNodeB). All user IP packets are transmitted through the S-GW 40. In addition, the S-GW 40 determines that the UE 10 is in an idle state known as the ECM-IDLE state, and the MME 30 controls paging of the UE 10 to re-establish a bearer. When temporarily buffering downlink data during initiation, bearer-related information is maintained. In addition, it serves as a mobility anchor for inter-working with other 3GPP technologies such as GRPS (General Packet Radio Service) and UMTS (Universal Mobile Telecommunications System).

In the present invention, the S-GW 40 is an entity implemented with functions necessary for processing user data routing/forwarding, and has been described as an embodiment. Accordingly, other devices as well as the S-GW 40 can perform the corresponding function.

The P-GW 50 performs IP address allocation for the UE and performs QoS enforcement and flow-based charging according to rules from the PCRF 60. The P-GW 50 performs QoS enforcement for Guaranteed Bit Rate (GBR) bearers. It also serves as a mobility anchor for interworking with non-3GPP technologies such as CDMA2000 or WiMAX networks.

In the present invention, the P-GW 50 is an entity implemented with functions necessary for processing user data routing/forwarding, and has been described as an embodiment. Accordingly, other devices as well as the P-GW 50 can perform the corresponding function.

The PCRF 60 performs policy control decision-making and performs flow-based charging.

The HSS 70 is also called a Home Location Register (HLR) and includes SAE subscription data including an EPS-subscribed QoS profile and access control information for roaming. In addition, information about the PDN that the user accesses is also included. This information can be maintained in the form of an Access Point Name (APN), which is a DNS (Domain Name system)-based label, which identifies an access point to a PDN or a PDN address indicating a subscribed IP address. it is a technique

As shown in FIG. 1, various interfaces such as S1-U, S1-MME, S5/S8, S11, S6a, Gx, Rx, and SG may be defined between EPS network elements.

Hereinafter, the concept of mobility management (MM) and a mobile line management (MM) back-off timer will be described in detail. Mobility Management (MM) is a procedure for reducing overhead on the E-UTRAN and processing at the UE.

When mobility management (MM) is applied, all information related to the UE in the access network may be released during a period in which data is deactivated. The MME may maintain information related to a UE context and configured bearer during the idle period.

So that the network can contact the UE in the ECM-IDLE state, the UE can notify the network about its new location whenever it leaves its current TA (Tracking Area). This procedure may be called "Tracking Area Update", and this procedure may be called "Routing Area Update" in a universal terrestrial radio access network (UTRAN) or GSM EDGE Radio Access Network (GERAN) system. The MME performs a function of tracking the user location while the UE is in the ECM-IDLE state.

If there is downlink data to be delivered to the UE in the ECM-IDLE state, the MME transmits a paging message to all base stations (eNodeBs) on a tracking area (TA) in which the UE is registered.

Then, the base station starts paging the UE over the radio interface. As the paging message is received, a procedure for making the state of the UE transition to the ECM-CONNECTED state is performed. This procedure can be called "Service Request Procedure". Accordingly, information related to the UE is generated in the E-UTRAN, and all bearers are re-established. The MME plays a role of resetting a radio bearer and updating a UE context on a base station.

When the aforementioned mobility management (MM) procedure is performed, a mobility management (MM) backoff timer may be additionally used. Specifically, a Tracking Area Update (TAU) may be transmitted to renew the TA, and the MME may reject the TAU request due to core network congestion. In this case, the time related to the MM backoff timer value can be provided. Upon receiving the corresponding time value, the UE may activate the MM backoff timer.

2 is a block diagram illustrating an example of a radio protocol architecture to which technical features of the present invention may be applied.

(a) of FIG. 2 shows an example of a radio protocol architecture for a user plane, and (b) of FIG. 2 shows an example of a radio protocol architecture for a control plane. It is a block diagram showing an example.

The user plane is a protocol stack for transmitting user data, and the control plane is a protocol stack for transmitting control signals.

Referring to (a) and (b) of FIG. 2, a physical (PHY) layer provides an information transfer service to an upper layer using a physical channel. The physical layer is connected to a medium access control (MAC) layer, which is an upper layer, through a transport channel. Data moves between the MAC layer and the physical layer through the transport channel. Transmission channels are classified according to how and with what characteristics data is transmitted through the air interface.

Data moves between different physical layers, that is, between physical layers of a transmitter and a receiver through a physical channel. The physical channel may be modulated using OFDM (Orthogonal Frequency Division Multiplexing) and utilizes time and frequency as radio resources.

The function of the MAC layer is mapping between logical channels and transport channels and multiplexing/demultiplexing ('/' includes the concepts of 'or' and 'and'). The MAC layer provides services to the Radio Link Control (RLC) layer through logical channels.

Functions of the RLC layer include concatenation, segmentation, and reassembly of RLC SDUs. In order to guarantee various Quality of Service (QoS) required by the Radio Bearer (RB), the RLC layer has transparent mode (TM), unacknowledged mode (UM), and acknowledged mode (Acknowledged Mode). , AM) provides three operation modes. AM RLC provides error correction through automatic repeat request (ARQ).

The Radio Resource Control (RRC) layer is defined only in the control plane. The RRC layer is responsible for the control of logical channels, transport channels, and physical channels in relation to configuration, re-configuration, and release of radio bearers. RB means a logical path provided by the first layer (PHY layer) and the second layer (MAC layer, RLC layer, PDCP layer) for data transfer between the terminal and the network.

The functions of the Packet Data Convergence Protocol (PDCP) layer in the user plane include delivery of user data, header compression, and ciphering. Functions of a Packet Data Convergence Protocol (PDCP) layer in the control plane include transmission of control plane data and encryption/integrity protection.

Establishing an RB means a process of defining characteristics of a radio protocol layer and a channel and setting specific parameters and operation methods to provide a specific service. RBs can be further divided into two types: Signaling RBs (SRBs) and Data RBs (DRBs). The SRB is used as a path for transmitting RRC messages in the control plane, and the DRB is used as a path for transmitting user data in the user plane.

When an RRC connection is established between the RRC layer of the terminal and the RRC layer of the E-UTRAN, the terminal is in an RRC connected state, otherwise it is in an RRC idle state.

A downlink transmission channel for transmitting data from a network to a terminal includes a broadcast channel (BCH) for transmitting system information and a downlink shared channel (SCH) for transmitting user traffic or control messages. Traffic or control messages of a downlink multicast or broadcast service may be transmitted through a downlink SCH or may be transmitted through a separate downlink multicast channel (MCH). Meanwhile, an uplink transmission channel for transmitting data from a terminal to a network includes a random access channel (RACH) for transmitting an initial control message and an uplink shared channel (SCH) for transmitting user traffic or control messages.

Logical channels located above transport channels and mapped to transport channels include BCCH (Broadcast Control Channel), PCCH (Paging Control Channel), CCCH (Common Control Channel), MCCH (Multicast Control Channel), MTCH (Multicast Traffic Channel) Channel), etc.

A physical channel is composed of several OFDM symbols in the time domain and several sub-carriers in the frequency domain. One sub-frame is composed of a plurality of OFDM symbols in the time domain. A resource block is a resource allocation unit and is composed of a plurality of OFDM symbols and a plurality of sub-carriers. In addition, each subframe may use specific subcarriers of specific OFDM symbols (eg, the first OFDM symbol) of the corresponding subframe for a Physical Downlink Control Channel (PDCCH), that is, an L1/L2 control channel. TTI (Transmission Time Interval) is a unit time of subframe transmission.

3 is a diagram for explaining physical channels used in a 3GPP LTE/LTE-A system to which the present invention can be applied and a general signal transmission method using them.

In a state where the power is turned off, the power is turned on again or the terminal that newly enters the cell performs an initial cell search operation such as synchronizing with the base station in step S3010. To this end, the terminal synchronizes with the base station by receiving a primary synchronization channel (P-SCH) and a secondary synchronization channel (S-SCH) from the base station, and obtains information such as a cell ID (identifier). do.

Thereafter, the terminal may acquire intra-cell broadcast information by receiving a physical broadcast channel (PBCH) signal from the base station. Meanwhile, the terminal may check the downlink channel state by receiving a downlink reference signal (DL RS) in the initial cell search step.

After completing the initial cell search, the UE can obtain more specific system information by receiving the PDCCH and the PDSCH according to the PDCCH information in step S3020.

Thereafter, the terminal may perform a random access procedure such as steps S3030 to S3060 in order to complete access to the base station. To this end, the UE may transmit a preamble through a physical random access channel (PRACH) (S3030), and receive a response message to the preamble through a PDCCH and a corresponding PDSCH (S3040). . In the case of contention-based random access, the UE may perform a contention resolution procedure such as transmitting an additional PRACH signal (S3050) and receiving a PDCCH signal and a corresponding PDSCH signal (S3060).

After performing the procedure as described above, the terminal receives a PDCCH signal and/or a PDSCH signal as a general uplink/downlink signal transmission procedure (S3070) and a physical uplink shared channel (PUSCH) signal and/or a physical uplink control channel. (PUCCH) signal transmission (S3080) may be performed.

Control information transmitted from the terminal to the base station is collectively referred to as uplink control information (UCI). UCI includes HARQ-ACK/NACK, scheduling request (SR), channel quality indicator (CQI), precoding matrix indicator (PMI), rank indicator (RI) information, and the like.

In the LTE/LTE-A system, UCI is generally transmitted periodically through PUCCH, but may be transmitted through PUSCH when control information and traffic data need to be simultaneously transmitted. In addition, UCI may be transmitted aperiodically through the PUSCH according to a request/instruction of the network.

4 shows the structure of a radio frame in a radio communication system to which the present invention can be applied.

3GPP LTE/LTE-A supports a type 1 radio frame structure applicable to frequency division duplex (FDD) and a type 2 radio frame structure applicable to time division duplex (TDD).

In FIG. 4, the size of a radio frame in the time domain is expressed as a multiple of a time unit of T_s=1/(15000*2048). Downlink and uplink transmission consists of radio frames having a section of T_f = 307200 * T_s = 10 ms.

(a) of FIG. 4 illustrates the structure of a type 1 radio frame. Type 1 radio frames can be applied to both full duplex and half duplex FDD.

A radio frame consists of 10 subframes. One radio frame consists of 20 slots with a length of T_slot = 15360 * T_s = 0.5 ms, and each slot is given an index from 0 to 19. One subframe is composed of two consecutive slots in the time domain, and subframe i is composed of slot 2i and slot 2i+1. The time taken to transmit one subframe is referred to as a transmission time interval (TTI). For example, one subframe may have a length of 1ms and one slot may have a length of 0.5ms.

In FDD, uplink transmission and downlink transmission are distinguished in the frequency domain. While there is no limitation in full-duplex FDD, in half-duplex FDD operation, a terminal cannot simultaneously transmit and receive.

One slot includes a plurality of orthogonal frequency division multiplexing (OFDM) symbols in the time domain and includes a plurality of resource blocks (RBs) in the frequency domain. Since 3GPP LTE uses OFDMA in downlink, an OFDM symbol is used to represent one symbol period. An OFDM symbol may be referred to as one SC-FDMA symbol or symbol period. A resource block is a resource allocation unit and includes a plurality of consecutive subcarriers in one slot.

(b) of FIG. 4 shows a type 2 frame structure (frame structure type 2).

A type 2 radio frame consists of two half frames each having a length of 153600*T_s=5 ms. Each half frame consists of 5 subframes with a length of 30720*T_s = 1 ms.

In the type 2 frame structure of the TDD system, the uplink-downlink configuration is a rule indicating whether uplink and downlink are allocated (or reserved) for all subframes.

Table 1 shows an uplink-downlink configuration.

[Table 1]

Referring to Table 1, for each subframe of a radio frame, 'D' represents a subframe for downlink transmission, 'U' represents a subframe for uplink transmission, and 'S' represents a DwPTS (Downlink Pilot Time Slot), Guard Period (GP), Uplink Pilot Time Slot (UpPTS) represents a special subframe composed of three fields.

DwPTS is used for initial cell search, synchronization or channel estimation in a UE. UpPTS is used to match channel estimation in the base station with uplink transmission synchronization of the terminal. The GP is a section for removing interference generated in uplink due to multipath delay of a downlink signal between uplink and downlink.

Each subframe i consists of a slot 2i and a slot 2i+1 each having a length of T_slot=15360*T_s=0.5 ms.

Uplink-downlink configurations can be classified into seven types, and the positions and/or numbers of downlink subframes, special subframes, and uplink subframes are different for each configuration.

The point at which the downlink is changed to the uplink or the point at which the uplink is switched to the downlink is referred to as a switching point. Switch-point periodicity refers to a period in which an uplink subframe and a downlink subframe are switched in the same way, and both 5 ms and 10 ms are supported. In the case of having a period of 5 ms downlink-uplink switching time, the special subframe (S) exists in every half-frame, and in the case of having a period of 5 ms downlink-uplink switching time, it exists only in the first half-frame.

In all configurations, subframes 0 and 5 and DwPTS are intervals only for downlink transmission. A subframe immediately following the UpPTS and subframe subframe is always a period for uplink transmission.

This uplink-downlink configuration may be known to both the base station and the terminal as system information. The base station can inform the terminal of the change in the uplink-downlink allocation state of the radio frame by transmitting only the index of the configuration information whenever the uplink-downlink configuration information changes. In addition, the configuration information is a kind of downlink control information and can be transmitted through PDCCH (Physical Downlink Control Channel) like other scheduling information, and is commonly transmitted to all terminals in a cell through a broadcast channel as broadcast information. It could be.

Table 2 shows the configuration of the special subframe (length of DwPTS/GP/UpPTS).

[Table 2]

The structure of the radio frame according to the example of FIG. 4 is only one example, and the number of subcarriers included in the radio frame, the number of slots included in the subframe, and the number of OFDM symbols included in the slot may be variously changed. can

5 is a diagram illustrating a resource grid for one downlink slot in a wireless communication system to which the present invention can be applied.

Referring to FIG. 5, one downlink slot includes a plurality of OFDM symbols in the time domain. Here, it is described as an example that one downlink slot includes 7 OFDM symbols and one resource block includes 12 subcarriers in the frequency domain, but is not limited thereto.

Each element on the resource grid is a resource element, and one resource block (RB) includes 12 × 7 resource elements. The number of resource blocks N^DL included in a downlink slot depends on a downlink transmission bandwidth.

A structure of an uplink slot may be the same as that of a downlink slot.

6 shows a structure of a downlink subframe in a wireless communication system to which the present invention can be applied.

Referring to FIG. 6, up to three OFDM symbols in the first slot of a subframe are a control region to which control channels are allocated, and the remaining OFDM symbols are a data region to which a Physical Downlink Shared Channel (PDSCH) is allocated ( data region). Examples of downlink control channels used in 3GPP LTE include a Physical Control Format Indicator Channel (PCFICH), a Physical Downlink Control Channel (PDCCH), and a Physical Hybrid-ARQ Indicator Channel (PHICH).

The PCFICH is transmitted in the first OFDM symbol of a subframe and carries information about the number of OFDM symbols (ie, the size of the control region) used for transmission of control channels in the subframe. The PHICH is a response channel for uplink and carries an Acknowledgement (ACK)/Not-Acknowledgement (NACK) signal for Hybrid Automatic Repeat Request (HARQ). Control information transmitted through the PDCCH is referred to as downlink control information (DCI). The downlink control information includes uplink resource allocation information, downlink resource allocation information, or an uplink transmission (Tx) power control command for a certain UE group.

The PDCCH includes resource allocation and transmission format of Downlink Shared Channel (DL-SCH) (this is also referred to as a downlink grant), resource allocation information of Uplink Shared Channel (UL-SCH) (this is also referred to as an uplink grant), and PCH (this is also referred to as an uplink grant). Resource allocation for upper-layer control messages such as paging information in paging channel, system information in DL-SCH, and random access response transmitted in PDSCH, arbitrary terminal A set of transmission power control commands for individual terminals in a group, activation of VoIP (Voice over IP), and the like can be carried. A plurality of PDCCHs may be transmitted within a control region, and a terminal may monitor a plurality of PDCCHs. The PDCCH is composed of a set of one or more consecutive control channel elements (CCEs). CCE is a logical allocation unit used to provide a PDCCH with a coding rate according to a state of a radio channel. CCE corresponds to a plurality of resource element groups. The format of the PDCCH and the number of usable bits of the PDCCH are determined according to the relationship between the number of CCEs and the coding rate provided by the CCEs.

The base station determines the PDCCH format according to the DCI to be transmitted to the terminal, and attaches a Cyclic Redundancy Check (CRC) to the control information. In the CRC, a unique identifier (referred to as a Radio Network Temporary Identifier (RNTI)) according to the owner or purpose of the PDCCH is masked. If the PDCCH is for a specific UE, a unique identifier of the UE, for example, C-RNTI (Cell-RNTI) may be masked to the CRC. Alternatively, if the PDCCH is for a paging message, a paging indication identifier, for example, Paging-RNTI (P-RNTI) may be masked to the CRC. If the PDCCH is for system information, more specifically, a system information block (SIB), a system information identifier and a system information RNTI (SI-RNTI) may be masked to the CRC. In order to indicate a random access response that is a response to the transmission of the random access preamble of the UE, a random access-RNTI (RA-RNTI) may be masked to the CRC.

7 shows a structure of an uplink subframe in a wireless communication system to which the present invention can be applied.

Referring to FIG. 7, an uplink subframe can be divided into a control region and a data region in the frequency domain. A physical uplink control channel (PUCCH) carrying uplink control information is allocated to the control region. The data area is allocated a Physical Uplink Shared Channel (PUSCH) carrying user data. In order to maintain single carrier characteristics, one UE does not simultaneously transmit PUCCH and PUSCH.

A pair of resource blocks (RBs) is allocated in a subframe to the PUCCH for one UE. RBs belonging to an RB pair occupy different subcarriers in each of two slots. This RB pair allocated to the PUCCH is said to be frequency hopping at a slot boundary.

Reference Signal (RS)

Since data is transmitted through a radio channel in a wireless communication system, the signal may be distorted during transmission. In order to accurately receive the distorted signal at the receiving end, the distortion of the received signal must be corrected using channel information. In order to detect channel information, a signal transmission method known to both the transmitting side and the receiving side and a method of detecting channel information using a distortion degree when a signal is transmitted through a channel are mainly used. The above signal is referred to as a pilot signal or a reference signal (RS).

In addition, when transmitting packets in most recent mobile communication systems, a method to improve transmission and reception data efficiency by adopting multiple transmission antennas and multiple reception antennas, breaking away from the previous use of one transmission antenna and one reception antenna, has been devised. use. When data is transmitted/received using multiple input/output antennas, a channel state between a transmit antenna and a receive antenna must be detected in order to accurately receive a signal. Therefore, each transmit antenna must have an individual reference signal.

In a mobile communication system, RS can be largely classified into two types according to its purpose. There is an RS for the purpose of obtaining channel information and an RS used for data demodulation. Since the purpose of the former is for the UE to acquire downlink channel information, it must be transmitted over a wide band, and even a UE that does not receive downlink data in a specific subframe must be able to receive and measure the RS. In addition, it is also used for measurement such as handover. The latter is an RS sent along with a corresponding resource when the base station transmits downlink, and the UE can estimate a channel by receiving the corresponding RS, and thus demodulate data. This RS must be transmitted in the area where data is transmitted.

The downlink reference signal includes one common reference signal (CRS: common RS) for acquisition of channel state information shared by all terminals in a cell, measurement of handover, etc., and a dedicated reference used for data demodulation only for a specific terminal. There is a signal (DRS: dedicated RS). Information for demodulation and channel measurement can be provided using these reference signals. That is, DRS is used only for data demodulation, and CRS is used for both purposes of acquiring channel information and demodulating data.

The receiving side (i.e., UE) measures the channel state from the CRS, and transmits indicators related to channel quality such as CQI (Channel Quality Indicator), PMI (Precoding Matrix Index) and/or RI (Rank Indicator) to the transmitting side (i.e., base station). The CRS is also referred to as a cell-specific RS. On the other hand, a reference signal related to channel state information (CSI) feedback may be defined as a CSI-RS.

DRS may be transmitted through resource elements when data demodulation on the PDSCH is required. The UE may receive the presence or absence of the DRS through a higher layer, and it is effective only when the corresponding PDSCH is mapped. The DRS may be referred to as a UE-specific RS or a Demodulation RS (DMRS).

Synchronization Signal (SS)

When the UE is powered on or newly enters a cell, an initial cell search process such as obtaining time and frequency synchronization with the cell and detecting a physical cell identity of the cell (procedure) is performed. To this end, the UE synchronizes with the eNB by receiving a synchronization signal, for example, a primary synchronization signal (PSS) and a secondary synchronization signal (SSS) from the eNB, and a cell identifier (ID : identity) and the like can be obtained.

8 shows an example of a radio frame structure for transmission of a synchronization signal in a wireless communication system to which the present invention can be applied.

In particular, FIG. 8 illustrates a radio frame structure for transmission of a synchronization signal and PBCH in frequency division duplex (FDD), and FIG. 8(b) shows transmission positions of SS and PBCH in a radio frame configured as an extended CP.

SS is divided into PSS and SSS. PSS is used to obtain time domain synchronization and/or frequency domain synchronization such as OFDM symbol synchronization, slot synchronization, etc., and SSS is used for frame synchronization, cell group ID and/or cell CP configuration (i.e., use of normal CP or extended CP). information) is used.

Referring to FIG. 8, in the time domain, PSS and SSS are transmitted in two OFDM symbols of each radio frame. Specifically, SS is performed in the first slot of subframe 0 and the first slot of subframe 5 in consideration of the GSM (Global System for Mobile communication) frame length of 4.6 ms for ease of inter radio access technology (RAT) measurement. are sent to each In particular, the PSS is transmitted in the last OFDM symbol of the first slot of subframe 0 and the last OFDM symbol of the first slot of subframe 5, respectively, and the SSS is transmitted in the second to last OFDM symbol of the first slot of subframe 0 and the subframe. It is transmitted in the second to last OFDM symbol of the first slot of frame 5, respectively.

A boundary of the corresponding radio frame may be detected through SSS. The PSS is transmitted in the last OFDM symbol of the corresponding slot, and the SSS is transmitted in the OFDM symbol immediately preceding the PSS. The transmit diversity method of SS uses only a single antenna port and is not separately defined in the standard. That is, a single antenna port transmission or a transmission method that is transparent to the UE (eg, Precoding Vector Switching (PVS), Time Switched Diversity (TSTD), Cyclic Delay Diversity (CDD)) can be used for transmit diversity of the SS. .

Since the PSS is transmitted every 5 ms, the UE can know that the corresponding subframe is one of subframe 0 and subframe 5 by detecting the PSS, but it cannot know which subframe is specifically between subframe 0 and subframe 5. . Therefore, the UE cannot recognize the boundary of the radio frame only with the PSS. That is, frame synchronization cannot be obtained only with the PSS. The UE detects the boundary of the radio frame by detecting the SSS transmitted twice in one radio frame but transmitted in different sequences.

In the frequency domain, PSS and SSS are mapped to 6 RBs located at the center of the downlink system bandwidth. In downlink, the total number of RBs may be composed of different numbers of RBs (eg, 6 RBs to 110 RBs) according to system bandwidth, but PSS and SSS are 6 RBs located in the center of the downlink system bandwidth Since they are mapped, the UE can detect the PSS and SSS in the same way regardless of the downlink system bandwidth.

Both PSS and SSS are composed of sequences of length 62. Therefore, among the 6 RBs, 62 subcarriers in the middle located on both sides of the DC subcarrier are mapped, and the DC subcarrier and 5 subcarriers located at both ends are not used.

The UE can obtain a physical layer cell ID according to a specific sequence of PSS and SSS. That is, the SS may indicate a total of 504 unique physical layer cell IDs through a combination of 3 PSSs and 168 SSSs.

In other words, the physical layer cell IDs are 168 physical-layer cell-identifier groups where each group contains 3 unique identifiers such that each physical-layer cell ID is part of only one physical-layer cell-identifier group. are grouped by Therefore, the physical layer cell identifier Ncell ID = 3N(1) ID + N(2) ID is a number N(1) ID in the range of 0 to 167 representing a physical-layer cell-identifier group and the physical-layer cell-identifier group. It is uniquely defined by a number N(2) ID from 0 to 2 representing the physical-layer identifier within the identifier group.

The UE can detect the PSS to know one of the three unique physical-layer identifiers, and detect the SSS to identify one of the 168 physical layer cell IDs associated with the physical-layer identifier.

SSS is generated based on M-sequence. Each SSS sequence is generated by interleaved concatenation of two SSC 1 sequences and SSC 2 sequences having a length of 31 in the frequency domain. By combining the two sequences, 168 cell group IDs are transmitted. As a sequence of SSS, m-sequence is robust in a frequency selective environment, and the amount of computation can be reduced by fast m-sequence transformation using fast Hadamard transform. In addition, configuring the SSS with two short codes has been proposed to reduce the amount of operation of the terminal.

9 shows an example of a structure of a secondary synchronization signal in a wireless communication system to which the present invention can be applied.

9 illustrates a structure in which two sequences for generating a secondary synchronization signal are interleaved and mapped in the physical domain.

When the two m-sequences used for SSS code generation are defined as SSS 1 and SSS 2, respectively, if the SSS of subframe 0 transmits the cell group identifier in two combinations (SSS 1, SSS 2), subframe 5 By swapping and transmitting the SSS of (SSS 2, SSS 1), it is possible to distinguish the 10 ms frame boundary. At this time, the SSS code used uses a generation polynomial of x ⁵ +x ² +1, and a total of 31 codes can be generated through different circular shifts.

In order to improve reception performance, two different PSS-based sequences are defined and scrambled in SSS, but different sequences are scrambled in SSS 1 and SSS 2. Then, by defining a scrambling code based on SSS 1, scrambling is performed on SSS 2. At this time, the SSS code is exchanged every 5 ms, but the PSS-based scrambling code is not exchanged.

The PSS-based scrambling code is defined as 6 cyclic shift versions according to the PSS index in the m-sequence generated from the generation polynomial of x ⁵ +x ² +1, and the SSS 1-based scrambling code is x ⁵ +x ⁴ +x It is defined as 8 cyclic shift versions according to the index of SSS 1 in the m-sequence generated from the polynomial of ² +x ¹ +1.

10 and 11 are diagrams illustrating an example of a method of receiving a pilot signal for channel estimation using a directional beam to which the present invention can be applied.

Referring to FIGS. 10 and 11 , in order to design channel estimation-based beamforming, channel information must first be obtained through a pilot transmission step. However, unlike the Conventional MIMO system that transmits channel estimation pilots in omni-direction, in millimeter band communication, in order to overcome the degradation of the SNR of the received signal due to the high path loss characteristic, FIG. 10 (a) and (b) ), the pilot is transmitted using a directional beam through analog beamforming.

That is, in the case of the Conventional MIMO system, since pilot signals are transmitted in all directions, the terminal can estimate the channel by checking the path of the pilot signals transmitted through each cluster through the pilot signal broadcast from the base station.

However, in the case of millimeter band communication, since a pilot signal is transmitted through a specific path using a directional beam through analog beamforming to reduce path loss, the terminal can estimate only a specific channel through the received pilot signal.

In this case, the terminal needs to estimate a channel through pilot signals received through several transmit/receive beam pairs.

However, since the pilot signals are transmitted to the terminal through different paths as shown in (a) and (b) of FIG. 11 due to different directional beam pairs, they arrive at the terminal through different path delays. , there is a problem in that it is difficult to obtain time synchronization of pilot signals in an asynchronous situation such as a cell search process.

That is, when a pilot signal is received through a directional beam, since each pilot signal has a different path delay, time synchronization obtained through the pilot signal is different for each beam pair, and for a specific pilot beam pair, a very low In some cases, time/frequency synchronization cannot be obtained according to the SNR.

Also, since pilot signals are transmitted through different night-directional beams, estimated channels may be different for each channel.

Therefore, in order to solve this problem, the present invention proposes a method of estimating a channel by aligning a plurality of pilot signals obtained through directional beams according to time, as in the case of pilot signals received in all directions.

12 is a diagram illustrating an example of a structure of a frame and a pilot signal proposed in the present invention.

,

(또는

,

) 안테나 수를 가지고 있으며 같은

RF 체인 수를 가지고 있다. 하나의 OFDM 심볼 길이는

로 CP길이와 데이터길이의 합으로 표현된다. 하나의 시간 도메인에서 OFDM 심볼은

샘플로 이루어져 있다.In the present invention, a cellular-based OFDM millimeter communication cell search environment is considered in a hybrid MIMO system. base station and terminal respectively

,

(or

,

) has the number of antennas and has the same

It has the number of RF chains. The length of one OFDM symbol is

It is expressed as the sum of CP length and data length. An OFDM symbol in one time domain is

made up of samples.

As shown in FIG. 12, the downlink frame structure proposed in the present invention is composed of a synchronization subframe D _sync-sb and a data subframe D _data-sb in D _frame cycles.

One synchronization subframe consists of N _sync OFDM symbols. 12 shows a frequency domain pilot structure of a synchronization frame structure in a downlink system.

로 표현 가능하다. 동기 서브 프레임의 OFDM 심볼은 다운링크 동기를 위한 SS(sync signal), 빔 정보를 담고 있는 ESS(extended sync signal), 채널 추정을 위한 RS(reference signal) 그리고 브로드캐스팅 되는 정보를 포함하고 있다.The frequency domain OFDM symbol of the p-th synchronization subframe is a vector of length K (DFT size).

can be expressed as The OFDM symbol of the synchronization subframe includes a sync signal (SS) for downlink synchronization, an extended sync signal (ESS) containing beam information, a reference signal (RS) for channel estimation, and broadcast information.

,

,

로 정의한다. SS 시퀀스

는 k가 짝수 인 곳에 zadoff-chu 시퀀스 값을 가지고 홀수 인 곳에는 0으로 할당한다. 이는 시간 도메인에서 신호가 반복되게 함으로써 주파수 동기 추정에 용이하기 위함이다.The set of sub-carrier indices where SS, ESS, and RS are located are each

,

,

is defined as SS sequence

assigns a zadoff-chu sequence value where k is even and 0 where k is odd. This is to facilitate frequency synchronization estimation by repeating the signal in the time domain.

개의 송신 DFT빔을 통해 동기 서브 프레임의 p-th OFDM심볼에 p-th DFT빔을 사용한다. (

) 단말은 수신 빔으로

개의 DFT 빔을 사용하며 이는

의 배수로 표현된다. 셀 탐색 과정 동안 단말은

개의 수신 빔을 하나의 프레임 길이와 하나의 심볼 길이(

)를 주기로 빔을 바꿔간다. 하나의 심볼 길이를 더 보는 것은 단말이 셀 탐색 과정을 시작할 때 기지국에서 전송한 OFDM 심볼 중간에 진입하는 경우 때문이다. 이에 따라 단말이 모든 방향에서 오는 파일럿 신호를 받으려면

시간이 필요하다.OFDM symbols of sync subframes are transmitted through different transmit and receive directional DFT beams. Base station per symbol

The p-th DFT beam is used for the p-th OFDM symbol of the synchronization subframe through the transmission DFT beams. (

) The terminal uses the receive beam

uses DFT beams, which are

expressed as a multiple of During the cell discovery process, the terminal

The number of receive beams is one frame length and one symbol length (

) to change the beam. One more symbol length is seen because the terminal enters the middle of the OFDM symbol transmitted by the base station when starting the cell search process. Accordingly, in order for the UE to receive pilot signals from all directions,

It takes time.

The present invention proposes a method for solving the problem of the method for receiving a pilot signal based on channel estimation in the process of searching for a millimeter communication cell described above.

13 is a flowchart illustrating an example of a method for estimating a channel using a pilot signal proposed in the present invention.

Referring to FIG. 13, in order to estimate a channel through a plurality of pilot signals received through different directional beams, the present invention arranges the received pilot signals in time as if they were received through omni-directional beams to estimate the channel. .

Specifically, the terminal receives a plurality of pilot signals transmitted from the base station through a plurality of directional beams (S13010).

Thereafter, since signals received through different directional beams are asynchronous without acquiring synchronization, time synchronization of the pilot signals is acquired and frequency offset is compensated (S13020).

, 시간 동기 획득 및 주파수 오프셋이 보상된 신호들의 집합을

라 한다.At this time, the received signal is

, a set of signals with time synchronization acquisition and frequency offset compensation

say

Thereafter, the terminal performs a time alignment process for compensating for an effect caused by a plurality of different directional beams of a plurality of pilot signals (S13030).

라 한다.At this time, the time alignment process is called a space-time aligner procedure, and the point corrected through the space-time alignment process is

say

지점으로부터 채널을 추정하기 위한 파일럿 심볼들을 선택하고, 선택된 파일럿 심볼들을 DFT 하여 채널 추정을 위한 신호들을 획득한다(S13040).Terminals are sorted

Pilot symbols for channel estimation are selected from the point, and signals for channel estimation are obtained by DFTing the selected pilot symbols (S13040).

Thereafter, the terminal estimates a channel based on the acquired signals (S13050).

Using this method, the terminal can estimate the same channel as the pilot signal of the omni-directional beam by compensating for the path loss and delay of the pilot signals acquired through the directional beam.

Hereinafter, each step will be described in detail.

14 is a diagram illustrating an example of a method for obtaining time synchronization of a pilot signal proposed in the present invention.

Referring to FIG. 14 , the terminal may obtain time synchronization by finding a time at which a plurality of pilot signals received through a plurality of directional beams are received, and compensate for a frequency offset.

Specifically, the terminal selects a signal to be used for channel estimation from among a plurality of pilot signals received through a plurality of directional beams through a pre-procedure, and synchronizes the time of the selected signal through a synchronization procedure. and frequency offset can be compensated.

Pre-Procedure

를 동기를 획득하기 위한 동기 신호인 SS가 위치하는 서브캐리어 인덱스

에 있는 파일럿 신호를 추출하기 위해서 저역 통과 필터(Low Pass Filter:LPF)를 통과시킨다(S14010).The terminal is a signal received through different transmit and receive directional beams.

Subcarrier index where SS, which is a synchronization signal for obtaining synchronization of

In order to extract the pilot signal in , a low pass filter (LPF) is passed (S14010).

를

라 호칭한다.At this time, passing the LPF

cast

call it

는 Moving Window를 통해 신호 검출 판단 및 동기를 획득하기 위한 구간이 결정된다(S14020).Signal passed through LPF

A section for signal detection determination and synchronization acquisition is determined through the Moving Window (S14020).

으로 홀, 짝으로 구분시켜 윈도우 크기의 50%씩 교차시키는 구조를 가진다.Moving Window is a window that determines the interval for signal detection judgment and synchronization. The size of the window is twice the symbol length.

It has a structure in which 50% of the window size is crossed by dividing it into odd and pair.

Through this, as shown in FIG. 12, one complete symbol is included in one window.

가 아래 수학식 1과 같이 정의될 때, N_j는 수학식 2와 같이 정의될 수 있다.Range of signals included in the window of j-th

When is defined as in Equation 1 below, N _j may be defined as in Equation 2.

의 구간에서 아래 수학식 3을 통해 신호의 반복성을 이용한 normalized auto-correlation을 구할 수 있다(S14030).The terminal that obtains the complete symbol through the window

In the interval of , normalized auto-correlation using signal repeatability can be obtained through Equation 3 below (S14030).

)보다 작을 경우 신호를 버리고 클 경우 최대값을 가지는 시간 인덱스를 대략적인 신호의 수신 지점으로 정의한다(S14040).After that, the terminal determines that C _j , which is the maximum value of normalized auto-correlation within the j-th window, is the threshold value set by the system (

), the signal is discarded if it is less than, and the time index having the maximum value is defined as the approximate signal reception point (S14040).

라 호칭한다.At this time, the reception point of the approximate signal

call it

Synchronization Procedure

에서의 각도를 구하여 주파수 오차를 구할 수 있다(S14050).After that, the terminal through Equation 4 below

A frequency error may be obtained by obtaining an angle at (S14050).

에 곱해줌으로써 주파수 오차를 보상하여

를 획득할 수 있다.The terminal uses the estimated frequency error as the received signal as shown in Equation 5 below.

By multiplying by, the frequency error is compensated for

can be obtained.

는 아래 수학식 6과 같이 주파수 오프셋인 CFO 보상을 마친 신호인

와 시간영역 파일럿 신호의 cross-correlation을 통해 획득될 수 있다.time synchronous

Is a signal after completing the frequency offset CFO compensation as shown in Equation 6 below.

It can be obtained through cross-correlation of and the time domain pilot signal.

은 단말에서 미리 알고 있는 SS 파일럿 신호를 의미한다.in Equation 6

denotes an SS pilot signal previously known by the terminal.

와 같은 정확한 시간 동기를 획득하는 경우, 시간 동기를 획득한 파일럿 신호들의 수를

라 한다.Through this procedure, the terminal

In the case of obtaining accurate time synchronization such as

say

라 하고, j_i는

를 만족한다.In addition, a set of indexes of windows in which pilot signals obtained for time synchronization are found are

, and j _i is

satisfies

라 한다.The point at which the pilot signal obtained for time synchronization is found is set

say

Through this procedure, the terminal can obtain time synchronization of pilot signals received through different directional beams.

15 to 17 are diagrams illustrating an example for receiving a pilot signal by compensating for time synchronization according to a path delay of a pilot signal proposed in the present invention.

15 to 17, by temporally aligning a plurality of pilot signals received through a plurality of different directional beams having time synchronization obtained, the plurality of pilot signals are aligned like signals received through omnidirectional beams. It can be.

Specifically, in order to perform channel estimation using a plurality of pilot signals received through different directional beams, a plurality of pilot signals received through time must be aligned in time as if they were received through omnidirectional beams.

In order to time-align pilot signals obtained from time synchronization, a beam sweeping effect due to a directional beam and a propagation delay due to a plurality of paths must be checked.

That is, as shown in FIG. 15, in order to compensate for the beam sweeping effect caused by different directional beams, the received pilot signals must be aligned in time as if they were received through omnidirectional beams.

To this end, a virtual system that is identical to the mmWave system but uses a transmit/receive antenna capable of beamforming in all directions rather than a directional beam is considered in order to temporally align the pilot signals obtained for time synchronization.

개수의 경로를 얻을 수 있다고 가정한다.In the virtual system in the omni direction, the transmit power is sufficiently large even with beamforming, so that it is the same as in the original mmWave system.

Suppose we can get the number of paths.

Thereafter, the received pilot signal may be aligned in time through a spatial alignment procedure as received through the omnidirectional beam, and a signal for channel estimation may be obtained by compensating for a time delay on a path through a temporal alignment procedure. .

Spatial alignment procedure

16 shows an example of a method of aligning received pilot signals in time as if they were received through an omnidirectional beam.

), omni방향으로의 수신빔 1개(

)인 mmWave 시스템을 가정한다.16 shows three transmission beams (

), one reception beam in the omni direction (

) assuming a mmWave system.

는 가상 시스템에서의 i-번째 경로로 도착한 파일럿 신호를 의미하며, 각 경로의 파일럿 신호간 차이를

로 정의한다.in Figure 16

denotes a pilot signal arriving at the i-th path in the virtual system, and the difference between the pilot signals of each path is

is defined as

와

간에는 일대일 대응이 이루어진다.

에 해당되는 가상 시스템에서의 파일럿 신호를

라 정의한다. 이 때,

는

에 대응되는 가상 시스템의 파일럿 신호로 mapping하는 함수이다. 예를 들면,

,

,

를 만족하는 쌍은 가상시스템에서

에 대응하는 경로가

에 대응되는 경로보다 더 일찍 도착함을 나타낸다.At this time,

Wow

There is a one-to-one correspondence between

The pilot signal in the virtual system corresponding to

define it as At this time,

Is

It is a function that maps to the pilot signal of the virtual system corresponding to . For example,

,

,

The pair that satisfies

the path corresponding to

Indicates that it arrives earlier than the route corresponding to .

로 정의하고, 여기서

은 방향성 빔을 통해 수신된 파일럿 신호를 시간 상으로 정렬한 이후의 파일럿 신호를 의미하며,

를 의미한다.Also, for all i

defined as, where

denotes a pilot signal after aligning pilot signals received through a directional beam in time,

means

,

,

을 만족한다.16, S is {m1, m2, m3}, NPAT is 3, f(1)=(1,1), f(2)=(2,1), f(3)=(3, 1), assuming g(1)=3, g(2)=1, and g(3)=2,

,

,

satisfies

,

,

이고,

및

내의 파일럿 심볼들은 각각

,

크기만큼 지연이 발생한다.Referring to Figure 16,

,

,

ego,

and

Pilot symbols in

,

The delay occurs according to the size.

를 이용하면

와

사이의 관계를 도출 할 수 있다.This delay between pilot symbols is referred to as a first delay. The difference between each pilot symbol is

using

Wow

relationship between them can be derived.

In the case of m1 and m2, it can be expressed as in Equation 7 below.

m1, m2 and m1, m3 can also be expressed as in Equation 8 below.

를 만족하는

와

이 동일한 수신 빔(

)으로부터 얻어졌다고 가정하는 경우,

와

은 아래 수학식 9 및 수학식 10과 같다.

to satisfy

Wow

This same receive beam (

), assuming that it is obtained from

Wow

Is equal to Equation 9 and Equation 10 below.

는 j번째 경로와 i번째 경로의 전파 지연 사이의 차이를 나타내며 아래 수학식 11과 같이 정의된다.In Equations 9 and 10

Represents the difference between propagation delays of the j-th path and the i-th path and is defined as in Equation 11 below.

Equation 11 can be proved through Equations 12 and 13 below.

일 때,

과

는 아래 수학식 12와 같이 정의될 수 있다.

when,

class

Can be defined as in Equation 12 below.

일 때,

과

는 아래 수학식 13과 같이 정의될 수 있다.In addition,

when,

class

Can be defined as in Equation 13 below.

임에 따라,

조건은

번째,

번째 TD-OFDM파일럿 심볼이 모두 같은 sync-subframe에 속함을 의미한다.In addition,

As a result,

condition is

th,

This means that all of the th TD-OFDM pilot symbols belong to the same sync-subframe.

은 수학식 9를 따른다. 수학식 10의 ①의 3번째 항은

조건을 만족할 때

번째와

번째 파일럿 심볼 사이의 TD-OFDM심볼 수를 나타낸다. 수학식 10의 ①에서 3번째 항은 빔 스위핑 과정으로 인해 발생한다. 반면에

조건에서, 각각 의 수신 빔이

구간 동안 켜져있기 때문에

번째와

번째 TD-OFDM 파일럿 심볼을 포함하는 동기 subframe들은 하나의 데이터 subframe으로 분리 된다.The second term in ① of Equation 10

follows Equation 9. The third term of ① in Equation 10 is

when the condition is satisfied

second and

Indicates the number of TD-OFDM symbols between the pilot symbols. The third term in ① of Equation 10 is generated due to the beam sweeping process. on the other side

In the condition, each receive beam

because it is on during the

second and

Synchronization subframes including the th TD-OFDM pilot symbol are separated into one data subframe.

경우에서

번째와

번째 TD-OFDM 파일럿 심볼 사이의 TD-OFDM심볼 수를 나타낸다. 이는 빔 스위핑 과정으로 인해 발생한다.The second term in ② of Equation 10 follows Equation 9. The sum of the 3rd, 4th, and 5th terms in ② of Equation 10 is

in case

second and

Indicates the number of TD-OFDM symbols between the th TD-OFDM pilot symbols. This occurs due to the beam sweeping process.

쌍 에서

이다. 수학식 10에서 경로 지연 차이를 나타내는

는

인지 아닌지 여부를 판단하는데 사용될 수 있다.given

in pairs

to be. Representing the path delay difference in Equation 10

Is

It can be used to determine whether it is or not.

In FIG. 16, h ₁₂ (Δ), h ₁₃ (Δ), and h ₂₃ (Δ) satisfy Equation 14 below.

을 만족하고, 이는

를 나타낸다.Therefore, in FIG. 16

is satisfied, which is

indicates

는

의 배수로 표현된다. 따라서 수학식 10의 ①과 ②는 아래 수학식 15와 같이 정의될 수 있다.In ② of Equation 10

Is

expressed as a multiple of Therefore, ① and ② of Equation 10 can be defined as Equation 15 below.

는 양의 정수로

를 만족하는 경우, 아래 수학식 16을 통해서 계산될 수 있다.here

is a positive integer

When it satisfies, it can be calculated through Equation 16 below.

'는

보다 작거나 같은 제일 큰 정수를 의미한다.in Equation 16

'Is

The largest integer less than or equal to.

를 더하면, 아래 수학식 17과 같을 수 있다.On both sides of Equation 15

By adding , it may be as shown in Equation 17 below.

이므로,

을 만족한다. 따라서 수학식 15 에서

와

는

의 몫과 나머지를 각각 나타낸다. 이는 수학식 16을 의미한다.in Equation 17

Because of,

satisfies Therefore, in Equation 15

Wow

Is

represent the quotient and remainder of , respectively. This means Equation 16.

와

사이의 관계가 도출 될 수 있다.From Equation 16

Wow

A relationship between them can be derived.

은 CP이후 TD 파일럿 심볼의 시작 지점이기 때문에

를 만족하고.

인 경우에는 아래 수학식 18을 만족한다.Specifically,

Since is the starting point of the TD pilot symbol after CP

to be satisfied.

In the case of Equation 18 below is satisfied.

인 경우,

에 의해서 아래 수학식 19를 만족한다.In Equation 10,

If

Satisfies Equation 19 below by

)에도 적용될 수 있다.Equation 15 is a general case of using different receive beams (

) can also be applied.

번 째 파일럿 심볼과

번 째 파일럿 심볼 사이의 TD OFDM 파일럿 심볼 수는

의 배수로서

,

,

과 같은 용어로 표현될 수 있기 때문이다.this is

th pilot symbol and

The number of TD OFDM pilot symbols between the th pilot symbols is

as a multiple of

,

,

This is because it can be expressed in terms such as

인 경우, 앞에서 설명한 것과 같이 아래 표 3의 알고리즘 1과 같이 나타낼 수 있다.Therefore, this method can also be applied in general cases,

In the case of , as described above, it can be expressed as Algorithm 1 in Table 3 below.

[Table 3]

Temporal alignment procedure

을 이용하여 가상시스템인 전 방향 빔을 이용하여 수신된 파일럿 심볼과 동일한 시퀀스를 구해야 한다.Thereafter, the received signal r[n] and the signal aligned in time through the procedure described above,

It is necessary to obtain the same sequence as the pilot symbol received using the omnidirectional beam, which is a virtual system, using .

, 가상시스템으로 얻은 전체 신호를

이라 정의하고,

과

과의 관계는 아래 수학식 20과 같다.To this end, the received signal of the virtual system through the i-th path

, the total signal obtained by the virtual system

is defined as,

class

The relationship is as shown in Equation 20 below.

로 정의된 가상시스템에서 수신된 파일럿 심볼은

지점으로부터

개의 연속적인 샘플들로 획득된다. 가상시스템에서 얻은 파일럿 심볼은 아래 수학식 21과 같이 표현된다.

The pilot symbol received in the virtual system defined as

from the branch

is obtained with two consecutive samples. The pilot symbol obtained from the virtual system is expressed as Equation 21 below.

In Equation 21, the first equal sign is derived from Equation 20, and the second equal sign is obtained from Equation 22 below.

로 얻어진다.In Equation 21, the third equal sign is equal to Equation 19

is obtained with

Therefore, Equation 23 below is satisfied.

는 실제로 수신된 신호 r[n]으로부터 얻은 벡터

가 Shift된 신호를 의미한다.Each vector in Equation 23

is a vector obtained from the actually received signal r[n]

means a shifted signal.

이고,

인 경우,

를

지점부터

개의 연속적인 샘플들로 정의하면

를 만족하고, 아래 수학식 24와 같이 정의 된다.

ego,

If

cast

from the point

If we define the number of consecutive samples as

Satisfies, and is defined as in Equation 24 below.

를 얻는 과정은 파일럿 수신과정에서 서로 다른 지연 효과를 보정하기 위해 시간적 정렬 절차를 적용하여

만큼 이동시킨다. 아래 표 4의 알고리즘 2는 는 서로 다른 송수신 빔쌍들로부터 들어온 수신된 신호들의 시간적 정렬 절차를 통해 파일럿 수신을 위한 프로세스의 일 예를 나타낸다,

The process of obtaining is by applying a temporal alignment procedure to compensate for different delay effects in the pilot reception process.

move as much Algorithm 2 of Table 4 below shows an example of a process for receiving a pilot through a temporal alignment procedure of signals received from different transmit/receive beam pairs.

[Table 4]

Through this method, pilot signals for channel listening can be received in a beam sweeping environment using different directional beams. That is, it is determined that a pilot signal exists only when the maximum value of the normalized autocorrelation signal of signals received through different directional beams is greater than or equal to the threshold value set by the receiving system, and the time at which the received signal is received through the synchronization procedure is determined. can be found

In addition, by aligning signals having different delays through different directional beams in time as if they were received through omni-directional beams, and estimating a channel through the aligned signals, the signals received through different directional beams It is possible to prevent performance degradation that may occur when estimating a channel using

FIG. 17 shows an example of a comparison graph comparing performances of methods for receiving a pilot signal for channel estimation according to the methods described in FIGS. 14 to 16 .

The simulation environment in FIG. 17 is shown in Table 5 below.

[Table 5]

The channel estimation performance was analyzed through the OMP algorithm with 8 RS subcarriers. The index of performance analysis can be defined as Equation 25 below as NMSE (normalized mean square error).

는 추정한 채널 값을 의미한다.in Equation 25

denotes an estimated channel value.

(a) of FIG. 17 shows OMP-based channel estimation performance according to threshold values between 0.1 and 0.9 at different SNRs when the method described above is used.

As the threshold value becomes smaller, channel estimation is possible through more RSs. Depending on each SNR, the threshold value at which the channel estimation performance is optimized is different.

From the left of the optimization point, there is a section where the spatial-temporal alignment is not perfect due to incorrect estimation. In the right section of the optimization point, the number of RSs used for channel estimation is reduced, which is a major factor in performance deterioration. For this reason, the optimal threshold represents a different point depending on the SNR.

17(b) shows the performance of 'Ginie-aided' implemented under the assumption that the method proposed by the present invention through the 0MP-based channel estimator according to the SNR and the relative delay between pilot beams is known when the method is not applied. is a comparison.

In the process of channel estimation, the method described in the present invention (hereinafter, referred to as ST-alignment) is essential for channel estimation in the process of cell search in which pilots are transmitted through different directional beams, and is close to 'Ginie-aided'. can confirm.

However, it can be confirmed that there is a serious performance degradation in frequency selective channel estimation without the ST-alignment of the present invention.

18 is a diagram illustrating an example of an IDFT (Inverse Discrete Fourier Transform) value of a pilot signal.

As shown in FIG. 18, in order to estimate a channel using the pilot signals described in FIGS. 13 to 17, IDFT is performed on pilot signals known to the terminal and the base station in units of frequency resource blocks at a specific time point.

Hereinafter, the IDFT value of the pilot signal known by the terminal and the base station is referred to as St, and is expressed as in Equation 26 below.

The terminal can estimate a channel from the received pilot signal using a pilot signal known to both the terminal and the base station.

In the present invention, F and W represent analog coefficients of Tx/Rx, respectively. The UE needs to know the values of F and W in order to calculate Ht representing channel information.

At this time, the values of F and W may be defined using the columns of the DFT matrix, and the base station may transmit the values of F and / or W through higher layer signals (e.g. RRC or MAC-CE) specific to the UE or cell. have.

Signal Model

,

안테나 수를 가지고 있으며 같은

RF 체인 수를 가지고 있다.In the present invention, an OFDM millimeter communication system is considered in a hybrid MIMO system. base station and terminal respectively

,

has the same number of antennas as

It has the number of RF chains.

,

개의 트레이닝 빔을 사용하며, 각각의 트레이닝 빔은

,

로 이루어져 있다.In order to estimate the channel, the transmitter and receiver are respectively

,

training beams are used, and each training beam is

,

Consists of

을 만족한다. 트레이닝 구간동안 송신단은

를 하나씩 사용하며 각각의 송신 트레이닝 빔은 트레이닝 수신 빔

로 받아진다.Each training beam is for all p,q

satisfies During the training period, the transmitter

are used one by one, and each transmit training beam is a training receive beam

is accepted as

로 이루어져 있고

로 나타나며

구간은 CP (Cyclic Prefix)를

구간은 수학식 25의 주파수 도메인의 파일럿 심볼의 IDFT 값을 나타낸다.In the time domain, the pilot symbol is length

is made up of

appears as

The section uses CP (Cyclic Prefix)

The interval represents the IDFT value of the pilot symbol in the frequency domain of Equation 25.

의 RF 체인 수를 가지고 있기 때문에 p번째 송신 트레이닝 빔에 대한 수신 벡터

를 얻어 낼 수 있다. 이 때, 아래 수학식 27을 만족한다.the receiver is

Since it has the number of RF chains of , the receive vector for the p-th transmit training beam

can be obtained At this time, Equation 27 below is satisfied.

는 아래 수학식 28과 같이 나타낼 수 있다.

Can be expressed as in Equation 28 below.

이고,

는 채널 행렬을 의미하고, 지연을 나타내는 d는

를 만족한다.In Equation 28, P means transmission power,

ego,

is the channel matrix, and d representing the delay

satisfies

은 AWGN을 의미하며,

분포를 나타낸다.

means AWGN,

represents the distribution.

를 모아서

을 획득할 수 있다. 이때, y_p[n] 는 아래 수학식 29와 같이 나타낼 수 있다.The terminal is the received pilot signal

by collecting

can be obtained. In this case, y _p [n] can be expressed as in Equation 29 below.

을 의미하고, v_p[n] 은

를 의미한다.In Equation 29, W is

, and v _p [n] is

means

에서 송신 트레이닝 빔에 따라 파일럿 신호를 모으면 아래 수학식 30과 같이 수신신호를 획득할 수 있다.At this time,

When the pilot signals are collected according to the transmission training beam in , a received signal can be obtained as shown in Equation 30 below.

is obtained and expressed as:

In this case, Y[n] may be defined as in Equation 31 below.

이고,

를 의미한다.in Equation 31

ego,

means

과 같이 정의하면, M은

과 같이 나타낼 수 있다.At this time, y[n]

Defined as, M is

can be expressed as

의 정의에 따라, y[n]은 아래 수학식 32를 만족한다.

According to the definition of , y [n] satisfies Equation 32 below.

In Equation 32, v[n], Q, Ht and Sn satisfy Equation 33 below.

를 모으면,

를 획득할 수 있으며, Y_t는 아래 수학식 34와 같이 정의될 수 있다.Finally, the terminal

If you collect

Can be obtained, and Y _t can be defined as in Equation 34 below.

는 도 13 내지 도 16에서 설명한 방법을 통해서 수신한 파일럿 신호를 나타내며, 관측 신호 또는 수신 파일럿 신호라 호칭될 수 있다.

represents a pilot signal received through the method described in FIGS. 13 to 16, and may be referred to as an observation signal or a received pilot signal.

Ht represents a channel to be estimated through a pilot signal, and may be referred to as a channel, a channel value, or channel information.

The channel means a physical channel between a base station physical antenna and a terminal physical antenna. For example, when the base station and the terminal each have one panel, and the number of antennas of the terminal panel of the base station is 64 and 8, respectively, the size of Ht may be 64×8.

Meanwhile, when there are two TXRUs in each panel of the base station and the terminal, an effective channel between the base station and the terminal may be 2×2.

In Equation 34, S satisfies Equation 35 below, and Vt satisfies Equation 36 below.

를 만족하게 된다. 이는 파일럿 신호가 equally spaced 조건을 만족하기 때문이다.In addition, the synchronization signal SS is

will be satisfied. This is because the pilot signal satisfies the equally spaced condition.

Channel Model

In the present invention, a frequency selective mmWave channel in which L paths enter at different times is considered. At this time, the d-th delay tap of the channel is as shown in Equation 37 below.

은 scatter의 수를 나타내고

은 각각 l번째 path의 complex gain, delay, AoA, AoD를 의미한다.

는 kroneker delta function이다.in Equation 37

represents the number of scatter

means the complex gain, delay, AoA, and AoD of the lth path, respectively.

is the kroneker delta function.

, 수신단은

로 표기한다.In the present invention, it is assumed that both the transmitter and receiver have a ULA antenna structure, and the array response vector is

, the receiving end

marked with

The array response vector of the ULA structure can be expressed as in Equation 38 below.

는 신호의 파장이며

는 안테나 간격으로

를 만족한다. 채널 complex gain

은 i.i.d random variable이며 분포는

이다.in Equation 38

is the wavelength of the signal

is the antenna spacing

satisfies Channel complex gain

is an iid random variable and the distribution is

to be.

은

구간에서 다 같은 확률을 가진다. AoA와 AoD는

구간에서 uniform분포를 가진다. 수학식 36의 채널 모델은 아래 수학식 39와 같이 다시 정의될 수 있다.each

silver

They all have the same probability in the interval. AoA and AoD are

It has a uniform distribution over the interval. The channel model of Equation 36 may be redefined as Equation 39 below.

In Equation 39, H _a [d], AR and AT may be defined as in Equation 40 below.

Hereinafter, a method of estimating a channel based on a received pilot signal will be described.

19 is a diagram illustrating an example of a method for estimating a channel using a pilot signal to which the present invention can be applied.

Referring to FIG. 19, a terminal may estimate a channel based on a received pilot signal through an orthogonal matching pursuit (OMP) algorithm of a compressive sensing (CS) technique.

Specifically, in Equation 39, H[d] can be expressed as Equation 41 below according to dividing Grid.

, 및

는 Grid에 따른 array response vector를 나타내며, G는 Grid의 수로

를 만족한다.in Equation 41

, and

represents the array response vector according to the grid, and G is the number of grids

satisfies

는 sparse한 매트릭스로 지연이 d인 지점에서의 AoA/AoD에 해당되는 값에만 non-zero값을 가지고 있다.

is a sparse matrix, and has a non-zero value only for the value corresponding to AoA/AoD at the point where the delay is d.

는 Grid를 나눔에 따른 에러 매트릭스이며, Grid로 나눈 채널

는 아래 수학식 42와 같이 정의될 수 있다.

is the error matrix according to dividing the grid, and the channel divided by the grid

Can be defined as in Equation 42 below.

와 관련되어 나타낼 수 있다.In Equation 32, the channel matrix Ht is as shown in Equation 43 below.

can be expressed in relation to

In Equation 43, Et satisfies Equation 44 below.

관계를 이용하면 수학식 34에서

는 아래 수학식 45와 같이 나타낼 수 있다.

Using the relationship, in Equation 34

Can be expressed as in Equation 45 below.

및

는 아래 수학식 46과 같을 수 있다.in Equation 45

and

May be as shown in Equation 46 below.

If Equation 45 is vectorized, it can be expressed as Equation 47 below.

를 추정하기 위해서는 일반적으로 CS 기법을 이용한다.As shown in FIG. 19, the channel in the received signal Yt

In order to estimate , a CS technique is generally used.

를 구해야 한다.In order to estimate a channel using the CS technique, the L-sparse vector is used in Equation 47 through Equation 48 below.

should save

를 계산하기 위해서 아래 표 6의 OMP 알고리즘을 이용할 수 있다.The vector in Equation 48

In order to calculate , the OMP algorithm in Table 6 below can be used.

In this case, F and/or W in Equation 46 may be transmitted from the base station to the terminal through pilot signal configuration information.

[Table 6]

,

의 관계를 만족한다.in table 6

,

satisfies the relationship of

Thereafter, the terminal selects a channel with a better weight among several channels estimated through the pilot signal transmitted from the base station, and reports channel information of the selected channel to the base station.

In this case, the base station Tx beam and the analog coefficient may be mapped 1:1. Accordingly, the base station can implicitly recognize which analog coefficient is an optimized analog coefficient for the UE, based on the channel information reported by the UE.

However, this method has a problem that calculation becomes complicated because the dimension of the received signal and the dimension of the sparse vector are too large.

Therefore, the present invention proposes a method of estimating a channel using the OMP algorithm of the CS technique after removing the characteristics of a pilot signal from a received signal in order to reduce the complexity of such calculation.

20 to 22 are diagrams illustrating an example of a method for estimating a channel using a pilot signal proposed in the present invention.

20 to 22, the terminal removes the characteristics of the transmitted pilot signal transmitted from the base station known to both the terminal and the base station from the received pilot signal, and uses the received pilot signal from which the characteristic of the transmitted pilot signal has been removed. Therefore, the channel can be estimated through the OMP algorithm of the CS technique.

20 is a flowchart illustrating an example of a method for channel estimation proposed in the present invention.

In order to estimate the channel of the received pilot signals received through the method described in FIGS. 13 to 16, the terminal removes the digital characteristics of the transmitted pilot signal, which is a signal known to both the terminal and the base station, from each received pilot signal (S20010). .

That is, the received pilot signal received by the terminal is represented by a combined value of the transmission pilot signal value, which is a digital characteristic known to both the base station and the terminal, channel information of a channel through which the pilot signal is transmitted, and analog characteristics of the received signal. .

Therefore, in order to obtain channel information from such a received pilot signal, a first pilot signal (or first signal characteristic) representing digital characteristics and a second pilot signal (or second signal characteristic) representing analog characteristics are obtained from the received pilot signal. signal characteristics).

To this end, the terminal removes the first pilot signal representing the digital characteristics of the signal known to both the terminal and the base station from the received pilot signal received through step S20010.

Thereafter, the terminal can estimate the channel of the received pilot signal by estimating the complex gain of each channel by removing the second pilot signal from the received pilot signal from which the first pilot signal has been removed through the OMP algorithm of the CS technique (S20020). .

By using this method, by first removing the signal characteristics known to the terminal and the base station from the received pilot signal, the complexity of calculation for obtaining channel information through the OMP algorithm of the CS technique can be greatly reduced.

21 is an example of expressing the method described in FIG. 20 using a mathematical formula.

Referring to FIG. 21, in the present invention, in the received pilot signal received through the first procedure for each training beam pair, a channel from which the first pilot signal of digital characteristics known to the terminal and the base station is removed (hereinafter, an effective channel, After estimating 1 channel state information or the first channel state value), the complex gain of the channel according to each delay tap is estimated using the OMP algorithm of the CS technique to obtain the channel information (or the second channel state information, or A two-step channel estimator performing a second procedure for obtaining a channel state value) is proposed.

In this case, the effective channel refers to a channel from which a first pilot signal having a digital characteristic known to the terminal and the base station is removed from the received pilot signal.

That is, a value obtained by combining the second pilot signal representing analog characteristics and channel information or a value obtained by multiplying Ht by an analog coefficient matrix of a transmitting/receiving terminal may be referred to as an effective channel value.

Specifically, the terminal can estimate the effective channel Z by removing St, which is a transmitted pilot signal known to the terminal and the base station, from the received pilot signal received through the method described in FIGS. 14 to 16 (S21010).

라 정의하면, 수학식 45에서 Yt는 아래 수학식 49와 같이 정의될 수 있다.That is, Z in FIG. 21

When defined as , Yt in Equation 45 can be defined as Equation 49 below.

의 i번째 row 벡터들을

,

및

로 정의하면, yi는 아래 수학식 50과 같이 정의될 수 있다.In Equation 49, Y _t , Z, and

The i-th row vectors of

,

and

When defined as , yi may be defined as in Equation 50 below.

는 i번째 트레이닝 빔 pair에 따른 길이 D의 지연을 가지는 유효 채널이라고 정의한다.in Equation 50

is defined as an effective channel having a delay of length D according to the i th training beam pair.

In this case, the index iT of the transmission training beam and the index iR of the reception training beam can be expressed as Equation 51 below.

의 제 1 절차를 통해 획득한

는 아래 수학식 52와 같이 나타낼 수 있다.

obtained through the first procedure of

Can be expressed as in Equation 52 below.

를 모은다면 전체 트레이닝 빔에 대한 effective channel인

를 획득할 수 있다.on each training beam pair

If , the effective channel for the entire training beam is

can be obtained.

는 아래 수학식 53과 같이 나타낼 수 있다.At this time,

Can be expressed as in Equation 53 below.

Thereafter, channel information may be estimated by removing the second pilot signal, which is an analog characteristic of the received pilot signal, using the OMP algorithm based on the CS technique from the obtained effective channel (S21020).

는

로부터 CS기법의 OMP 알고리즘을 통해 얻을 수 있다. 제 2 절차는 수학식 48과 같이 일반 적인 CS 기법의 OMP 알고리즘과 같이 공식화할 수 있다.Specifically, a sparse matrix

Is

can be obtained through the OMP algorithm of the CS technique. The second procedure can be formulated like the OMP algorithm of a general CS technique as shown in Equation 48.

Equation 54 below shows an example of formulating the second procedure.

In Equation 54, Q can be expressed as Equation 55 below.

In this method, an actual physical channel, not an effective channel, is estimated, and the terminal can directly calculate an analog coefficient of a transmit antenna of the base station from the estimated physical channel.

Through this method, the terminal can estimate an actual physical channel, not an effective channel, and directly calculate an analog coefficient of a transmit antenna of the base station through the estimated physical channel.

In this case, the base station's Tx beam and analog coefficients may no longer be mapped 1:1.

For example, the base station actually transmits the pilot signal through 64 beams, but the terminal has the same effect as if the base station transmitted the pilot signal through 256 beams.

That is, the terminal has an effect of selecting an optimal beam from among more beams than beams through which the base station transmits a pilot signal.

, 및

라는 점에서 차이점이 존재한다.At this time, Equation 54, Equation 48 and Table 6 are

, and

There is a difference in that.

The channel estimation method using the two steps described in FIG. 21 can obtain the same results as the channel estimation method described in FIG. 19 as follows.

Hereinafter, the method proposed in FIG. 21 will be described using a suffix P, and the method described in FIG. 19 will be described using a suffix C.

Specifically, the terminal acquires the same subset as the OMP algorithm of the existing CS technique in the first iteration step 3 of FIGS. 20 and 21 through Equation 56 below.

At this time, iteration step 3 means the third step of the OMP algorithm of Table 6 when the value of k, which is an iteration counter, is 1 in the 0MP algorithm of Table 6.

로부터 획득된다. 등호 (b)는

와

로부터 획득된다. 등호 (c)는

로부터 획득된다.In Equation 56, the equal sign (a) is

is obtained from Equal sign (b) is

Wow

is obtained from Equal sign (c) is

is obtained from

를 만족하기 때문에

이다.from Equation 56

because it satisfies

to be.

Then, when the method described in FIGS. 20 and 21 acquires the same support set as the method described in FIG. 19, the methods in FIGS. 20 and 21 and the method in FIG. 19 have the same sparseness in the k-th iteration as shown in Equation 58 below. solution can be obtained.

를 획득하는 단계를 의미한다.At this time, the k-th iteration is when the iteration counter, which is the 5th step in the OMP algorithm of Table 6, is k,

means the step of obtaining

이면

를 만족할 수 있다.At this time, in Equations 57 and 58

the other side

can be satisfied.

는

의 부분 행렬이고

는

의 부분 벡터이기 때문에 수학식 56과 수학식 57로부터 수학식 58의 등호 (a)를 만족시킬 수 있다.in Equation 57

Is

is a submatrix of

Is

Since it is a partial vector of , it is possible to satisfy the equal sign (a) of Equation 58 from Equations 56 and 57.

In addition, when the method described in FIGS. 20 and 21 obtains the same support set as the method described in FIG. 19, the method of FIGS. 20 and 21 and the method of FIG. 19 obtain an index in the k-th iteration through Equation 59 below. You can get it in step 3.

This means the third step when the value of k, which is an iteration counter in the OMP algorithm of Table 6, satisfies k>1.

In this case, the equal sign (a) in Equation 59 can be obtained from Equations 56 and 57.

와 sparse solution

은 도 19에서 설명한 CS 기법을 통한 OMP 알고리즘을 이용하는 방법과 동일한 결과를 획득할 수 있다.As such, the support set of the method described in FIGS. 20 and 21

with sparse solution

can obtain the same result as the method using the OMP algorithm through the CS technique described in FIG. 19.

At this time, complexity between the method described in FIG. 19 and the method described in FIGS. 20 and 21 may be compared as shown in Equation 60 below.

As described above, in the present invention, the terminal can reduce complexity for channel estimation in the time domain in wideband mmWave communication by first removing the digital characteristics of the received pilot signal.

That is, the present invention estimates the effective channel corresponding to each transmission/reception terminal training beam pair in the first procedure, and estimates the AoD and AoA of each delay lap using the OMP algorithm of the CS method through the second procedure to estimate the channel By doing so, the complexity of channel estimation can be reduced.

FIG. 22 shows an example of a comparison graph comparing performance of channel estimation according to the methods described in FIGS. 20 and 21 .

The simulation environment in FIG. 22 is shown in Table 7 below.

[Table 7]

는 추정한 채널 값을 의미한다.22 is a graph comparing performances of the two-step channel estimation method described in FIGS. 20 and 21 and the channel estimation method described in FIG. 19 . The index of performance analysis is NMSE (normalized mean square error), as shown in Equation 61 below,

denotes an estimated channel value.

The complexity at this time is shown in Equation 62 below.

As shown in FIG. 22, the two methods show the same performance, but the computational complexity of the methods described in FIGS. 20 and 21 is lowered by about 10 ⁴ as in Equation 62.

When a channel is estimated using this method, the same effect as channel estimation received through an omni-directional beam can be obtained.

That is, a plurality of pilot signals transmitted through a plurality of directional beams are aligned identically to pilot signals received through omnidirectional beams, a channel is estimated through the aligned signals, and one channel is selected from among the estimated channels. Therefore, there is an effect that one channel can be selected from among more channels than channels through which pilot signals are transmitted.

For example, when pilot signals are transmitted through 64 channels, the receiving end can estimate 256 channels using the pilot signal reception method and channel estimation method described above, and select one of the 256 estimated channels. There is an effect.

When channel estimation is performed using the pilot signal transmitted through the directional beam described in FIGS. 10 to 22, or when data is transmitted after channel estimation, when there is an obstacle in the direction of the reception beam, data communication may fail. There is a problem with that.

Accordingly, the present invention proposes a beamformer design that maintains a communication link even when a communication failure occurs due to an obstacle in order to solve this problem.

23 is a diagram illustrating an example of a case in which a communication failure occurs due to an obstacle in a MIMO system using a directional beam.

Unlike the Conventional MIMO system that transmits data in all directions, data is transmitted using directional beams through analog beamforming to overcome the degradation of SNR of the received signal due to high path-loss characteristics in millimeter band communication.

However, when an obstacle (e.g. bus, human-body) appears in the direction of the receiving beam when transmitting through a directional beam, the communication link may deteriorate rapidly as shown in FIG. 23, which is referred to as blockage.

Therefore, it is necessary to design a beamformer that maintains a communication link even when blockage occurs by considering not only throughput but also robustness.

24 is a diagram illustrating an example of a double space time space diversity (DSTTD) system of multi-input multi-output (MIMO) to which the present invention can be applied.

Advantages of a multi-input multi-output (MIMO) system can be largely divided into spatial multiplexing and diversity.

A transmission method capable of maximally obtaining transmit diversity is a space time block code (STBC).

Layered STBC (L-STBC) is a transmission method that simultaneously considers a multiplexing gain and a diversity gain in a MIMO system. In the L-STBC system, several antennas are divided into groups that transmit independent streams, and each group passes STBC to a corresponding transmission data stream. DSTTD (Double space time space diversity) shown in FIG. 24 means a system that operates two alamouti's STBCs.

는 Alamouti's STBC로 아래 수학식 63과 같이 나타낼 수 있다.Data stream transmitted through the DSTTD system shown in FIG. 24

Is Alamouti's STBC and can be expressed as Equation 63 below.

는 아래 수학식 64와 같이 orthogonality를 만족한다.At this time, the STBC symbol

satisfies orthogonality as shown in Equation 64 below.

는 아래 수학식 65와 같이 orthogonality를 만족하지 않는다. (

)However, the 4×2 DSTTD symbol matrix

does not satisfy orthogonality as shown in Equation 65 below. (

)

Since there is interference between each STBC symbol, a successive interference cancellation (SIC) reception method such as V-BLAST is required to discover the symbol. It is assumed that the channel is completely known at the receiving end. The MIMO channel shown in FIG. 24 can be expressed as Equation 66 below.

은 n번째 time에 2x4 complex channel matrix를 의미하고,

는 j번째 송신 안테나와 i번째 수신 안테나의 채널을 의미한다.channel in Equation 66

means a 2x4 complex channel matrix at the nth time,

denotes channels of the j-th transmit antenna and the i-th receive antenna.

If the channel does not change while the terminal receives the received signals y(2n) and y(2n+a) of Equation 67 below, the received signal during the two symbol times can be expressed as Equation 68 below.

의 AWGN (Additive White Gaussian Noise)를 나타낸다. 수학식 68은 아래 수학식 69와 같이 effective channel linearlized form형태로 표현할 수 있다.The 2x2 matrix in Equation 68 is the variance

AWGN (Additive White Gaussian Noise). Equation 68 can be expressed in an effective channel linearized form as shown in Equation 69 below.

As described above, when the signal is modeled, the receiving end can decode the signal through a receiving process such as ZF-SIC and MMSE-SIC as shown in Equation 70 below.

25 is a diagram illustrating an example of a DSTTD system to which an antenna shuffling technique to which the present invention can be applied is applied.

The performance of a DSTTD system may be sensitive to spatial correlation between transmit and receive antennas. Therefore, the antenna shuffling technique can be applied to the DSTTD system as shown in FIG. 25 to improve performance in response to spatial correlation.

As shown in FIG. 25, an additional antenna shuffling process for performance improvement may be included in the existing 4x2 DSTTD system.

Each transmitted stream goes through the antenna shuffling process of the ASG-Preprocessing block of FIG. 25 after the STBC Encoder.

To improve performance, the receiving end determines a shuffling matrix based on characteristics sensitive to spatial correlation and feeds back the index to the transmitting end.

Here, the matrix set for shuffling is as shown in Equation 71 below. The above scheme can be viewed as a closed-loop system because an additional feedback process at the receiving end is added.

은

로부터 획득될 수 있다.In Equation 71, the effective channel

silver

can be obtained from

26 and 27 are diagrams illustrating an example of a DSTTD system to which a beam shuffling technique proposed in the present invention is applied.

Referring to FIGS. 26 and 27 , an open-loop transmission system robust against blockage can be designed based on a DSTTD system in a hybrid MIMO communication environment using mmWave frequencies.

Specifically, while extending to a hybrid MIMO structure based on the structure of the DSTTD system described in FIG. 24, the transceiver structure may be changed as follows.

In the existing MIMO structure, the signal from the STBC can be directly transmitted to each antenna domain, but as the hybrid MIMO structure is extended, the signal can be transmitted to the analog beam domain instead of the antenna domain.

As shown in FIG. 26, assume a system in which the transmitting end is composed of two panel structures and the receiving end is composed of one panel structure.

At this time, the number of panels of the transmitting end and the receiving end may be extended. Each panel covers a different coverage in the angle region.

In FIG. 26, the transmitting end panel #1 covers the 0 to 90 area, the panel #2 can cover the 90 to 180 area, and the receiving end panel can cover the 0 to 180 area.

At this time, the transmitter has two RF chains connected to each panel and performs analog beamforming in the same direction. The analog beam in Panel #1 points in the channel direction toward the upper receive beam, and the analog beam in Panel # 2 points in the channel direction toward the lower receive beam.

At this time, in the Hybrid BF structure shown in FIG. 25, when two signals from the STBC are transmitted in one reception beam direction, if blockage occurs in the reception beam direction, it is difficult to receive a complete signal in the reception beam in the blockage direction. Therefore, this method does not obtain the diversity effect through STBC#.

Therefore, in order to be robust against the blockage effect, a beam shuffling method through a preprocessing block is required as shown in FIG. 25 .

As shown in FIG. 25, by transmitting the streams that have passed through the STBC encoder in different reception beam directions through beam shuffling, one coded signal can enter both reception beams.

Therefore, even if blockage occurs in one reception beam direction, a signal can be received through another reception beam, and thus a diversity effect can be obtained.

를 구성하는 matrix는 아래 수학식 72와 같이 나타낼 수 있다.At this time, shuffling between transmit beams heading in the same analog receive beam direction has no effect on blockage.

The matrix constituting may be expressed as in Equation 72 below.

set 안에서 임의의 수신단과 약속된 matrix를 사용하여 open-loop시스템에 약속된 matrix를 적용할 수 있다.In Equation 72, the preprocessor is

In the set, the promised matrix can be applied to the open-loop system by using the promised matrix with any receiving end.

The purpose of the beam shuffling proposed in FIG. 26 of the present invention is to allow signals that have passed through STBC to be received as different reception beams, and to respond to the spatial correlation described in FIG. may have the following differences.

The present invention is a beam shuffling method for selecting an analog beam while extending an antenna shuffling method considering channel correlation for performance improvement of a DSTTD system in an existing MIMO channel to a hybrid MIMO structure.

At this time, the following analog beam operation is required.

In FIG. 26, at least two beams must be used in one panel of the transmitter. At this time, two streams should be received as one reception beam by forming two transmission beams in the same direction.

If one analog beam is used in one panel, two beams in the same direction cannot be formed because the range that can be covered is different for each panel, which guarantees that each stream transmits a signal as a reception beam. can't be

In addition, as shown in FIG. 26, the reception beams must point to different beam directions in order to prevent blockage from the proposed beam shuffling technique.

If Rx beams are formed in one direction, it may be difficult to receive signals due to the effect of blockage on all Rx beams.

Through this method, even if an obstacle occurs in a beam of a specific panel, the terminal can receive data without loss through a beam of another panel.

That is, the present invention can design an open-loop transmission method robust against blockage suitable for the mmWave communication hybrid MIMO structure.

Through this, the transmission method proposed in the present invention shows more robust performance in a blockage environment.

FIG. 27 shows an example of a comparison graph comparing performances of the transmission methods described in FIG. 26 .

The simulation environment in FIG. 27 is shown in Table 8 below.

[Table 8]

System performance can be checked through BER (Bit Error Rate).

27 shows the BER performance of the existing open-loop transmission technique and the proposed beam shuffling-based DSTTD transmission technique in the simulation environment of Table 6.

By confirming that the proposed beam shuffling-based DSTTD technique has the best ber performance than comparison systems, it is confirmed that it is robust against blockage. Conventional DSTTD without beam shuffling can confirm performance degradation due to blockage loss.

28 is a flowchart illustrating an example of a method for estimating a channel using a pilot signal proposed in the present invention.

Referring to FIG. 28, a terminal receives a plurality of pilot signals for channel estimation through a plurality of directional beams, corrects and aligns the received plurality of pilot signals as if they were received through omnidirectional beams, and estimates a channel. can

Specifically, the terminal receives a plurality of pilot signals for channel estimation through a plurality of beams from the base station (S28010).

In this case, the plurality of pilot signals may have a symbol frame and symbol structure as shown in FIG. 12 .

Thereafter, the terminal acquires at least one pilot signal having an SNR value equal to or greater than the threshold among a plurality of pilot signals, and acquires time synchronization of the obtained at least one pilot signal (S28020).

At this time, time synchronization can be obtained through the method described in FIGS. 13 and 14 .

Thereafter, the terminal acquires time synchronization and arranges in time as if at least one pilot signal having an accurately known reception time was received through an omnidirectional beam (S28030).

In this case, alignment of at least one pilot signal in time may be performed through the spatial alignment procedure described in FIGS. 15 and 16 .

Thereafter, the terminal calculates and compensates for a frequency offset, which is a propagation delay due to the CFO of each of the at least one pilot signal, in order to compensate for the path delay of the at least one pilot signal (S28040).

In this case, the frequency offset compensation of each of the at least one pilot signal may be performed through the temporal alignment procedure described in FIGS. 15 and 16 .

Thereafter, the terminal may perform channel estimation based on at least one pilot signal for which the frequency offset value is compensated, and may select one channel from among the estimated channels and report channel information to the base station (S28050).

At this time, channel estimation may be performed through the 1-step channel estimation technique described in FIG. 19 or the 2-step channel estimation technique described in FIGS. 20 and 21 .

29 is a diagram illustrating an example of an internal block diagram of a wireless device to which the present invention can be applied.

Here, the wireless device may be a base station and a terminal, and the base station includes both a macro base station and a small base station.

As shown in FIG. 29, a base station 2910 and a UE 2920 include communication units (transmitting and receiving units, RF units, 2913 and 2923), processors 2911 and 2921, and memories 2912 and 2922.

In addition, the base station and the UE may further include an input unit and an output unit.

The communication units 2913 and 2923, processors 2911 and 2921, input units, output units, and memories 2912 and 2922 are functionally connected to perform the method proposed in this specification.

When the communication unit (transmitter/receiver unit or RF unit, 2913, 2923) receives information generated from the PHY protocol (Physical Layer Protocol), it transfers the received information to the RF spectrum (Radio-Frequency Spectrum), and performs filtering and amplification. ) and the like are transmitted to the antenna. In addition, the communication unit functions to move a radio frequency signal (RF signal) received from an antenna to a band that can be processed by the PHY protocol and performs filtering.

And, the communication unit may also include a switch function for switching such transmission and reception functions.

The processors 2911 and 2921 implement functions, processes and/or methods proposed in this specification. Layers of the air interface protocol may be implemented by a processor.

The processor may be expressed as a control unit, a controller, a control unit, a computer, or the like.

The memories 2912 and 2922 are connected to the processor and store protocols or parameters for performing an uplink resource allocation method.

The processors 2911 and 2921 may include application-specific integrated circuits (ASICs), other chipsets, logic circuits, and/or data processing devices. Memory may include read-only memory (ROM), random access memory (RAM), flash memory, memory cards, storage media, and/or other storage devices. The communication unit may include a baseband circuit for processing a radio signal. When the embodiment is implemented as software, the above-described technique may be implemented as a module (process, function, etc.) that performs the above-described functions.

A module can be stored in memory and executed by a processor. The memory may be internal or external to the processor, and may be coupled with the processor in a variety of well-known means.

An output unit (display unit or display unit) is controlled by a processor and outputs information output from the processor together with a key input signal generated by the key input unit and various information signals from the processor.

Furthermore, although each drawing has been divided and described for convenience of explanation, it is also possible to design to implement a new embodiment by merging the embodiments described in each drawing. And, according to the needs of those skilled in the art, designing a computer-readable recording medium in which programs for executing the previously described embodiments are recorded is also within the scope of the present invention.

In the direction-based device search method according to the present specification, the configuration and method of the above-described embodiments are not limitedly applicable, but all or part of each embodiment is selectively applied so that various modifications can be made. It may be configured in combination with.

Meanwhile, the direction-based device search method of the present specification can be implemented as a processor-readable code on a processor-readable recording medium included in a network device. The processor-readable recording medium includes all types of recording devices in which data readable by the processor is stored. Examples of the processor-readable recording medium include ROM, RAM, CD-ROM, magnetic tape, floppy disk, optical data storage device, etc., and also include those implemented in the form of carrier waves such as transmission through the Internet. . In addition, the processor-readable recording medium is distributed in computer systems connected through a network, so that processor-readable codes can be stored and executed in a distributed manner.

In addition, although preferred embodiments of the present specification have been illustrated and described above, the present specification is not limited to the specific embodiments described above, and the technical field to which the present invention belongs without departing from the gist of the present invention claimed in the claims. Of course, various modifications are possible by those skilled in the art, and these modifications should not be individually understood from the technical spirit or prospect of the present invention.

And, in this specification, both the product invention and the method invention are described, and the description of both inventions can be supplementarily applied as needed.

2910: base station 2920: terminal
2911: processor 2922: memory
2911: RF unit

Claims (18)

In a method for performing channel estimation by a terminal in a wireless communication system,
Receiving, from a base station, a plurality of pilot signals for channel estimation based on a plurality of beam pairs between a base station transmission beam and a terminal reception beam;
Each of the plurality of beam pairs (i) is associated with one of the plurality of pilot signals, and (ii) has different directivity;
Acquiring a reception time point of at least one pilot signal among the plurality of pilot signals based on a reception window for receiving the pilot signal;
The receiving window is set so that (i) the time length of the receiving window is twice the length of an orthogonal frequency-division multiplexing (OFDM) symbol, and (ii) the time length between adjacent receiving windows cross each other by half;
performing (i) spatial alignment and (ii) temporal alignment on the at least one pilot signal;
Based on the spatial alignment, (i) calculating a path delay for a path difference between the at least one pilot signal according to beam sweeping based on the plurality of beam pairs, and (ii) calculating the path delay for the at least one pilot signal. By compensating the calculated path delay value for , the at least one pilot signal is aligned in time,
Based on the time alignment, (i) calculating a propagation delay of each of the at least one pilot signal aligned in time along a plurality of paths through which pilot signals are transmitted from the base station to the terminal; compensating the calculated propagation delay for the at least one pilot signal aligned in phase, so that the at least one pilot signal aligned in time is further aligned in time; and
and performing channel estimation based on at least one pilot signal aligned based on the (i) spatial alignment and (ii) temporal alignment. According to claim 1,
calculating a delay between the at least one pilot signal; and
and aligning the at least one pilot signal in time according to a transmission order in the base station based on the calculated delay between the at least one pilot signal. According to claim 2,
The calculated delay between the at least one pilot signal includes the number of OFDM symbols between the at least one pilot signal and a transmission delay. According to claim 1,
comparing the plurality of pilot signals with a threshold value; and
and determining the at least one pilot signal greater than the threshold among the plurality of pilot signals. The method of claim 1, wherein performing channel estimation comprises:
calculating a channel state value of each of the at least one pilot signal; and
and selecting the highest channel state value based on the calculated channel state value. The method of claim 5, wherein calculating the channel state value comprises:
obtaining a first channel state value based on a first signal characteristic value in each of the at least one pilot signal; and
And obtaining a second channel state value from the first channel state value using a specific algorithm. According to claim 6,
The first signal characteristic value is a value representing a digital characteristic of a pilot signal transmitted from the base station,
The first channel state value is obtained by removing the first signal characteristic value from each of the at least one pilot signal. According to claim 6,
Wherein the specific algorithm is an orthogonal matching pursuit (OMP) algorithm. According to claim 1,
The method further comprising receiving configuration information of the at least one pilot signal from the base station. In a terminal performing channel estimation in a wireless communication system,
A radio unit (Radio Frequency Unit) for transmitting and receiving radio signals to and from the outside;
a processor functionally coupled to the radio unit; and
at least one computer memory operably connectable to the processor and storing instructions that, when executed by the processor, perform operations;
These actions are
Receiving, from a base station, a plurality of pilot signals for channel estimation based on a plurality of beam pairs between a base station transmission beam and a terminal reception beam;
Each of the plurality of beam pairs (i) is associated with one of the plurality of pilot signals, and (ii) has different directivity;
Acquiring a reception time point of at least one pilot signal among the plurality of pilot signals based on a reception window for receiving the pilot signal;
The receiving window is set so that (i) the time length of the receiving window is twice the length of an orthogonal frequency-division multiplexing (OFDM) symbol, and (ii) the time length between adjacent receiving windows cross each other by half;
performing (i) spatial alignment and (ii) temporal alignment on the at least one pilot signal;
Based on the spatial alignment, (i) calculating a path delay for a path difference between the at least one pilot signal according to beam sweeping based on the plurality of beam pairs, and (ii) calculating the path delay for the at least one pilot signal. By compensating the calculated path delay value for , the at least one pilot signal is aligned in time,
Based on the time alignment, (i) calculating a propagation delay of each of the at least one pilot signal aligned in time along a plurality of paths through which pilot signals are transmitted from the base station to the terminal; compensating the calculated propagation delay for the at least one pilot signal aligned in phase, so that the at least one pilot signal aligned in time is further aligned in time; and
and performing channel estimation based on at least one pilot signal aligned based on the (i) spatial alignment and (ii) temporal alignment. 11. The method of claim 10, wherein the operations,
calculating a delay between the at least one pilot signal; and
and aligning the at least one pilot signal in time according to a transmission order in the base station based on the calculated delay between the at least one pilot signal. According to claim 11,
The calculated delay between the at least one pilot signal includes the number of OFDM symbols between the at least one pilot signal and a transmission delay. 11. The method of claim 10, wherein the operations,
comparing the plurality of pilot signals with a threshold value; and
and determining the at least one pilot signal greater than the threshold among the plurality of pilot signals. The method of claim 10, wherein the operations,
calculating a channel state value of each of the at least one pilot signal;
The terminal further comprising the step of selecting the highest channel state value based on the calculated channel state value. 11. The method of claim 10, wherein the operations,
obtaining a first channel state value based on a first signal characteristic value in each of the at least one pilot signal;
The terminal further comprising obtaining a second channel state value from the first channel state value by using a specific algorithm. According to claim 15,
The first signal characteristic value is a value representing a digital characteristic of a pilot signal transmitted from the base station,
The first channel state value is obtained by removing the first signal characteristic value from each of the at least one pilot signal. According to claim 15,
The terminal, characterized in that the specific algorithm is an orthogonal matching pursuit (OMP) algorithm. 11. The method of claim 10, wherein the operations,
The terminal further comprising receiving configuration information of the at least one pilot signal from the base station.

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