KR102128327B1 - Llc resonant converter and operation method thereof - Google Patents

Abstract

This embodiment is an LLC resonant converter including a primary side circuit for converting a DC input voltage Vin to an AC voltage of a predetermined level and a secondary side circuit for rectifying the converted AC voltage to supply a DC output voltage Vo. The method of claim 1, wherein the primary circuit includes: a first resonant circuit unit including a transformer T1, a resonant inductor Lpl1 and a resonant capacitor Cr1; A second resonant circuit unit including a transformer T2, a resonant inductor Lpl2, and a resonant capacitor Cr2; And a plurality of switching elements that convert the DC input voltage Vin to an AC voltage through a switching operation and apply it to the first resonant circuit unit and the second resonant circuit unit, wherein the secondary-side circuit includes the transformer T1 and the transformer T2 It includes a rectifying unit including a plurality of diodes that are connected to the secondary coil of the rectifying the AC voltage converted by the primary circuit, the resonance capacitor Cr1 and the resonance capacitor Cr2 is the tolerance of the resonance element (Tolerance) In order to prevent the current imbalance of the secondary-side circuit, an LLC resonant converter having one end and the other end in common connection is provided.

Description

LLC Resonant Converter and its operation method{LLC RESONANT CONVERTER AND OPERATION METHOD THEREOF}

The present invention relates to an LLC resonant converter, and more particularly, to an LLC resonant converter having a wide input/output voltage control range and a method for operating the same.

The contents described in this section merely provide background information for this embodiment, and do not constitute a prior art.

2. Description of the Related Art Recently, demand for a power conversion device having a wide input/output control range is increasing in various application fields such as a microgrid, an energy storage system (ESS), an electric vehicle (EV charger), and a forklift charging system.

Accordingly, in the case of an E-mobility-related charging system, research and development of a DC-DC converter power converter having a wide input/output control range of 2 times or more to charge various types of batteries has been actively conducted.

In particular, in order to achieve high integration and high efficiency, research and development of an LLC resonant converter capable of high-frequency switching and zero voltage switching (ZVS) under all output voltage and load conditions has been actively conducted.

1 is a view showing a full-bridge LLC resonant converter according to the prior art, and FIG. 2 is a view showing the gain characteristics of the full-bridge LLC resonant converter of FIG. 1.

In FIG. 2, the horizontal axis represents the switching frequency, the vertical axis represents the DC input/output voltage ratio (ie, gain), and the load resistance values of the resistors R1 to R8 are'R8<R7<R6...<R1'.

Referring to FIG. 1, the full-bridge LLC resonant converter 100 includes switches Q1 and Q2 of the primary circuit 110 and switches Q3 and Q4 at a duty ratio of 50%. By alternately switching operations, a voltage having the same magnitude as the input voltage Vin is applied between the terminals a and b to be transferred to the secondary circuit 130. Specifically, when the switches Q1 and Q4 are turned on and the switches Q2 and Q3 are turned off, a voltage of +Vin is applied between the terminals a and b. Conversely, when the switches Q1 and Q4 are turned off and the switches Q2 and Q3 are turned on, a voltage of -Vin is applied between the terminals a and b.

Referring to FIG. 2, the full-bridge LLC resonant converter 100 may control gains by controlling switches Q1 to Q4 in a switching frequency modulation method. Specifically, the gain characteristic of the full-bridge LLC resonant converter 100 is higher as the switching frequency is lowered to the minimum control frequency fmin based on the resonant frequency fr, and lower as the switching frequency is increased. The full-bridge LLC resonant converter 100 can control the input/output voltage over a wide range using such gain characteristics.

The gain characteristic of the LLC resonant converter 100 is strongly load-dependent, and thus the gain characteristic is different according to the size of the load. Specifically, when the LLC resonant converter 100 is light-loaded (for example, V(f, R1) in FIG. 2), since the gain characteristic changes according to the switching frequency is large, the control range of the input/output voltage is wide. Lose. However, when the LLC resonant converter 100 operates at heavy load (for example, V(f, R7)), the change in the gain characteristic according to the switching frequency is small, so the control range of the input/output voltage is narrowed.

The magnetization inductance (Lpm1) of the transformer (T1) is designed to be very small to improve the gain characteristics, but in this case, there is a problem that the conduction loss increases and the efficiency decreases as the magnetizing current increases.

In addition, in order to reduce the load dependence of the gain characteristics, an LLC resonant converter that operates in parallel has been proposed, but in this case, the gain according to the tolerance of the resonant elements of individual converters (eg, transformer leakage inductance and magnetization inductance, resonant capacitor, etc.) There is a problem that current imbalance may occur in the secondary circuit due to the difference.

3 is a view showing a 6-switch LLC resonant converter according to the prior art, and FIG. 4 is a view showing a gain characteristic of the 6-switch LLC resonant converter of FIG. 3.

The 6-switch LLC resonant converter of FIG. 3 is disclosed in Korean Registration No. 10-1837603 (Registration Date: 2018.03.06).

In addition, in Figure 4, the horizontal axis represents the switching frequency, and the vertical axis represents the DC input/output voltage ratio (ie, gain).

3 and 4, the 6-switch LLC resonant converter 300, in order to prevent the current imbalance of the secondary circuit 330, in the 0th mode (Mode 0) and the first mode (Mode 1), The first resonant circuit unit 311 and the second resonant circuit unit 313 operate in series, and the secondary windings NS11, NS22, NS12, and NS21 of each transformer of the secondary circuit 330 operate in parallel, respectively, and rectifying diodes D1 and D3, D4 and D6) transmit power by supplying the rectified resonant current to the load.

In addition, the 6-switch LLC resonant converter 300, in the second mode (Mode 2) and the third mode (Mode 3), the first resonant circuit unit 311 and the second resonant circuit unit 313 operates in parallel, The secondary windings NS11, NS21, NS12, and NS22 of each transformer of the rectifying part of the secondary circuit 330 operate in series, respectively, and transmit power by supplying the rectified resonant current through the rectifying diodes D2 and D5 to the load.

Therefore, even if there are tolerances in the resonant elements of the resonant circuits 311 and 313, the diode output rectifier operates stably without current imbalance in each mode, and the LLC resonant converter 300 has a wide input/output control range (input voltage: Vin/ 16 ~ Vin, output voltage: Vo ~ 16Vo).

However, in each mode of the resonant circuit units 311 and 313, the primary or secondary sides of the transformers T1 and T2 operate in series, and thus, when switching frequency modulation control, leakage inductances Lpl1 and Lpl2 and magnetization inductances Lpm1, Due to Lpm2), there is a problem that it is difficult to obtain desired gain characteristics.

In addition, in the case of low-voltage large-current output, the number of diodes required for the secondary-side rectifier increases, and in the case of the second mode (Mode 2) and the third mode (Mode 3), some diodes D2 and D5 according to the secondary-side series operation There is a problem in that it must bear all the large currents.

To solve this problem, the primary and secondary rectifiers of the resonant circuit units 311 and 313 can be operated in parallel by changing the primary side switching pattern and the primary and secondary winding polarities of the transformers T1 and T2. However, in this case, there is a problem in that current imbalance may be largely generated in the secondary circuit according to the tolerance of the resonance element.

This embodiment is to provide an LLC resonant converter capable of controlling input and output in a wide range without current imbalance in the secondary circuit and a method of operating the same.

According to an aspect of this embodiment, the LLC resonant converter including a primary side circuit for converting a DC input voltage Vin to an AC voltage of a predetermined level and a secondary side circuit for rectifying the converted AC voltage to supply a DC output voltage Vo ( In the resonant converter), the primary circuit includes: a first resonant circuit unit including a transformer T1, a resonant inductor Lpl1 and a resonant capacitor Cr1; A second resonant circuit unit including a transformer T2, a resonant inductor Lpl2, and a resonant capacitor Cr2; And a plurality of switching elements that convert the DC input voltage Vin to an AC voltage through a switching operation and apply it to the first resonant circuit unit and the second resonant circuit unit, wherein the secondary-side circuit includes the transformer T1 and the transformer T2. It includes a rectifying unit including a plurality of diodes that are connected to the secondary coil of the rectifying the AC voltage converted by the primary circuit, the resonance capacitor Cr1 and the resonance capacitor Cr2 is the tolerance of the resonance element (Tolerance) In order to prevent the current imbalance of the secondary-side circuit, an LLC resonant converter having one end and the other end in common connection is provided.

According to another aspect of this embodiment, the LLC resonant converter including a primary side circuit for converting a DC input voltage Vin into an AC voltage of a predetermined level and a secondary side circuit for rectifying the converted AC voltage to output a DC output voltage Vo ( In the resonant converter), the primary circuit includes: a first resonant circuit unit including a transformer T1, a resonant inductor Lpl1 and a resonant capacitor Cr1; A second resonant circuit unit including a transformer T2, a resonant inductor Lpl2, and a resonant capacitor Cr2; And a plurality of main switching elements and at least one auxiliary switching element applied to the first resonant circuit unit and the second resonant circuit unit by converting the DC input voltage Vin into an AC voltage through a switching operation. , A rectifying unit including a plurality of diodes connected to the secondary coils of the transformer T1 and the transformer T2 to rectify the AC voltage converted by the primary circuit, wherein the resonant capacitor Cr1 and the resonant capacitor Cr2 resonate An LLC resonant converter in which one end and the other end are commonly connected to each other is provided to prevent current imbalance of the secondary circuit due to element tolerance.

The LLC resonant converter according to this embodiment has an effect of improving converter efficiency by increasing converter capacity through primary and secondary current sharing and controlling the output voltage at a constant switching frequency.

1 is a view showing a full-bridge LLC resonant converter according to the prior art.
FIG. 2 is a diagram showing the gain characteristics of the full-bridge LLC resonant converter of FIG. 1.
3 is a view showing a 6-switch LLC resonant converter according to the prior art.
FIG. 4 is a diagram showing gain characteristics of the 6-switch LLC resonant converter of FIG. 3.
5 is a view showing an LLC resonant converter according to an embodiment of the present invention.
FIG. 6 is a view showing gain characteristics of the LLC resonant converter of FIG. 5.
7 is a view exemplarily showing a secondary side rectifying part of the LLC resonant converter of FIG. 5.
FIG. 8A is a diagram illustrating an operation waveform when the LLC resonant converter of FIG. 5 operates as an example of the first mode.
8B to 8E are diagrams showing a current path when the LLC resonant converter of FIG. 5 operates as an example of the first mode.
9A is a diagram illustrating an operation waveform when the LLC resonant converter of FIG. 5 operates as another example of the first mode.
9B to 9E are diagrams showing a current path when the LLC resonant converter of FIG. 5 operates as another example of the first mode.
FIG. 10A is a diagram illustrating an operation waveform when the LLC resonant converter of FIG. 5 operates as an example of the second mode.
10B to 10E are diagrams showing a current path when the LLC resonant converter of FIG. 5 operates as an example of the second mode.
11 is a view showing an application example of the LLC resonant converter of FIG. 5.
12 is a graph showing the experimental results of comparing the operating performance of the LLC resonant converter of FIG. 5 and the LLC resonant converter according to the prior art.
13 is a view showing an LLC resonant converter according to another embodiment of the present invention.
14 is a view showing gain characteristics of the LLC resonant converter of FIG. 13.
15A is a diagram illustrating an operation waveform when the LLC resonant converter of FIG. 13 operates as an example of the first mode.
15B to 15G are diagrams showing a current path when the LLC resonant converter of FIG. 13 operates as an example of the first mode.
FIG. 16A is a diagram illustrating an operation waveform when the LLC resonant converter of FIG. 13 operates as an example of the second mode.
16B to 16E are diagrams showing a current path when the LLC resonant converter of FIG. 13 operates as an example of the second mode.
17 is a diagram illustrating an application example of the LLC resonant converter of FIG. 13.
18 is a graph showing experimental results comparing the performance of the LLC resonant converter of FIG. 13 and the LLC resonant converter according to the prior art.

Hereinafter, some embodiments of the present invention will be described in detail through exemplary drawings. It should be noted that in adding reference numerals to the components of each drawing, the same components have the same reference numerals as possible, even if they are displayed on different drawings. In addition, in describing the embodiments of the present invention, when it is determined that detailed descriptions of related well-known structures or functions may obscure the subject matter of the present invention, detailed descriptions thereof will be omitted.

In addition, in describing the components of the present invention, terms such as first, second, A, B, (a), and (b) may be used. These terms are only for distinguishing the component from other components, and the nature, order, or order of the component is not limited by the term. Throughout the specification, when a part is'included' or'equipped' a component, this means that other components may be further included rather than excluded other components unless specifically stated to the contrary. . In addition,'… The terms "unit" and "unit" mean a unit that processes at least one function or operation, which may be implemented by hardware or software or a combination of hardware and software.

Hereinafter, exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings.

5 is a view showing an LLC resonant converter according to an embodiment of the present invention, and FIG. 6 is a view showing a gain characteristic of the LLC resonant converter of FIG. 5.

7 is a view exemplarily showing a secondary side rectifying part of the LLC resonant converter of FIG. 5.

First, referring to FIG. 5, the LLC resonant converter 500 may include six switching elements Q1 to Q6, a first resonant circuit unit 511 and a second resonant circuit unit 513 as the primary side circuit 510. have. In addition, the LLC resonant converter 500 may include a rectifier as the secondary circuit 530.

In the primary-side circuit 510, switching elements Q1 and Q2 are connected in series to a first bridge arm Br1, switching elements Q3 and Q4 are connected in series to a second bridge arm Br2, and switching elements Q5 and Q6 are connected in series. 3 The bridge arm Br3 is connected in parallel between the + terminal and the-terminal of the input power Vin.

The six switching elements Q1 to Q6 convert a DC input voltage into an AC voltage through a switching operation and apply a function to the resonant circuit, and may be implemented as a transistor.

The first resonant circuit unit 511 is connected between the node a between the source of the switching element Q1 and the drain of Q2 and the node b between the source of the switching element Q3 and the drain of Q4.

Further, the second resonant circuit unit 513 is connected between the node b between the source of the switching element Q3 and the drain of Q4, and the node c between the source of the switching element Q5 and the drain of Q6.

The first resonance circuit unit 511 may include a primary coil of the first transformer T1, a resonance inductor Lpl1, and a resonance capacitor Cr1. According to an aspect of this embodiment, the resonant inductor Lpl1 may be a leakage inductor of the first transformer T1.

The second resonance circuit unit 513 may include a primary side of the second transformer T2, a resonance inductor Lpl2, and a resonance capacitor Cr2. According to an aspect of this embodiment, the resonant inductor Lpl2 may be a leakage inductor of the second transformer T2.

The first and second resonant circuit units 511 and 513 perform a function of transmitting the AC voltage transformed according to the primary and secondary turns ratios of each transformer T1 and T2 to the secondary circuit 530.

In FIG. 5, the LLC resonant converter 500 shows that each transformer T1 and T2 is individually used, but this is exemplary, and the LLC resonant converter 500 may be implemented using a single transformer.

One end of the first and second resonant capacitors Cr1 and Cr2 is commonly connected to the node b between the source of the switching element Q3 and the drain of Q4, and the other ends of the first and second resonant capacitors Cr1 and Cr2 are connected in common. do. Here, the capacitances of the first and second resonant capacitors Cr1 and Cr2 may be the same (ie, Cr1=Cr2=C).

By connecting the resonant capacitors (Cr1, Cr2) in common, the tolerance to the resonant elements (e.g., leakage inductances (Lpl1, Lpl2), magnetization inductances (Lpm1, Lpm2), and resonant capacitors (Cr1, Cr2), etc.) (within 5%) Even if the secondary circuit 530 can perform a current sharing operation without large current imbalance.

The rectifiers (diodes Dr1 to Dr4) of the secondary-side circuit 530 are respectively configured in parallel through secondary windings of the transformers T1 and T2. According to an aspect of the present embodiment, the secondary rectifier may be configured as a parallel connection using a center tap of each secondary winding of each transformer T1 and T2, as shown in FIG. 5. According to another aspect of this embodiment, the secondary-side rectifier may be configured as a parallel connection using a full-bridge of each secondary winding of each transformer T1, T2. However, it should be noted that this is an example, and the present embodiment is not limited thereto.

Including such a rectifying unit, the secondary-side circuit 530 may be implemented in various forms, such as the 3-bridge rectifying circuit shown in FIG. 7(a) or the parallel bridge rectifying circuit shown in FIG. 7(b). .

Referring to FIG. 6, the LLC resonant converter 500 operates in two modes (that is, the first mode and the second mode) according to the switching pattern of the primary circuit 510, and input voltage through switching frequency modulation control. Can be widely controlled in the '1/4Vin to Vin' range or the output voltage in the'Vo to 4Vo' range.

The input/output voltage control range for each mode of the LLC resonant converter 500 is summarized as follows.

1. Mode 1

During the primary half-bridge switching operation, the input voltage can be controlled in the '1/2Vin to Vin' range or the output voltage in the'Vo to 2Vo' range through switching frequency modulation control.

2. Mode 2

During the primary full-bridge switching operation, the input voltage can be controlled in the '1/4Vin to 1/2Vin' range or the output voltage in the '2Vo to 4Vo' range through switching frequency modulation control.

Hereinafter, a mode of the LLC resonant converter 500 according to an aspect of the present embodiment will be described in detail with reference to FIGS. 5 and 8A to 8E.

The first mode (Mode 1)-Example 1

8A is a diagram illustrating an operating waveform when the LLC resonant converter of FIG. 5 operates as an example of the first mode, and FIGS. 8B to 8E illustrate current paths when the LLC resonant converter of FIG. 5 operates as an example of the first mode. It is a figure to show.

Referring to Figure 8a, LLC resonant converter 500, in the first mode (Mode 1), switching elements Q2/Q6 and Q1/Q5 alternately at a duty ratio of 50% to perform a switching operation, , Since the switching element Q3 is turned on and Q4 is turned off (or vice versa), the primary switching elements Q1 to Q6 perform half-bridge switching operations.

During the half-bridge switching operation of the primary-side circuit 510, the current path for each time interval of the LLC resonant converter 500 is as illustrated in FIGS. 8B to 8E. Specifically, FIG. 8B is a't0 ~ t1 interval', FIG. 8C is a't1 ~ t2 interval', FIG. 8D is a't2 ~ t3 interval', and FIG. 8E is a't3 ~ t4 interval' of the LLC resonant converter 500. Indicates the current path.

The LLC resonant converter 500 controls the primary side switch elements Q1 to Q6 by a switching frequency modulation method under a half-bridge switching operation to adjust the gain, thereby adjusting the output voltage in the range of'Vo to 2Vo' Can be controlled widely.

The first mode (Mode 1)-Example 2

9A is a diagram showing an operating waveform when the LLC resonant converter of FIG. 5 operates as another example of the first mode, and FIGS. 9B to 9E show current paths when the LLC resonant converter of FIG. 5 operates as another example of the first mode. It is a figure to show.

Referring to FIG. 9A, in the first mode (Mode 1) of the LLC resonant converter 500, the switching elements Q3 and Q4 alternately perform a switching operation at a duty ratio of 50%, and the switching elements Q2 and Q6 are turn-on When Q1 and Q5 are turned off, the primary switching elements Q1 to Q6 perform half-bridge switching operations.

Alternatively, in the first mode (Mode 1) of the LLC resonant converter 500, the switching elements Q3 and Q4 alternately perform a switching operation at a duty ratio of 50%, and the switching elements Q2 and Q6 are turned off and Q1 and Q5 When is turned on, the primary-side switching elements Q1 to Q6 perform a half-bridge switching operation.

During the half-bridge switching operation of the primary-side circuit 510, the current path for each time period of the LLC resonant converter 500 is as illustrated in FIGS. 9B to 9E. Specifically, FIG. 9B is a section't0 to t1', FIG. 9C is a section't1 to t2', FIG. 9D is a section't2 to t3', and FIG. 9E is a section't3 to t4' of the LLC resonant converter 500 Indicates the current path.

The LLC resonant converter 500 controls the primary side switch elements Q1 to Q6 by a switching frequency modulation method under a half-bridge switching operation to adjust the gain, thereby adjusting the output voltage in the range of'Vo to 2Vo' Can be controlled widely.

2nd mode (Mode 2)

FIG. 10A is a diagram showing an operating waveform when the LLC resonant converter of FIG. 5 operates as an example of the second mode, and FIGS. 10B to 10E show current paths when the LLC resonant converter of FIG. 5 operates as an example of the second mode. It is a figure to show.

10A, in the second mode (Mode 2) of the LLC resonant converter 500, the switching elements Q2/Q3/Q6 and Q1/Q4/Q5 alternately switch at a duty ratio of 50%, so 1 The secondary switching elements Q1 to Q6 perform a full-bridge switching operation.

During the full-bridge switching operation of the primary circuit 510, the current path for each time period of the LLC resonant converter 500 is as shown in FIGS. 10B to 10E. Specifically, FIG. 10B is a't0 ~ t1 interval', FIG. 10C is a't1 ~ t2 interval', FIG. 10D is a't2 ~ t3 interval', and FIG. 10E is a't3 ~ t4 interval' of the LLC resonant converter 500. Indicates the current path.

The LLC resonant converter 500 controls the primary side switch elements Q1 to Q6 by a switching frequency modulation method under a full-bridge switching operation to adjust the gain, thereby adjusting the output voltage in the range of '2Vo to 4Vo' Can be controlled widely.

11 is a view showing an application example of the LLC resonant converter of FIG. 5.

Referring to FIG. 11, the LLC resonant converter application circuit 1100 applies a three-phase AC/DC converter (eg, a Vienna rectifier, a boost converter, a PWM rectifier, etc.) to the input terminal to control the V _LINK voltage constantly, and the LLC It is implemented by connecting two resonant converters in parallel. At this time, the LLC resonant converters can operate in parallel to control each input/output voltage over a wide range (input voltage: 1/4 Vin to Vin, output voltage: Vo to 4Vo).

12 is a graph showing the experimental results of comparing the operating performance of the LLC resonant converter of FIG. 5 and the LLC resonant converter according to the prior art.

12(a) shows the operating waveform of the LLC resonant converter to which the resonant capacitor is not commonly connected, and FIG. 12(b) shows the operating waveform of the LLC resonant converter to which the resonant capacitor is commonly connected according to the present embodiment. .

In the experiment of FIG. 12, (1) the first resonant circuit, the second resonant circuit, and the secondary-side rectifier were operated in parallel, and (2) the resonant capacitors Cr1 and Cr2 applied to the resonant circuit used the same value, ( 3) The input is DC 400V and the output is DC 36V / 700W, (4) It is performed under the assumption that the magnetization inductances (Lpm1, Lpm2) of the transformers (T1, T2) differ by more than the tolerance (5%) by 6%.

Referring to (a) of FIG. 12, it can be confirmed that, in the LLC resonant converter in which the resonant capacitors Cr1 and Cr2 are not commonly connected, the current imbalance between the currents I _Dr2 and I _Dr4 of the secondary circuit is large.

On the other hand, referring to (b) of FIG. 12, it can be seen that the LLC resonant converters to which the resonant capacitors Cr1 and Cr2 are commonly connected show little current imbalance between the currents I _Dr2 and I _Dr4 of the secondary circuit.

In summary, the LLC resonant converter 500 according to the present embodiment includes resonant capacitors Cr1 and Cr2 connected in common, and has two modes (half-bridge switching in the first mode and full-in the second mode). In the bridge switching operation), the switching element is controlled by the switching frequency modulation control method, thereby effectively controlling the input/output voltage.

13 is a view showing an LLC resonant converter according to another embodiment of the present invention, and FIG. 14 is a view showing the gain characteristics of the LLC resonant converter of FIG. 13.

Referring to FIG. 13, the LLC resonant converter 1300 is a primary circuit 1310 as two capacitors C1 and C2, six main switching elements Q1 to Q6, and two auxiliary switching elements SA1 and SA2. , The first resonant circuit unit 1311 and the second resonant circuit unit 1313 may be included. In addition, the LLC resonant converter 1300 may include a rectifier as the secondary-side circuit 1330.

Hereinafter, a description of contents overlapping with the LLC resonant converter 500 described above with reference to FIG. 5 will be omitted or simplified.

In the primary-side circuit 1310, the first bridge arm (Br1) in which capacitors C1 and C2 are connected in series, the second bridge arm (Br2) in which the main switching elements Q1 and Q2 are connected in series, and the main switching elements Q3 and Q4 are connected in series. The third bridge arm Br3 and the fourth bridge arm Br4 connected to the main switching elements Q5 and Q6 in series are connected in parallel between the + terminal and the-terminal of the input power supply Vin.

Further, the auxiliary switching elements SA1 and SA2 are connected in series between the connection terminals of the capacitors C1 and C2 and the node b between the source of the main switching element Q3 and the drain of Q4.

The rectifiers (diodes Dr1 to Dr4) of the secondary-side circuit 1330 are respectively configured in parallel through secondary windings of the transformers T1 and T2. According to an aspect of the present embodiment, the secondary rectifier may be configured as a parallel connection using a center tap of each secondary winding of each transformer T1 and T2, as shown in FIG. 5. According to another aspect of this embodiment, the secondary-side rectifier may be configured as a parallel connection using a full-bridge of each secondary winding of each transformer T1, T2. However, it should be noted that this is an example, and the present embodiment is not limited thereto.

Including such a rectifying unit, the secondary-side circuit 1330 may be implemented in various forms such as various rectifying circuits such as the three-bridge rectifying circuit or the parallel bridge rectifying circuit described above with reference to FIG. 7.

Referring to FIG. 14, the LLC resonant converter 1300 operates in two modes (that is, the first mode and the second mode) according to the switching pattern of the primary circuit 1310, and the input voltage is '1/3Vin'. ~Vin' range, or the output voltage can be widely controlled over the range of'Vo ~ 3Vo'. Hereinafter, each mode of the LLC resonant converter 1300 will be described in detail.

The first mode (Mode-1)

In the first mode (Mode-1) of the LLC resonant converter 1300, (1) the primary-side switching elements Q2/Q6 and Q1/Q5 are alternately switched at a constant switching frequency and a duty ratio of 50%. And (2) primary-side switching elements Q3 and Q4 and auxiliary switching elements SA1 and SA2 perform a switching operation using a pulse width modulation (PWM) control method.

The LLC resonant converter 1300, in the first mode (Mode-1), does not use a switching frequency modulation control method, and primary switching elements (Q3) at a fixed switching frequency near the resonant frequency (fr). And Q4, SA1 and SA2) by controlling the gain by controlling the pulse width modulation method, so that the magnetization inductances Lpm1 and Lpm2 can be applied without significantly reducing the increase in magnetization current and the resulting conduction loss. loss) can be prevented.

In addition, the LLC resonant converter 1300, in the first mode (Mode-1), pulse width modulation of the primary side switching elements (Q3 and Q4, SA1 and SA2) at a fixed switching frequency near the resonant frequency fr. By controlling the gain by controlling the method, the input voltage can be widely controlled in the '1/2Vin to Vin' range or the output voltage in the'Vo to 2Vo' range.

2nd mode (Mode-2)

In the second mode (Mode-2) of the LLC resonant converter 1300, (1) the primary-side auxiliary switching elements SA1 and SA2 are turned off, the switching elements Q2/Q3/Q6 and Q1/Q4/ Q5 alternates at a duty ratio of 50% to perform the switching operation, and (2) primary switching elements Q2/Q3/Q6 and Q1/Q4/Q5 perform switching operations using a switching frequency modulation control method.

The LLC resonant converter 1300 controls the primary side switch elements Q2/Q3/Q6 and Q1/Q4/Q5 in a second mode (Mode-2) by switching frequency modulation. By doing so, when the output voltage Vo is constant, the input voltage can be widely controlled in the range of '1/4Vin to 1/2Vin' or, when the input voltage Vin is constant, in the range of '2Vo to 4Vo'.

Hereinafter, with reference to FIGS. 15A to 16E, the mode of the LLC resonant cover 1300 according to another aspect of the present embodiment will be described in detail.

15A is a diagram showing an operating waveform when the LLC resonant converter of FIG. 13 operates as an example of the first mode, and FIGS. 15B to 15G show a current path when the LLC resonant converter of FIG. 13 operates as an example of the first mode. It is a figure to show.

15A, in the first mode (Mode-1) of the LLC resonant converter 1300, (1) the primary switching elements Q2/Q6 and Q1/Q5 are fixed switching frequencies near the resonance frequency fr and Switching operation is performed by alternating with a 50% duty ratio, and (2) primary switching elements Q3 and Q4 and auxiliary switching elements SA1 and SA2 are switched using a pulse width modulation (PWM) control method. Perform an action.

The current path for each time period of the LLC resonant converter 1300 is as illustrated in FIGS. 15B to 15G.

1.t0-t1 section

Referring to FIG. 15B, the primary-side switching elements Q1, Q4, and Q5 are turned off during this period, so that the parasitic capacitors Cp of these switching elements are charged and charged with the input voltage Vin.

Since the primary switching elements Q3, Q2, and Q6 are turned on, the input voltage Vin is applied in parallel to the first resonant circuit and the second resonant circuit, and resonant currents IP1 and IP2 flow respectively.

The first resonant circuit and the second resonant circuit are composed of transformers T1 and T2 for resonance, leakage inductances Lpl1 and Lpl2, magnetization inductances Lpm1 and Lpm2, and resonant capacitors Cr1 and Cr2.

The secondary side rectification part is applied with a voltage by the secondary turn-ratio and gain according to the polarity of each transformer (T1, T2) winding, and the resonant current rectified through each center tap rectifying diode (Dr1, Dr3) flows into the load. Power is delivered.

During this period, since the primary secondary switching element SA2 is turned on and SA1 is turned off, the voltage across the parasitic capacitor Cp of the secondary switching element SA1 is charged by 1/2 of the input voltage Vin. have.

2. t1-t2 section

Referring to FIG. 15C, when the primary switching element Q3 is turned off at a time t1, the parasitic capacitor Cp of Q3 starts charging, and the voltage of the parasitic capacitor Cp of Q4 charged with the input voltage Vin Starts discharging. In addition, the voltage (Vin/2) charged in the parasitic capacitor Cp of the primary-side auxiliary switching element SA1 is discharged to a '0' voltage through the primary-side switching elements Q2 and Q6 through the first and second resonant circuits. do. The voltage of the parasitic capacitor Cp of the switching element Q3 is charged by 1/2 of the input voltage Vin, and the voltage of the parasitic capacitor Cp of the switching element Q4 is discharged by 1/2 of the input voltage Vin, When the voltage of the parasitic capacitor Cp of the primary-side auxiliary switching element SA1 is discharged to a voltage of 0, this section operation ends.

When the voltage of the parasitic capacitor Cp of the primary-side auxiliary switching element SA1 is discharged to a voltage of '0' and then turned on, the zero-voltage switching (ZVS) turns-on, thereby reducing switching loss. .

3. t2-t3 section

Referring to FIG. 15D, since the voltage of the parasitic capacitor Cp of the primary-side auxiliary switching element SA1 is discharged to a voltage of 0 before time t2, the zero-voltage switching is turned on, so the secondary-side auxiliary switching element is in the t2-t3 section. SA1 and SA2 are both turned on so that half of the input voltage Vin is applied in parallel to the first resonant circuit and the second resonant circuit, respectively, and the resonant currents (IP1, IP2) through the primary switching elements Q2 and Q6 respectively. ) Will continue to flow.

Since the secondary side rectifying part is applied to the first resonant circuit and the second resonant circuit, 1/2 of the input voltage Vin is applied to the voltage of the primary and secondary turn-ratios and 1/2 gain of each transformer T1 and T2 It is applied, and the resonant current rectified through each center tap rectifying diode (Dr1, Dr3) flows to the load to transfer power.

4. t3-t4 section

Referring to FIG. 15E, as in the previous section t2-t3, the secondary auxiliary switching elements SA1 and SA2 are both turned on, so that half of the input voltage Vin is the first resonant circuit and the second resonant circuit. Is applied, and current flows, but this section (t3-t4) is a section where only the magnetizing current flows through the magnetizing inductances (Lpm1, Lpm2) of the primary transformers T1 and T2, and the resonance current does not flow to the load.

5. t4-t5 section

Referring to FIG. 15F, in the t4-t5 dead time period, the primary switching elements Q2 and Q6 and the auxiliary switching elements SA2 are turned off at time t4. Therefore, the voltage of the parasitic capacitor Cp of the primary-side switching element Q4, which was charged at half of the input voltage Vin, and the parasitic capacitor Cp of the primary-side switching elements Q1 and Q5, which were charged at the input voltage Vin. The voltage of) starts discharging, and the parasitic capacitor Cp of the primary-side auxiliary switching elements SA2 and the primary-side switching elements Q2 and Q6 starts charging. During the t4-t5 dead time period, the parasitic capacitors Cp of the primary switching element and the secondary switching element are charged/discharged by the magnetizing currents of the transformers T1 and T2, and the primary switching elements Q2, Q6 and Q3 are input. It is charged with the voltage Vin and the primary switching devices Q1, Q5 and Q4 are turned on when current flows through the anti-parallel diode after the primary switching devices Q1, Q5 and Q4 are discharged to the '0' voltage. At this time, the primary-side switching elements Q1, Q5 and Q4 become zero voltage switching (ZVS) to reduce switching loss. In addition, during the dead time period t4-t5, the parasitic capacitor Cp of the primary secondary switching element SA2 is charged to 1/2 of the input voltage Vin.

6. t5-t6 section

Referring to FIG. 15G, at t5, the primary switching elements Q1, Q5 and Q4 are turned on at zero voltage, and the input voltage Vin is applied in parallel to the first resonant circuit and the second resonant circuit, respectively. Resonant currents IP1 and IP2 flow through the circuit. At this time, voltages according to the primary and secondary turn-ratios of the transformers T1 and T2 and the gain of the resonance circuit are applied.

The secondary side rectifier is applied with voltage by the secondary turn-ratio and gain according to the polarity of each transformer (T1, T2) winding, and is rectified through each center tap secondary side rectifying diode (Dr2, Dr4) to resonate the load in parallel Electric current flows and power is transferred.

Subsequently, in the remaining half period, the LLC resonant converter 1300 repeats the operation of the previous half period (ie, t0-t6 period).

As described above, the LLC resonant converter 1300, in the first mode, according to the pulse width modulation (PWM) control of the primary primary switching elements (Q3, Q4) and secondary switching elements (SA1, SA2), The voltage applied to each of the 1 resonant circuit and the 2nd resonant circuit is applied as 1/2 of the input voltage (Vin) and the input voltage (Vin), and the output voltage (Vo) can be controlled in the range of'Vo to 2Vo'. have.

FIG. 16A is a diagram showing an operating waveform when the LLC resonant converter of FIG. 13 operates as an example of the second mode, and FIGS. 16B to 16E show current paths when the LLC resonant converter of FIG. 13 operates as an example of the second mode. It is a figure to show.

Referring to FIG. 16A, in the second mode (Mode-2) of the LLC resonant converter 1300, (1) the primary side auxiliary switching elements SA1 and SA2 are turned off, and the switching elements Q2/Q3/ Q6 and Q1/Q4/Q5 alternately perform a switching operation at a duty ratio of 50%, and (2) primary-side switching elements Q2/Q3/Q6 and Q1/Q4/Q5 control switching frequency modulation (Switching-Frequency Modulation) In a full-bridge switching operation.

Under the full-bridge switching operation, the input voltage Vin is applied in parallel to the first resonant circuit and the second resonant circuit of the LLC resonant converter 1300, so that the resonant current flows, and the secondary-side rectifier includes transformers T1 and T2. The voltage by the secondary turn-ratio and gain is applied according to the polarity of each winding, and is rectified through the secondary-side rectifying diodes Dr1 and Dr3 of each center tap so that load resonance current flows in parallel to transmit power.

Under the full-bridge switching operation, the current path for each time period of the LLC resonant converter 1300 is as shown in FIGS. 16B to 16E. Specifically, FIG. 16B is a't0 to t1 section', FIG. 16C is a't1 to t2 section', FIG. 16D is a't2 to t3 section', and FIG. 16E is an LLC resonant converter 1300 in the't3 to t4 section'. Indicates the current path.

As described above, the LLC resonant converter 1300 controls the gain by controlling the primary side switch elements Q1 to Q6 by a switching frequency modulation method under a full-bridge switching operation, thereby adjusting the output voltage from '2Vo ~ It can be widely controlled in the 4Vo' range. This is because the output voltage of the second mode (Mode-2) at the resonant frequency (fr) is doubled than that of the first mode (Mode-1), so that the magnetization current induced to the primary side increases to improve the gain. .

FIG. 17 is a diagram showing an application example of the LLC resonant converter of FIG. 13.

Referring to FIG. 17, the LLC resonant converter application circuit 1700 applies a three-phase AC/DC converter (eg, a Vienna rectifier, a boost converter, a PWM rectifier, etc.) to the input terminal to control the V _LINK voltage constantly, and the LLC It is implemented by connecting two resonant converters in parallel. At this time, each LLC resonant converter individually modulates the pulse width or the switching frequency of the primary switching elements Q3 and Q4 and the secondary switching elements SA1 and SA2 individually in the first mode (Mode-1) and the second mode (Mode-2). By controlling by the modulation method, each input/output voltage range can be widely controlled (input voltage: 1/4 Vin to Vin, output voltage: Vo to 4Vo).

18 is a graph showing experimental results comparing the performance of the LLC resonant converter of FIG. 13 and the LLC resonant converter according to the prior art.

The experiment in FIG. 18 is an experimental waveform when the output voltage is controlled to DC 48V in an environment in which the auxiliary switching elements SA1 and SA2 and the switching elements Q3 and Q4 are switched by a pulse width modulation method, and the output capacity is 2 kW. Shows.

18(a) shows the operating waveforms of the LLC resonant converter operating in the first mode (Mode-1) in the case of the primary and secondary terminal voltages (Vab, VS11) and currents (IP1, IS11). (b) shows the operating waveform of the LLC resonant converter operating in the first mode (Mode-1) in the case of the primary terminal voltage (Vab) and secondary current (IS11, IS21)/voltage (VS11).

In the first mode (Mode-1), the half-bridge and full-bridge modes coexist in accordance with pulse width modulation, and the auxiliary switching elements SA1 and SA2 and the switching elements Q3 and Q4 Mode switching is possible through the switching operation of the pulse width modulation control method.

In accordance with such operation control, the LLC resonant converter 1300 has a stable output voltage control range of'Vo to 2Vo' that is current-sharing without generating overshoot and undershoot.

The above description is merely illustrative of the technical idea of the present embodiment, and those skilled in the art to which this embodiment belongs may be capable of various modifications and variations without departing from the essential characteristics of the present embodiment. Therefore, the present embodiments are not intended to limit the technical spirit of the present embodiment, but to explain, and the scope of the technical spirit of the present embodiment is not limited by these embodiments. The protection scope of the present embodiment should be interpreted by the claims below, and all technical spirits within the equivalent range should be interpreted as being included in the scope of the present embodiment.

Claims (14)

In the LLC resonant converter comprising a primary side circuit for converting the DC input voltage Vin to an AC voltage of a predetermined level and a secondary side circuit for rectifying the converted AC voltage to supply the DC output voltage Vo,
The primary circuit,
A first resonant circuit unit including a transformer T1, a resonant inductor Lpl1, and a resonant capacitor Cr1;
A second resonant circuit unit including a transformer T2, a resonant inductor Lpl2, and a resonant capacitor Cr2;
And a plurality of switching elements that convert the DC input voltage Vin to an AC voltage through a switching operation and apply it to the first resonant circuit unit and the second resonant circuit unit,
The secondary circuit,
It includes a rectifying unit including a plurality of diodes (Dr1, Dr2, Dr3 and Dr4) rectifying the AC voltage converted by the primary-side circuit by being connected to the secondary coil of the transformer T1 and the transformer T2,
The resonant capacitor Cr1 and the resonant capacitor Cr2 are respectively connected to one end and the other end in common to prevent current imbalance of the secondary circuit due to tolerance of the resonant element,
The switching element,
The first switching element Q1 and the second switching element Q2 connected in series to the first bridge arm, the third switching element Q3 and the fourth switching element Q4 and the third bridge arm connected in series to the second bridge arm are connected in series. A fifth switching element Q5 and a sixth switching element Q6,
The first to third bridge arms are connected in parallel to the input terminal,
The first resonant circuit unit,
It is connected between a node between the source of the switching element Q1 and the drain of the switching element Q2, and a node between the source of the switching element Q3 and the drain of the switching element Q4,
The second resonant circuit unit,
It is connected between a node between the source of the switching element Q3 and the drain of the switching element Q4, and a node between the source of the switching element Q5 and the drain of Q6,
The primary circuit,
By controlling the switching element by a switching frequency modulation method to perform any one of the half-bridge (half-bridge) operation and full-bridge (full-bridge) operation, at least one of the DC input voltage and DC output voltage Control to a certain level,
The polarity directions of the primary coils of the transformer T1 and the transformer T2 are both directed to a node between the source of the switching element Q3 and the drain of the switching element Q4, and the first resonant circuit part and the second resonant circuit are driven by the polarity direction. Each furnace part operates in parallel, and the rectifier operates in parallel with respect to the secondary winding of each of the transformer T1 and the transformer T2,
The half-bridge (half-bridge) operation,
When the switching element Q3 and the switching element Q4 are turn-on and turn-off or turn-off and turn-on, respectively, the switching element Q2 and the switching element Q6 And, the switching element Q1 and the switching element Q5 alternately switch at a duty ratio of 50%,
The switching element Q2 and the switching element Q6 are turned on and the switching element Q1 and the switching element Q5 are turned off, or the switching element Q2 and the switching element Q6 are turned on. -When turned off and the switching element Q1 and the switching element Q5 are turned on, the switching element Q3 and the switching element Q4 alternate with a duty ratio of 50%. Is switching operation,
The full-bridge (full-bridge) operation,
The switching element Q2, the switching element Q3 and the switching element Q6, and the switching element Q1, the switching element Q4 and the switching element Q5 alternately switch at a duty ratio of 50%.
LLC resonant converter. According to claim 1,
The resonant inductor Lpl1,
Leakage inductor of the transformer T1
LLC resonant converter. According to claim 1,
The resonant inductor Lpl2,
Leakage inductor of the transformer T2
LLC resonant converter. According to claim 1,
The rectifying unit,
The transformer T1 and the transformer T2 are implemented as parallel center tap rectifier circuits for the secondary winding of each.
LLC resonant converter. According to claim 1,
The rectifying unit,
The transformer T1 and the transformer T2 are implemented as parallel full-bridge rectifier circuits for the secondary winding of each.
LLC resonant converter. delete delete delete delete delete delete delete delete delete

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