KR100535273B1 - Method and apparatus for wideband predistortion linearization - Google PatentsMethod and apparatus for wideband predistortion linearization Download PDF
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- KR100535273B1 KR100535273B1 KR10-2000-7006897A KR20007006897A KR100535273B1 KR 100535273 B1 KR100535273 B1 KR 100535273B1 KR 20007006897 A KR20007006897 A KR 20007006897A KR 100535273 B1 KR100535273 B1 KR 100535273B1
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- 230000003321 amplification Effects 0.000 claims description 4
- 238000003199 nucleic acid amplification method Methods 0.000 claims description 4
- 238000000034 methods Methods 0.000 description 31
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- 238000004458 analytical methods Methods 0.000 description 1
- 230000000694 effects Effects 0.000 description 1
- H03—BASIC ELECTRONIC CIRCUITRY
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/32—Modifications of amplifiers to reduce non-linear distortion
- H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
- H03F1/3247—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits using feedback acting on predistortion circuits
- H03—BASIC ELECTRONIC CIRCUITRY
- H03F1/00—Details of amplifiers with only discharge tubes, only semiconductor devices or only unspecified devices as amplifying elements
- H03F1/32—Modifications of amplifiers to reduce non-linear distortion
- H03F1/3241—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits
- H03F1/3258—Modifications of amplifiers to reduce non-linear distortion using predistortion circuits based on polynomial terms
- H03—BASIC ELECTRONIC CIRCUITRY
- H03F2200/00—Indexing scheme relating to amplifiers
- H03F2200/102—A non-specified detector of a signal envelope being used in an amplifying circuit
The present invention relates to a method and apparatus for compensating for distortion. More particularly, the present invention relates to a wideband predistortion linearization method and apparatus.
Radio frequency (RF) signals often include envelope deviation due to, for example, amplitude modulation or a combination of two or more RF carriers or RF tones. Inter Modulation Distortion (IMD) appears when an amplitude modulation or multi-tone RF signal is amplified by a nonlinear amplifier, for example. Due to the IMD, unwanted interference is generated at frequencies other than the tone frequency of the multi-tone RF signal. Typically, this interference occurs near the tone frequency, making it difficult to filter out. Therefore, certain forms of linearization are desirable to suppress IMD caused by nonlinear amplification.
In amplifier design, there is tradeoff between distortion performance and efficiency. Linear amplifiers operating under "Class A" conditions rarely produce distortion, but are inefficient. On the other hand, nonlinear amplifiers operating under "class C" conditions are somewhat efficient, but introduce significant distortion. In amplifier design, both efficiency and distortion are important considerations, but at higher power levels efficiency becomes more important. Because of such efficiency, there are distortion problems that must be solved, but nonlinear amplifiers account for the majority.
There are many known amplifier linearization techniques that reduce the distortion caused by nonlinear amplification. Conventional amplifier linearization techniques can be broadly classified into feedback, feedforward, and predistortion.
Feedback is a well-known linearization technique that is widely applied. For example, US Pat. No. 2,102,671 by Black discloses an early negative feedback linearization technique for reducing distortion at audio frequencies. H.A. Rosen and A.T. Owens, "Power Amplifier Linearity Studies for SSB Transmissions", IEEE Transactions on Communication Systems, pp. 150-159, June 1964 discloses a feedback linearization technique for reducing distortion at radio frequencies. Recent feedback techniques are karting feedback using negative feedback of the base quadrature modulation. An example of this kind of feedback is M.A. Briffa and M. Faulner, "Stability Analysis of Cartesian Feedback Linearization for Amplifiers with Weak Nonlinearities", IEEE Proceedings on Communications, Vol. 143, No. 4, pp. 212-218, Aug. Disclosed in 1996. Other modern feedback linearization techniques include Polar Feedback, disclosed in US Pat. No. 5,023,937 to Opas, and K.G. Voyce and J.H. Intermediate frequency (IF) feedback as disclosed in McCandles, "Power Amplifier Linearization Using IF Feedback", IEEE MTT-S Digest, pp. 863-866, 1989.
The problem with these feedback linearization techniques is that system delay typically limits the available linearization bandwidth. Therefore, feedback techniques are generally limited to narrowband systems, for example single carrier linear modulation schemes. Another disadvantage of these feedback linearization techniques is that they are potentially unstable.
Feed forward is another well-known linearization technique that has been successfully applied to radio frequencies. Typical feedforward RF power amplifiers are disclosed in US Pat. No. 5,157,346 to Powell et al. According to this technique, in the first comparison loop, the amplified output and the input signal are compared to output an error signal. In the second correction loop, the original distortion at the output and the error signal deviating 180 degrees out of phase are amplified and reentered at the output, thereby removing distortion at the final output. This technique provides excellent IMD suppression capability over a wide linearization bandwidth.
The disadvantage of the feedforward linearization technique is that it typically operates in class A, since it usually requires the use of an error amplifier required to be linear. However, although feedforward amplifiers are much more efficient with significant performance compared to class A amplifiers, they reduce the efficiency of the feedforward technique for the same reason.
Another well-known linearization technique is predistortion. According to this technique, linearization is achieved by distorting the input signal in a manner complementary to the amplifier distortion in accordance with the predistortion function such that the entire transmission from the predistorter input to the amplifier output is a linear function.
Predistortion may be applied at the actual RF frequency or baseband, ie prior to modulation with the RF carrier. When applied at RF frequencies, the predistortion technique exhibits wideband linearization performance. However, because the predistortion function is more complex and it is difficult to implement distortion at higher orders, the predistortion technique typically reduces the IMD product to the third order. T.Nojima and T.Konno, "Cuber Predistortion Linearizer for Relay Equipment in 800 MHz Band Land Mobile Telephone Systems", IEEE Transactions on Vehicular Technology, Vol.VT-34, No.4, pp.169-177, Nov.1985 And US Pat. No. 4,943,783 by Nojima discloses a typical third-order predistorter that reduces the IMD product in third order.
1A shows a conventional third order predistorter 1 disclosed in Nojima and Konno. The input RF signal received via the input terminal 2 is divided by the power divider 4 into two signals having substantially the same amplitude. One of the divided signals is provided in a linear signal path comprising a variable delay line 9. The other divided signal is provided to a nonlinear signal path comprising a cubic function generator 6, a variable phase adjuster 7, and a variable attenuator 8. The tertiary function generator 6 generates a third order predistortion signal based on the received input RF signal and outputs a predistortion signal at terminal 12. The variable phase adjuster 7 adjusts the phase of the predistortion signal, and the variable attenuator 8 adjusts the amplitude of the predistortion signal. The amplitude and phase adjusted predistortion signal is combined in the combiner 5 with the linear signal supplied in the delay line 9. The combined signal is sent from terminal 10 to an RF power amplifier (PA) 13. In this way, the third order IMD product caused by the RF PA 13 is removed from the amplified signal to linearize the RF PA 13. If the delay line 9 compensates for the delay that occurs when generating the predistortion signal, wideband linearization can be achieved.
FIG. 1B shows the spectrum of an RF signal comprising two tones at f 1 and f 2 that can be applied to the predistortion circuit 1. 1C shows the output spectrum of the RF PA 13. As it is shown in Figure 1C, the output spectrum f 1 and 2f 1 -f produced by the RF PA (13) as indicated by the primary component and a continuous line in the f 2 2 and Third order IMD distortion component at 2f 2 -f 1 . In addition, the output spectrum is frequency 2f 1 -f 2 and injected by the predistortion circuit 1 as indicated by the dotted line and Third order predistortion component at 2f 2 -f 1 . As can be seen from FIG. 1C, the injected third order predistortion component has the same amplitude and opposite phase as the third order IMD product of the RF PA 13. Therefore, the third order predistortion component cancels the third order IMD product.
The problem with this solution is that the RF power amplifier produces not only the 3rd order IMD product, but also the higher order IMD product. Typically, these higher order IMD distortion products are not taken into account, but actually occur when the third order IMD product is suppressed.
In an attempt to solve this problem, S.P. Stapleton and J.K. Cavers, "A New Technique for the Adaption of Linearizing Predistorters", Proceedings of the IEEE Vehicular Technology Conference, pp. 753-758, May 1991, describe a predistorter for compensating for more than three orders of magnitude of IMD products. 2 illustrates such an improved predistorter. Such a predistorter essentially comprises three main blocks: orthogonal gain phase adjuster (QGPA) 14, baseband polynomial predistortion circuit (PreD) 15, and controller 16.
As shown in FIG. 2, an input RF signal at the input terminal 17 is supplied to both the QGPA 14 and the PreD 15. The PreD 15 circuit detects the envelope of the input RF signal at the detector 22, and processes the detection envelope through the nonlinear function generators F 1 (x) 23 and F 2 (x) 24, Generate two predistortion signals. Function generators F 1 (x) 23 and F 2 (x) 24 are in-phase and quadrature (I & Q) signals that are multiplied by the input signal at QGPA 14 to form third and fourth order predistortion components. Create The complex multiplication process combined with the use of the detection envelope allows both amplitude modulation / amplitude modulation (AM / AM) and amplitude modulation / phase modulation (AM / PM) distortion to be corrected, but results in a karting form.
The complex multiplication is achieved by splitting the input RF signal into two paths in the QGPA 14 circuit, one path being input to multiplier 19 and the other being input to multiplier 21. The signals on both paths are the same except that the signal input to the multiplier 21 is 90 ° phase shifted by the phase shifter 20 with respect to the signal input to the multiplier 19.
The coefficients of the two polynomial functions generated in the function generators F 1 (x) 23 and F 2 (x) 24 are provided by a microprocessor (p) 25 in the controller 16. The microprocessor 25 adjusts the coefficients based on the magnitude of the in-phase and quadrature feedback signals from the output of the linearized RF PA (not shown). The I & Q feedback signal is bandpass filtered at filters 28 and 29 to measure the IMD by separating the actual desired signal from the IMD. This is only possible if the signal to be amplified is a single carrier signal in which the IMD product is known to be in either band of single carrier modulation. Detectors 26 and 27 determine the size of the band-filtered IMD so that microprocessor 25 can adjust the coefficients of the distortion function to minimize this distortion, thereby minimizing the level of IMD products present at the output of the RF PA. do.
Although this technique compensates for more than 3rd order IMD products, it is only suitable for single carrier applications. In a multi-carrier system, the location of the carrier, and therefore the resulting IMD, may not always be obtained with a bandpass filtering approach. This fact makes this technique unsuitable for wideband applications that typically include multi-tone input signals. The broadband characteristics of these applications result from the combination of individual narrowband RF signals spaced between some given band plans (spectrums).
Digital signal processing (DSP) can be applied for more precise predistortion. For example, US Pat. No. 5,049,832 to Caver discloses an adaptive linearization technique using DSP. The problem with using a DSP is that the linearization bandwidth is greatly limited by the DSP sampling frequency and the required digital-to-analog converter. Therefore, systems using DSPs are usually not suitable for wideband applications.
None of these conventional techniques compensate for higher order IMD in wideband applications with multi-tone RF input signals. In addition, none of these prior arts distinguish between peak and average level signals.
When tones are combined in a multi-tone system, structural interference causes the peaks of the multi-tone RF signal to be very large relative to the average level. For an economical and relatively power efficient nonlinear multi-tone power amplifier, it cannot be dimensioned to adjust the peak of such a signal. As a result, some peaks in the multi-tone signal will be clipped if the amplifier saturation limit is exceeded.
In a conventional predistorter as shown in Figures 1A and 2, no attempt is made to distinguish operation below or above the saturation limit of the RF PA. As a result, predistortion is usually applied to the RF input signal causing the RF PA to saturate. Once the RF PA is saturated, injecting a predistortion signal has no effect on the amplitude of the output. This is because large complementary inputs to the RF PA do not raise the output amplitude above the saturated RF output limit. However, the situation with respect to phase is greatly different. Predistortion correction applied to the phase of the RF input signal is sent to the output. In conventional predistorters, these phase corrections are very inaccurate with respect to the peak input signal, resulting in a large degradation in the predistortion performance at that peak.
Therefore, there is a need for a wideband predistortion technique for multi-tone RF signals that compensates for higher order IMD products. There is also a need for predistortion techniques that are valid for peak RF input signals.
Embodiments of the present invention will be described in more detail with reference to the accompanying drawings, in which like components are designated by like reference numerals.
Is a block diagram of a conventional third order predistorter.
FIG. 1B shows the spectrum of a two tone input RF signal applied to the predistorter shown in FIG. 1A. FIG.
1C shows the spectrum of the RF PA amplified output of the third order predistorter shown in FIG. 1A.
2 is a block diagram of a conventional fifth order predistorter.
3 is a block diagram of an analog predistorter system in accordance with an embodiment of the present invention.
4 is a detailed block diagram of an example predistortion circuit in accordance with an embodiment of the present invention.
5A-5C illustrate aspects of terms of a predistorted RF signal in the frequency domain in accordance with an embodiment of the invention.
6A-6C illustrate aspects of terms of a predistorted RF signal in the time domain in accordance with an embodiment of the invention.
7A illustrates a method of compensating for predistortion in accordance with an embodiment of the present invention.
7B illustrates a method for generating a predistortion signal in accordance with an embodiment of the present invention.
It is therefore an object of the present invention to provide a technique for compensating higher order IMD products. It is yet another object of the present invention to provide very large advances in IMD compensation for average input signals while at the same time preventing degradation of IMD compensation for peak input signals.
According to an embodiment of the invention, a predistortion signal is generated that compensates for more than three orders of magnitude of the IMD product over a wideband, and the predistortion signal is provided to an input RF signal, for example a multi-tone RF signal. The predistortion function is a lower order polynomial with adjustable coefficients. Since the predistortion function is generated based on the detection envelope of the input RF signal, the predistortion is practically independent of the tone frequency. The detection envelope is clipped and scaled in an approximate hyperbolic tan shape. The clipping form is approximated in the form of a hyperbolic tanh function to prevent the detection envelope from exceeding a certain value, thus greatly preventing inaccurate predistortion compensation in the presence of large peaks in the detection envelope.
According to an embodiment, the predistortion signal compensates for the IMD product generated by the nonlinear amplifier. The coefficient of the polynomial is adjusted based on the difference between the amplifier output and the input RF signal.
In this specification, specific details such as specific circuits, circuit elements, techniques, and the like are provided for the purpose of description and not of limitation, to provide a thorough understanding of the invention. It will be apparent to one skilled in the art that the present invention may be practiced in the form of other embodiments that depart from these specific details. In the following description, well-known methods, devices, and circuits are omitted for clarity.
According to the present invention, the 3rd or higher IMD product is compensated by applying an input RF signal to the predistortion signal. According to an embodiment of the invention, the predistortion signal is generated by an analog processing component that allows wideband operation.
3 illustrates a predistortion system 100 according to an embodiment of the invention. The system includes a predistortion circuit (PreD) 37 for generating an analog predistortion signal and a quadrature gain phase adjuster (QGPA) 36 for applying the predistortion signal to the multi-tone RF input of the RF PA 13.
PreD 37 internally generates an appropriate predistortion signal based on the multi-tone RF input signal RF in received at terminal 30. The received input signal q from terminal 30 is divided into two branches of equal amplitude by the input coupler 33. (The bold lines represent the complex signal, and the time dependency is clear. One branch is supplied to the delay member 42. The other branch is fed to splitter 34 as signal x . Splitter 34 splits signal x into two branches of equal amplitude, one to PreD 37 and the other to 90 ° splitter 35. The 90 ° phase splitter splits the signal x into two branches, multiplying 0 ° in one branch by 90 ° in the other, resulting in a complex signal.
PreD 37 generates a predistortion signal p based on signal x . 90 ° splitter 35 adjusts the amplitude and phase of signal x by allowing QGPA 36 to multiply signal x by complex predistortion signal p from PreD 37. The adjusted signal is output from the QGPA 36 to the RF PA 13 as a signal r .
The operation of the predistortion system of the gain-based form may be mathematically expressed as Equation 1.
Where r is a predistorted RF signal, x is an RF input, and p is a predistortion signal (or dynamic complex gain signal) generated by PreD 37, i.e. p = p i + jp q . The predistortion signal p may be generated by the PreD 37 in a karting form as follows.
Where | x | is the magnitude of the signal x and C 2i , C 2q , C 1i , C 1q , C 0i and C 0q represent the coefficients of the predistortion signal that can be adjusted by the controller 40. The QGPA 36 may be implemented with, for example, two multipliers and one adder to multiply terms of the predistortion signal p by the input signal x .
Equation 1 may be rewritten in an expanded complex form as in the following equation.
Referring to FIG. 3, the controller 40 adjusts the predistortion signal coefficients by minimizing the difference between the input q and the scaled output of the RF PA 13. The output v of the RF PA 13 is coupled to the attenuator 43 via the coupler 47. The coupled output is scaled in an attenuator 43 and the scaled output is coupled to the input signal via the coupler 46 such that the input signal is delayed by the member 42 by an amount substantially equal to the delay of the predistortion branch. do. The attenuator 43 scales the output ν from the coupler 47 to the size of 1 / G D (G D corresponds to the gain of the RF PA 13, the gain of the coupler 47, etc.), thereby reducing the scale of the output signal. The gain is matched with the gain of the delayed gain signal. The scaled output signal and the delayed input signal are coupled by a coupler 45 to a detector 41 which detects the difference between both signals and reports this difference to the controller 40 as an error signal e. The controller 40 adjusts the predistortion coefficients C 1i , C 1q , C 2i , C 2q to minimize the error signal e. This explains the change in amplifier characteristics that occurs over time and over time. The controller 40 also corrects the static (static) portion of the difference between the RF amplifier input and output and generates a constant complex coefficients C 0i, C 0q which is added to the PreD output by the adder (38, 39). The controller 40 may be implemented by, for example, a microprocessor.
As can be seen from the gain based predistortion signal of Equation 3, without PreD (37), i.e., C 1i = 0, C 1q = 0, C 2i = 0, C 2q = 0 => C 1 = 0 In the case of C 2 = 0, the gain of QGPA 36 depends on the setting of the complex coefficient C 0 = C 0i + C 0q . Therefore, without PreD 37, QGPA 36 may adjust the complex gain of RF PA 13 regardless of the RF input signal supplied. Since the RF PA 13 has a complex gain that changes slightly as the RF input level changes, i.e., it is nonlinear, such fixed adjustments cause the input / output difference to zero at only one RF level. At other levels, the input / output difference is not zero. By injecting the PreD 37, it is possible to dynamically vary the complex gain as a function of the RF input level, and reduce the input / output difference over a range of RF levels, thereby efficiently linearizing the RF PA 13 do. C 1 Allowing the term allows PreD 37 to adjust the complex gain of QGPA 36 in proportion to the magnitude of the input. C 2 Allowing the term allows the complex gain to be changed in response to the square of the input magnitude.
4 shows a detailed block diagram of an example predistortion circuit. According to an embodiment, the predistortion circuit of FIG. 4 corresponds to PreD 37 of FIG. 3. However, the present invention is not limited thereto, and it is obvious that the predistortion circuit according to the present invention can be implemented in any system in which 3rd or more IMD compensation is desired.
Referring to FIG. 4, an input RF signal RFi (in accordance with an embodiment, corresponding to signal x of FIG. 3) is applied to an input detector comprising a mixer 48 and a limiting amplifier 49. The input detector detects an envelope of the input RF signal. This allows predistortion to be made substantially independent of tone frequency. The detected envelope is applied to a variable saturation amplifier (VSA) 50. The VSA 50 effectively makes it possible to clip the detected envelope to a clip level that can be controlled externally via a control voltage applied at terminal C3. According to an embodiment, the clipping form is approximated in the form of a hyperbolic tanh function that prevents the envelope from exceeding a certain value. For this reason, when a large peak exists in the detected envelope, it is possible to prevent the PreD 37 from significantly inaccurate pre-correction.
The signal clipped from the VSA 50 is applied to a variable gain amplifier (VGA) 51. The VGA 51 scales the clipped signal and enables the entire PreD 37 circuit to be switched off or gradually disabled in response to the control voltage applied at terminal C4.
The scaled output of the VGA 51 represents a preprocessed envelope of the RF input. This signal, | x | is squared in the squarer 52 to yield | x | 2 . To generate the function given by equations (2a) and (2b), | x | and | x | 2 are both applied to four linear output multiplier circuits 53, 54, 55, 56. These multipliers multiply C 1i , C 2i , C 1q , C 2q by | x | and | x | 2 , respectively. These multiplied signals are added by adders 57 and 59 and buffered in buffers 58 and 60 to yield two outputs PRI and PRQ, respectively. These outputs Equation 2a, 2b of the high-order term, i.e. PRI = | represents the C 1q | x | 2 C 2i + | x | C 1i, PRQ = | x | 2 C 2q + | x. The low order terms C 0i and C 0q of Equations 2a and 2b are sequentially added to PRI and PRQ by the adders 38 and 39 to output p i and p q , respectively.
Although described above in gain-based form, the predistortion system can be expressed by the input-to-output transfer function by expanding Equation 3 as follows.
Equation 5 shows that the predistorted signal r compensates for multiple IMD products above the first term x C 0 to compensate the first order IMD, the third term x | x | 2 C 2 , and the third order IMD product to compensate for the third order IMD. Since the additional term x | x | 2 C 1 is included, efficient linearization performance can be sufficiently achieved for higher order IMD products. This will be better understood with reference to FIGS. 5A-5C, which show aspects of terms of the predistorted signal r in the frequency domain, except for the coefficients. As can be seen from Figures 5A and 5C, the first term x provides the first order predistortion, and the term x | x | 2 provides substantially the third order predistortion. However, FIG. 5B shows that x | x | provides predistortion of at least three orders of magnitude.
In addition to providing predistortion for higher order IMD products, the x | x | term corresponds well to the dynamic range. As can be seen from FIGS. 6A-6C, which show the shape of the predistorted signal in the time domain, the amplitudes of these terms change rapidly, ie rise and fall faster every order of | x |. Similarly, if x | x | is compared with other higher-order terms, such as x 5 , x 7, etc., if the amplitude of x is relatively large (> 1), then the amplitude of the other higher-order term is greater than the amplitude of the x | x | term. As it rises rapidly, it will reach the clip limit earlier than the electronic embodiment. If x is relatively small (<1), the magnitude of the alternative higher order term drops faster than the size of x | x | and approaches the noise floor much faster in the electronic implementation. Therefore, when dealing with a multi-tone signal having a high dynamic range from the beginning, the x | x | term simplifies the electronic implementation of the predistortion circuit.
7A illustrates a method for compensating for distortion in accordance with an embodiment of the invention. The method begins at step 700 where an input multi-tone RF signal is received. In step 720, a predistorted signal is generated. In step 740, the predistortion signal is applied to the input multi-tone RF signal. After the predistortion signal is applied, the input multi-tone RF signal is amplified, for example, and the IMD generated by the amplifier is compensated by the predistortion signal. As shown in Fig. 7A, the predistortion method is repeated as long as the input RF signal is received.
7B illustrates a method for generating a predistortion signal in accordance with an embodiment of the invention. The method begins at step 722 where an envelope of the input multi-tone RF signal is detected. The detected envelope is clipped at step 724 and scaled at step 726. Next, in step 728, the controller 40 adjusts the coefficients based on, for example, the detected error between the input and the output of the amplifier. Finally, in step 730, the polynomial predistortion signal is calculated by multiplying the detected envelope by the coefficient adjusted by the controller 40.
According to the present invention, the average IMD performance can be improved over a wide bandwidth while reducing inaccurate predistortion at the peak input level. In experimental tests with an intermediate frequency of 1500 MHz and an average output power of 25 Watts (W), improvements over 10 decibels (dB) average IMD were realized without degrading peak IMD performance over at least a 10 MHz bandwidth. The predistortion technique according to the invention can be applied at almost all carrier frequencies with little modification.
It is apparent that the present invention is not limited to the specific embodiments described and described above. For example, while the above embodiment has been described with reference to compensating for distortion caused by a nonlinear amplifier, the present invention is applicable to compensating IMD from any source. In addition, although the input RF signal is described as a multi-tone RF signal, the present invention is also applicable to a single-tone RF input signal. The specification includes all modifications that fall within the scope of the invention as defined by the following claims.
- An apparatus for compensating for distortion over a broadband, the apparatus comprising:Means for receiving an input wideband multi-tone radio frequency signal;A predistortion circuit being low order polynomial and comprising means for generating a wideband predistortion signal that compensates for intermodulation distortion products of at least three orders of magnitude; AndA quadrature gain phase adjuster for applying said wideband predistortion signal to said input wideband multi-tone radio frequency signal;Including,And the lower order polynomial predistortion signal is a third order polynomial generated according to the following equation.Equation(Where r is a predistorted radio frequency signal, x represents an input radio frequency signal, and C 2 , C 1 , and C 0 represent adjustable complex control coefficients)
- The distortion compensating apparatus according to claim 1, further comprising a controller for adjusting the predistortion signal generated by the predistortion circuit.
- The distortion compensating apparatus of claim 1, wherein the intermodulation distortion product is generated by a nonlinear amplifier, and the predistortion signal is applied to the input radio frequency signal before amplification of the nonlinear amplifier.
- 6. The distortion compensator of claim 5, wherein the controller adjusts the predistortion signal based on a difference between the output of the amplifier and the input radio frequency signal.
- 4. The predistortion circuit of claim 1, wherein the predistortion circuit comprises an envelope detector for detecting an envelope of the input radio frequency signal, wherein the predistortion circuit generates the predistortion signal based on the detected envelope. Distortion compensation device.
- 8. The distortion compensator of claim 7, wherein the predistortion circuit comprises a variable saturation amplifier for clipping the detected envelope in an approximate hyperbolic tan shape.
- 9. The distortion compensator of claim 8, wherein the predistortion circuit comprises a variable gain amplifier for scaling the clipped envelope.
- In a method for compensating for distortion over a broadband,Receiving an input wideband multi-tone radio frequency signal;Generating a wideband predistortion signal that is a lower polynomial; AndApplying the wideband, lower order polynomial predistortion signal to the input wideband multi-tone radio frequency signal in quadrature using a quadrature gain phase adjuster, wherein the wideband, lower order predistortion signal is third order or more intermodulated Contains terms that compensate for distortion products-Including,The low-order polynomial predistortion signal is generated according to the following equation.Equation(Where r is a predistorted radio frequency signal, x represents an input radio frequency signal, and C 2 , C 1 , and C 0 represent adjustable complex control coefficients)
- 11. The method of claim 10, further comprising adjusting the predistortion signal.
- 11. The method of claim 10, wherein the intermodulation distortion product is generated by a nonlinear amplifier and the predistortion signal is applied to the input radio frequency signal prior to amplification by the nonlinear amplifier.
- 15. The method of claim 14, wherein the predistortion signal is adjusted based on a detected difference between the amplifier output and the input radio frequency signal.
- The method of claim 10, wherein the generating of the predistortion signal comprises:Detecting an envelope of the input radio frequency signal; AndGenerating the predistortion signal based on the detected envelopeDistortion compensation method comprising a.
- 17. The method of claim 16, wherein generating the predistortion signal comprises clipping the detected envelope in the form of an approximate hyperbolic tangent.
- 18. The method of claim 17, wherein generating the predistortion signal comprises scaling the detected envelope.
- The method of claim 1, wherein the predistortion circuit,A predistortion circuit for generating a non-zero order term associated with the lower polynomial predistortion signal; AndA controller for generating a zero order term related to the low order polynomial predistortion signalDistortion compensation device comprising a.
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|Application Number||Priority Date||Filing Date||Title|
|US08/995,663 US6075411A (en)||1997-12-22||1997-12-22||Method and apparatus for wideband predistortion linearization|
|PCT/SE1998/002439 WO1999033170A1 (en)||1997-12-22||1998-12-22||Method and apparatus for wideband predistortion linearization|
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|KR100535273B1 true KR100535273B1 (en)||2005-12-09|
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|EP (1)||EP1042864B1 (en)|
|JP (1)||JP4279453B2 (en)|
|KR (1)||KR100535273B1 (en)|
|CN (1)||CN1203611C (en)|
|AR (1)||AR015506A1 (en)|
|AU (1)||AU752058B2 (en)|
|DE (1)||DE69832198D1 (en)|
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|US6285412B1 (en) *||1997-07-23||2001-09-04||Harris Corporation||Adaptive pre-equalization apparatus for correcting linear distortion of a non-ideal data transmission system|
|FI105506B (en) *||1998-04-30||2000-08-31||Nokia Networks Oy||Linearization procedure for amplifiers and amplifier arrangements|
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|US9913194B2 (en)||Method and system for baseband predistortion linearization in multi-channel wideband communication systems|
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