KR100248297B1 - Voltage source converter for thyristor phase control - Google Patents

Abstract

BACKGROUND OF THE INVENTION Field of the Invention The present invention relates to an AC-DC stationary power converter, to provide an effective converter that can be applied to a large power voltage source inverter system having regenerative braking. Bidirectional power is adopted by adopting a commutated thyristor and a bidirectional output DC current characteristic of zero mode, rectifier mode, and inverter mode by phase-control as a control method. In order to achieve the effect of flow capability and output DC voltage regulation, and also to achieve the effect of excellent reactive power and harmonic characteristics, the present invention is particularly suitable for converters for large power voltage source inverter systems with regenerative braking functions. , And voltage source type high voltage direct current transmission systems.

Description

Thyristor Phase-Controlled Voltage Source Converters

The present invention relates to a large power AC-DC static power converter (hereinafter, referred to as an AC-DC static power converter), in particular from the DC side to the AC side. A new thyristor phase-controlled voltage source converter with bidirectional power flow capability for energy recovery.

As the depletion of natural resources and the problem of environmental pollution become serious, the importance of energy saving technology is increasing. In motor drives, regenerative braking for recovering the kinetic energy of the motor and the sign is such a technique, which is also emerging as a more important technique.

In order to perform regenerative braking, it is not possible with a conventional diode rectifier with only unidirectional power flow capability, and the converter must have bidirectional power flow capability. Pulse Width Modulation (PWM) for current source inverter systems (ref. 1-3) and PWM voltage source converters (ref. 3-6) for voltage source inverter systems are converters with bidirectional power current capability. . A PWM current source converter is a converter that provides the characteristics of a bipolar output DC voltage, and performs bidirectional power current by changing the polarity of the output DC voltage, and the PWM voltage source converter performs a bidirectional output DC current. A converter providing the characteristics, it performs a bidirectional power flow by changing the direction of the output direct current. PWM current source converters and PWM voltage source converters have the advantage of bidirectional power current capability over diode rectifiers, as well as the added advantage of being able to control the output DC voltage and minimizing reactive power and harmonics problems. Has However, these converters have a problem that they are expensive and difficult to implement because of forced-commutated switching devices such as BJT, FET, IGBT, and GTO.

In high power applications, conventional line-commutated thyristor phase-controlled current source converters and current source inverter systems are mainly used (Refs. 3 and 7). Although pre-current thyristors are incomplete switching devices that are controlled only on turn on, they are nearly unrivaled in high power due to their low power loss and high power rating. . However, the current source inverter system is not only limited in its use, but also requires a large and heavy reactor on the DC side, and the thyristor phase-controlled current source converter has a problem of generating large reactive power and harmonics on the input side. . In a voltage source inverter system, a dual converter (in which two thyristor phase-controlled current source converters are connected in parallel with the DC sides opposite polarities) can be used. However, there is a problem that the price is doubled and also generates large reactive power and harmonics (Refs. 6 and 7).

[Reference 1] E.P. Wiechmann, P. D. Ziogas, and V. R. Stefanovic, "A novel bilateral power conversion scheme for variable frequency static power supplies," IEEE Trans. Industry Applications, vol. 21, no. 5, pp. 1226-1233, 1985.

[Ref. 2] T. Salzmann and A. Weschta, "Progress in voltage source inverters (VSIs) and current source inverters (CSIs) with modern semiconductor devices," in Proceedings of the IEEE IAS'87 Conference, pp. 577-583, 1987.

[Reference 3] B. K. Bose, "Power electronics-an emerging technology," in Proceedings of the IEEE IECON'88 Coference, pp. 501-508, 1988.

[Reference 4] B. T. Ooi, J. C. Salmon, J. W. Dixon, and A. B. Kulkarni, "A three-phase controlled-current PWM converter with leading power factor," IEEE Trans. Industry Applications, vol. 23, no. 1, pp. 78-84, 1987.

[Reference 5] M. Nishimoto, J. W. Dixon, A. B. Kulkarni, and B. T. Ooi, "An integrated controlled-current PWM rectifier chopper link for sliding mode position control," IEEE Trans. Industry Applications, vol. 23, no. 5, pp. 894-900, 1987.

[Ref. 6] S. B. Dewan and R. Wu, "A microprocessor-based dual PWM converter fed four quadrant ac drive system," in Proceedings of the IEEE IAS'87 Conference, pp. 755-759, 1987.

Ref. 7 N. Mohan, T. M. Undeland, and W. P. Robbins, Power Electronics: converters, applications, and design-2nd ed., John Wiley & Sons, Inc., 1995, pp. 138-153, 418-428, 460-468, 494-500.

Summary of the Invention The present invention is directed to a large power inverter system having a regenerative braking function, the disadvantages of the conventional thyristor phase-controlled current source converter and current source inverter system, and of the conventional dual converter for voltage source inverter system. SUMMARY OF THE INVENTION The present invention provides a converter that can be applied to a large power voltage source inverter system having a regenerative braking function, employing a line-current thyristor, and improving the bidirectional power flow capability by changing the direction of the output DC current. To provide a new thyristor phase-controlled voltage source converter having an output voltage regulation capability and having reactive power and harmonic characteristics superior to the conventional thyristor phase-controlled current source converter or the conventional dual converter. .

1 is a power circuit diagram of the present invention.

2 is a waveform showing a control method and operation in a zero mode of the present invention.

Figure 3 is a waveform showing the control method and operation in the rectifier mode of the present invention.

Figure 4 is a waveform showing the control method and operation in the inverter mode of the present invention.

5A, 5B, and 5C are phasor diagrams for the waveforms of FIGS. 2, 3, and 4, respectively.

6 is the operating region of V _an assuming pure sinusoidal i _a .

FIG. 7A is an equivalent circuit of FIG. 1 for waveform analysis. FIG.

7B and 7C are circuits for applying the superposition theorem.

8 shows waveforms in zero mode operation.

Figure 9 shows the thresholds in zero mode operation.

10 shows waveforms in rectifier mode operation.

11 shows waveforms in inverter mode operation.

를 나타낸 그림.12 shows the maximum phase control angle under the condition that V _d is constant.

Figure showing.

13 shows the operating region of V _an .

14a and 14b show the operating trajectories of V _an to obtain a regulated DC voltage.

15a and 15b show the driving trajectories of V _asn -V _an and I _a for FIG. 14a.

16 is a reactive power of the present invention.

17 is an input current harmonic of the present invention.

FIG. 18 shows the experimental results for FIG. 2.

FIG. 19 shows the experimental results for FIG. 3.

20 is an experimental result of FIG.

* Explanation of symbols for main parts of the drawings

Q ₁ , Q ₂ , Q ₃ , Q ₄ , Q ₅ , and Q ₆ : Thyristors

D ₁ , D ₂ , D ₃ , D ₄ , D ₅ , and D ₆ : Diodes

n: Neutral point of three-phase AC power

a, b, and c: three phase nodes

as, bs, and cs: nodes on the power side three phases

o: virtual neutral point on dc side

i _a , i _b , and i _c : alternating current three-phase current

i _d : DC current

i _l : load current

i _G1 , i _G2 , i _G3 , i _G4 , i _G5 , and i _G6 : gate drive currents of Q ₁ , Q ₂ , Q ₃ , Q ₄ , Q ₅ , and Q ₆ , respectively

V _d : DC voltage

: 위상 제어각

Phase control angle

: i_a의 기본파 성분의 영점-통과(zero-crossing)각

: Zero-crossing angle of the fundamental wave component of i _a

k: conversion factor

P: AC side active power

Q: AC side reactive power

THD: total harmonic distortion

<Construction of invention>

The present invention largely consists of a power circuit and a control method. 1 shows a power circuit of the present invention. As shown in FIG. 1, the power circuit of the present invention is a power converter that converts a three-phase AC voltage into a DC voltage.

(1) a positive thyristor, an upper thyristor having an anode connected to the positive terminal, a lower thyristor having an anode connected to a cathode of the upper thyristor, and a lower thyristor A negative terminal connected to a cathode of the upper thyristor, an anode of the upper thyristor, and an upper diode connected to a cathode and an anode of the cathode, respectively, and an anode of the lower thyristor Three thyristor voltage source half-bridges consisting of a lower diode with a cathode and an anode connected to and a cathode respectively, and an AC terminal connected to the cathode of the upper thyristor and the anode of the lower thyristor. ;

(2) three inductors connected between each phase of said three-phase alternating voltage and each alternating terminal of said three thyristor voltage source half-bridges;

(3) an output positive bus connecting the positive terminals of the three thyristor voltage sources half-bridges;

(4) an output negative bus connecting the negative terminals of the three thyristor voltage sources half-bridges; And

(5) a direct current capacitor connected between the output positive bus and the output negative bus. In FIG. 1, two DC capacitors are separated, but this separation is not really necessary. In FIG. 1, DC capacitors are separated into two to show the virtual neutral point o of the DC side, which is only necessary to explain the operation of the invention. It is expressed.

2, 3, and 4 show the control method of the present invention and its working waveforms. 2, 3, and 4 are for each zero mode, rectifier mode, and inverter mode. As shown in Figures 2, 3, and 4, the control method of the present invention

일 때 그 상에 연결된 상기 사이리스터 전압원 반-브리지의 아래쪽 사이리스터의 게이트 전류 펄스를 인가하는 영(zero) 모드 제어방법;(1) when the phase angle of each phase of the three-phase alternating voltage is 0, a gate current pulse of an upper thyristor of the thyristor voltage source half-bridge connected thereon is applied, and the phase angle of each phase of the three-phase alternating voltage is

A zero mode control method of applying a gate current pulse of a lower thyristor of the thyristor voltage source half-bridge connected thereon when the thyristor voltage source is connected thereto;

보다 클 때 그 상에 연결된 상기 사이리스터 전압원 반-브리지의 아래쪽 사이리스터의 게이트 전류 펄스를 인가하는 정류기(rectifier) 모드 제어 방법; 및(2) when the phase angle of each phase of the three-phase alternating voltage is greater than zero, a gate current pulse of an upper thyristor of the thyristor voltage source half-bridge connected thereon is applied, and the phase angle of each phase of the three phase alternating voltage is

A rectifier mode control method for applying a gate current pulse of the lower thyristor of the thyristor voltage source half-bridge connected thereon when larger; And

보다 작을 때 그 상에 연결된 상기 사이리스터 전압원 반-브리지의 아래쪽 사이리스터의 게이트 전류 펄스를 인가하는 인버터(inverter) 모드 제어 방법을 포함하는 것을 특징으로 하는 것이다.(3) when the phase angle of each phase of the three-phase alternating voltage is less than zero, a gate current pulse of an upper thyristor of the thyristor voltage source half-bridge connected thereon is applied, and the phase angle of each phase of the three phase alternating voltage

And an inverter mode control method for applying a gate current pulse of the lower thyristor of the thyristor voltage source half-bridge connected thereon when smaller.

<Action of invention>

Hereinafter, the operation of the present invention will be described in detail with reference to FIGS. 1, 2, 3, and 4. FIG.

t=0이 될 때, i_G1의 게이트 전류 펄스를 사이리스터 Q₁에 인가한다. 그러면, 직류 전압 V_d에 의해 Q₁이 턴 온 되며 D₄는 턴 오프 되어, 전압

_ao의 스위칭이 일어난다. i_a＜0인 동안에는 Q₁은 계속 도통된다. 그러나, i_a＞0이 되면, Q₁은 도통을 멈추고 D₁이 도통되기 시작한다. 그러나,

_ao는 변하지 않는다. 다음으로,

t=

가 될 때, i_G4의 게이트 전류 펄스를 사이리스터 Q₄에 인가한다. 그러면, 직류 전압 V_d에 의해 Q₄가 턴 온 되며 D₁은 턴 오프 되어, 전압

_ao의 스위칭이 일어난다. i_a＞0인 동안에는 Q₄는 계속 도통된다. 그러나, i_a＜0이 되면, Q₄는 도통을 멈추고 D₄가 도통되기 시작한다. 그러나 역시,

_ao는 변하지 않는다. 그러면, 동작의 한 주기가 끝나며,

_ao는 a상 전원 전압

_asn과 동 위상인 구형파가 된다. b상과 c상의 동작은 a상과 같으며, 다만 각각 2/3

및 4/3

만큼 위상 천이 된다. 그러면, 상전압

_an은 잘 알려진 6-펄스 파형을 이루며, 이 전압은 전원 전압

_asn과 함께 a상 전류 i_a를 만든다.

_asn과

_an이 동 위상이기 때문에 i_a는

_asn보다

/2만큼 지연되며, 아무런 유효 전력 조류가 이루어지지 않는다. 그리고 직류 전류 i_d는 0의 평균값을 갖는다. 이것이 영(zero) 모드의 동작이다.In FIG. 2, just before wt = 0, the α phase current is ia <0 and flows through diode D ₄ . And

When t = 0, the gate current pulse of i _G1 is applied to the thyristor Q ₁ . Then, Q ₁ is turned on by the DC voltage V _d and D ₄ is turned off,

Switching of _ao occurs. Q ₁ continues to conduct while i _a <0. However, when i _a > 0, Q ₁ stops conducting and D ₁ begins to conduct. But,

_ao does not change. to the next,

t =

When is applied, the gate current pulse of i _G4 is applied to the thyristor Q ₄ . Then, Q ₄ is turned on by the DC voltage V _d and D ₁ is turned off,

Switching of _ao occurs. Q ₄ continues to conduct while i _a > 0. However, when i _a <0, Q ₄ stops conducting and D ₄ begins to conduct. But too,

_ao does not change. Then, one cycle of action ends

_ao is a phase power supply voltage

_A square wave is in phase with _asn . Phase b and phase c are the same as phase a, except 2/3 each.

And 4/3

As much as the phase shift. Then, phase voltage

_an forms the well-known 6-pulse waveform, which is the supply voltage

Create a phase current i _a with _asn .

_asn and

_{Since an} is in phase, i _a

than _asn

Delayed by / 2, and no active power flow occurs. DC current i _d has an average value of zero. This is the operation in zero mode.

만큼 지연(delay)되면, 전압

_ao는

_asn에 비해

만큼 지연된다. 그러면, 전류 i_a는 왼쪽으로 천이 되어 양(positive)의 전력 조류가 형성된다. 그리고 직류 전류 i_d는 양의 평균값을 갖는다. 이것이 정류기(rectifier) 모드의 동작이다. 역으로, 인가된 게이트 전류 펄스들이 제4도에 보인 것과 같이 위상 제어각

만큼 전진(advance)되면, 전압

_ao는

_asn에 비해

만큼 전진된다. 그러면, 전류 i_a는 오른쪽으로 천이 되어 음(negative)의 전력 조류가 형성된다. 그리고 직류 전류 i_d는 음의 평균값을 갖는다. 이것이 인버터(inverter) 모드의 동작이다.Phase control angle as applied gate current pulses as shown in FIG.

When delayed, voltage

_ao is

compared to _asn

Delayed by. The current i _a then transitions to the left to form a positive power flow. And the direct current i _d has a positive mean value. This is the operation in rectifier mode. Conversely, the phase control angle as applied gate current pulses are shown in FIG.

As advanced, the voltage

_ao is

compared to _asn

As advanced. The current i _a then transitions to the right to form a negative power current. DC current i _d has a negative average value. This is the operation of the inverter mode.

_asn,

_ao의 기본파 성분, 및 i_a의 기본파 성분이다. 그리고

₁은 i_a의 기본파 성분의 영점-통과(zero-crossing)각이다.I _a는V _asn-V _an에 비해

/2만큼 지연되며,V _asn에 비해서는

₁만큼 지연된다.Figures 5a, b and 5c show pacers 2, 3, and 4, respectively. In Figures 5a, b and 5c, V _asn , V _an , and I _a are respectively

_asn ,

fundamental wave component of _ao and fundamental wave component of i _a . And

₁ is the zero-crossing angle of the fundamental wave component of i _a . I _a is compared to V _asn - V _an

Delayed by / 2, compared to V _asn

₁ is delayed by.

<Operation area>

t=

에서 게이트 펄스 전류 i_G1이 사이리스터 Q₁에 인가될 때 전류(commutation)가 실패하지 않기 위해서는 전류 i_a는 음(negative)이어야 한다. 마찬가지로,

t=

+

에서 게이트 펄스 전류 i_G4가 사이리스터 Q₄에 인가될 때 역시 전류(commutation)가 실패하지 않기 위해서는 전류 i_a는 양(positive)이어야 한다.In the operation of the present invention there is an operating region in which the commutation does not fail. In Figures 2, 3, and 4,

t =

In order for the commutation not to fail when the gate pulse current i _G1 is applied to the thyristor Q ₁ , the current i _a must be negative. Likewise,

t =

+

When the gate pulse current i _G4 is applied to the thyristor Q ₄ , the current i _a must be positive so that the commutation does not fail.

_an에 비해 지연되어야 한다는 것과 동일하다. 그러면, 위의 조건은 제6도에 보인 것과 같은 페이서도로 표현될 수 있다. 제6도에서, 원(circle) 안의 어둡게 칠해진 부분이V _an의 동작 영역이다. 그러나, i_a의 실제의 모양은 정현파가 아니기 때문에 실제의 동작 영역을 구하기 위해서는 파형 해석이 필요하다.If the premise that now, for the sake of convenience current i _a pure sine wave, the condition of the above, the current i _a

Equivalent to delaying over _an . Then, the above condition can be expressed as a pacer as shown in FIG. In FIG. 6, the darkened portion in the circle is the operating region of V _an . However, since the actual shape of i _a is not a sine wave, waveform analysis is required to obtain the actual operating area.

_an은 컨버터에 의해 발생된 전압이다. 제7b도와 7c에는 제7a도의 회로에 중첩(superposition) 정리를 적용하기 위한 회로들을 나타내었다. 중첩 정리에 의해 전류 i_a는 i₁과 i₂의 차로서 주어진다.FIG. 7A shows an equivalent circuit of FIG. 1 for waveform analysis. In Figure 7a

_an is the voltage generated by the converter. 7b and 7c show circuits for applying superposition theorem to the circuit of FIG. 7a. By superposition theorem, the current i _a is given as the difference between i ₁ and i ₂ .

8 shows waveforms in zero mode operation. In the eighth waveform in Figure i ₁ is not changed since the current i ₁ is caused by the fixed (fixed) power. However, the waveform of i ₂ increases in magnitude as the DC voltage V _d increases. Showed it on ninth degree. Then, when the peak of i ₂ reaches the top of i _1, the threshold is for the commutation not to fail, because when i ₂ becomes the peak, a gate pulse is applied to the thyristor, where i ₂ becomes If it is larger than i ₁ , the commutation fails. The maximum value of the DC voltage V _d when i ₂ reaches the critical point is given as follows.

_asn의 크기이다. 지금, 변환비(conversion factor) k를 다음과 같이 도입하는 것이 편리하다.Where V _s is the supply voltage

The size of _asn Now, it is convenient to introduce the conversion factor k as follows.

_asn의 기본파 성분이다. 그리고, V₁은 직류 전압 V_d와 항상 다음과 같은 관계에 있다.Where V ₁ is the converter voltage

The fundamental wave component of _asn . And V ₁ always has the following relationship with the DC voltage V _d .

Then, the maximum value of the conversion ratio k is obtained from the above equations as follows.

_asn의 파형이 순수한 정현파라면, k의 최대값은 1이 된다. 그러나 실제로는,

_asn에 있는 고조파 때문에 변환비 k는 1이 되지 못한다.if

_If the waveform of _asn is a pure sine wave, then the maximum value of k is 1. But in reality,

Because of the harmonics in _asn , the conversion ratio k is not _equal to one.

를 이동시켜서 i₂의 첨두가 i₁과 만나는 점이 전류(commutation)가 실패하지 않기 위한 임계점이다. 제12도에 그것을 나타내었다. i₂가 임계점에 도달될 때의 최대 위상 제어각

는 다음과 같이 주어진다.10 and 11 show waveforms in rectifier mode and inverter mode operation, respectively. 10 and 11, the waveform of i ₁ does not change, and the waveform of i ₂ does not change under the condition that V _d is constant. However, the waveform of i _{2 changes} according to the phase angle. Then, phase control angle

Moved to a critical point for the peak of the i ₂ it is not the point is a current (commutation) of intersection with i ₁ fails. It is shown in FIG. phase control angle when i ₂ reaches the critical point

Is given by

The above equation is a circular function and in rectangular coordinates it is written as

)이며 y=ksin(

)이다. 제13도에 이 조건을 나타내었다. 원(circle) 안의 어둡게 칠해진 부분이V _an의 동작 영역이다. 비교를 위해서, 순수한 정현파를 전제한 제6도의 원을 점선으로 나타내었다.Where x = kcos (

) And y = ksin (

)to be. This condition is shown in FIG. The darkened area in the circle is the operating region of V _an . For comparison, the circle of FIG. 6 assuming pure sinusoids is indicated by a dotted line.

<Characteristic>

_an을 직접 제어할 필요는 없다. 다만, 위상 제어각

제어에 의해 유효 전력을 증가 또는 감소시켜 직류 전압 V_d를 원하는 값으로 제어(regulate)함으로써 충분하다. 그러면,V _an은 자동적으로 운전 궤적을 추종하며 전력 조류의 크기와 방향도 역시 부하 조건에 따라 자동적으로 제어된다.As described above, the present invention is operable in the area shown in FIG. We can then select or design the operating trajectories of V _an to achieve the desired characteristics. If we want to get a regulated output DC voltage, the magnitude of V _an must be controlled constantly. Then, the driving trajectory of V _an is arced as shown by the thick line in FIG. 14A and FIG. 14B. 14A and 14B are examples of driving trajectories when the conversion ratio k is 0.8 and 0.85, respectively. To make V _an follow the driving trajectory

You do not need to control _an directly. Phase control angle

It is sufficient by regulating the DC voltage V _d to a desired value by increasing or decreasing the active power by the control. V _an then automatically follows the driving trajectory and the magnitude and direction of the power current is also automatically controlled according to the load conditions.

Bi-Directional Power Flow and Output DC Voltage Regulation

As described above, the present invention employs a line-commutated thyristor as the switching element, and zero-mode, rectifier mode, and inverter by phase-control as a control method. By providing the bidirectional output DC current characteristics of the mode has the effect of enabling the bidirectional power flow, and also has the effect of enabling the regulation of the output DC voltage through the control of the power flow.

<Reduction of reactive power>

Figures 15a and 15b show the operating trajectories of V _asn - V _an and I _a for Figure 14a. As shown in FIG. 15B, the present invention always generates reactive power on the ground. However, the reactive power of the present invention is less at most operating conditions than conventional thyristor phase-controlled current source converters or dual converters, as shown in FIG. (In Fig. 16, the inductor assumed at k = 0.85 is not the same as the inductor assumed at k = 0.8, and is appropriately reduced to equalize the maximum active power characteristic. Assuming the same inductor, the reactive power is reduced but the maximum effective Power is also reduced.)

<Reduction of harmonics>

에서의 i_a의 파형 및 다음의 정의로부터 얻은 것이다.In the present invention, although the shape of the input current i _a varies greatly depending on the load condition, the harmonic component is constant regardless of the load condition under the condition that V _d or k is constant. 8, 10, and 11 clearly show this. Figure 17 shows the total harmonic distortion THD of the input current i _a of the present invention as a function of k. THD up

From the waveform of i _a in and the following definition:

For comparison, the THD of a conventional thyristor phase-controlled current source converter or dual converter is also indicated by dotted lines. As shown in FIG. 17, the harmonics of the input current of the present invention are significantly less than that of conventional thyristor phase-controlled current source converters or dual converters. The THD of the present invention does not exceed 10% when k is 0.8 and does not exceed 13% when k is 0.85.

<About k value>

16 and 17 show the characteristics of the conversion ratio k from 0.8 to 0.85. As the value of k decreases, the harmonics of the input current also decrease, but the reactive power increases. On the other hand, when k is about 0.88 or more, the larger the value of k, the worse the harmonic characteristics of the reactive power and the input current. As a result, around 0.8 to 0.85 is recommended as the area of k for obtaining good performance.

<Experiment result>

Experimental results for FIGS. 2, 3, and 4 are shown in FIGS. 18, 19, and 20, respectively. The converters used in the experiments were of small power designed for experimentation. The results of FIGS. 18, 19, and 20 well confirm the operation of the present invention.

<Application>

As described above, the present invention has the capability of bidirectional power flow, and the output DC voltage can be controlled to a desired value. The reactive power and harmonics of the present invention are much less than conventional thyristor phase-controlled current source converters or dual converters. Consequently, the present invention is expected to be particularly useful as a converter for voltage source inverter systems with regenerative braking at high power.

Since the present invention can also operate as a line-commutated inverter, it can be used in a voltage source type high voltage direct current (HVDC) transmission system (Refs. 8 and 9). The voltage source high voltage direct current transmission system has advantages over the conventional current source high voltage direct current transmission system in particular in a multi-terminal configuration.

[Reference 8] B. T. Ooi and X. Wang, "Voltage angle lock loop control of the boost type PWM converter for HVDC application," IEEE Trans. Power electronics, vol. 5, no. 2, pp. 229-235, 1990.

[Reference 9] B. T. Ooi, "Pulse width modulation power transmission system," U. S. Patent No. 4,941,079, 1990.

Claims (1)

In a power converter that converts a three-phase AC voltage into a DC voltage, (1) a positive thruster, an upper thyristor having an anode connected to the positive terminal, a lower thyristor having an anode connected to a cathode of the upper thyristor, and a negative electrode of the lower thyristor ( A negative terminal connected to the cathode, an upper diode connected to a cathode and an anode of the upper thyristor anode and a cathode, respectively an anode and a cathode of the lower thyristor three thyristor voltage source half-bridges each having a cathode connected to a cathode and an anode, and an AC terminal connected to a cathode of the upper thyristor and an anode of a lower thyristor; (2) three inductors connected between each phase of said three-phase alternating voltage and each alternating terminal of said three thyristor voltage source half-bridges; (3) an output positive bus connecting the positive terminals of the three thyristor voltage sources half-bridges; (4) an output negative bus connecting the negative terminals of the three thyristor voltage sources half-bridges; (5) a direct current capacitor connected between the output positive bus and the output negative bus; (6) When the phase angle of each phase of the three-phase alternating voltage is 0, a gate current pulse of an upper thyristor of the thyristor voltage source half-bridge connected thereon is applied, and the phase angle of each phase of the three-phase alternating voltage is

A zero mode control method of applying a gate current pulse of a lower thyristor of the thyristor voltage source half-bridge connected thereon when the thyristor voltage source is connected thereto; (7) When the phase angle of each phase of the three-phase alternating voltage is greater than zero, a gate current pulse of an upper thyristor of the thyristor voltage source half-bridge connected thereon is applied, and the phase angle of each phase of the three phase alternating voltage is

A rectifier mode control method for applying a gate current pulse of the lower thyristor of the thyristor voltage source half-bridge connected thereon when larger; And (8) when the phase angle of each phase of the three-phase alternating voltage is less than zero, a gate current pulse of an upper thyristor of the thyristor voltage source half-bridge connected thereon is applied, and the phase angle of each phase of the three phase alternating voltage

And an inverter mode control method for applying a gate current pulse of the lower thyristor of the thyristor voltage source half-bridge connected thereon when smaller.

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