KR100218521B1 - Switching mode power suppling apparatus for generating high voltage - Google Patents

Abstract

end. The technical field to which the invention described in the claims belongs:

Switched-mode power supplies capable of high voltage generation

I. The technical problem the invention is trying to solve:

The present invention provides a switching mode power supply that enables high voltage generation with a simple circuit configuration in a switching mode power supply.

All. The gist of the solution of the invention:

In a switched-mode power supply, current resonant waveforms are used to directly obtain high voltages with high boost ratios.

la. Important uses of the invention:

Used in circuits that require high voltage supply.

Description

Switching power supply for high voltage generation {SWITCHING MODE POWER SUPPLING APPARATUS FOR GENERATING HIGH VOLTAGE}

The present invention relates to a switching mode power supply (SMPS), and more particularly to a switching mode power supply capable of generating a high voltage.

Typically, televisions and monitors, which are home appliances, include cathode ray tubes used as video display devices. The cathode ray tube accelerates the hot electrons generated from the electron gun of the cathode ray tube, and passes them through the focusing and deflection portions to sprinkle the fluorescent surface to emit light of the fluorescent surface of the portion. In order to accelerate hot electrons in such a cathode ray tube, a high voltage of 25 to 30 kV is generally required. In order to provide such a high voltage, a high voltage generator is mounted on a television or a monitor.

1 shows an example of a high voltage generating device mounted on a television or a monitor. Referring to FIG. 1, a conventional high voltage generator includes a rectifying unit 10 which rectifies an AC input power Vin to a direct current, and a constant voltage power supply unit that boosts a voltage applied from the rectifying unit 10 to a stable medium voltage (eg, 130 V). 20 and the high voltage controller 30 for controlling the voltage applied from the constant voltage power supply unit 20 to the divided voltage of the high voltage output Vout for the stable output of the high voltage output Vout, and the high voltage generating the high voltage by boosting the voltage applied by the high voltage controller 30 secondly. It consists of the generator 40. In detail, the rectifier 10 includes a bridge diode circuit 12 and a capacitor C1 having one end grounded and the other end connected to the output terminal of the bridge diode circuit 12. The AC input power Vin is full-wave rectified in the bridge diode circuit 12, smoothed in the capacitor C1, and applied to the constant voltage power supply 20. The inductor L1 of the constant voltage power supply unit 20 receives a DC power output from the rectifier 10. The inductor L1 is controlled on / off by a certain duty cycle by the SMPS controller 22 and is connected in series with the collector terminal of the transistor Q1 which performs the switching operation. The emitter terminal of the transistor Q1 is grounded. have. Transformer T1 consists of inductor L1, the primary winding, and inductors L2, L3, the secondary winding. Based on the on / off operation of the transistor Q1, the voltage applied to the primary winding inductor L1 of the transformer T1 is boosted by the winding ratio N2 / N1 to be induced into the secondary winding inductors L2 and L3. The secondary windings L2, L3 are separated from each other and are connected in series with the diodes D1 or D2, respectively, in parallel with the capacitor C2 or C3, one end of which is then grounded. Voltages induced in the secondary windings L2 and L3, respectively, are rectified by the diodes D1 or D2 and smoothed by the capacitors C2 or C3 and output to the medium voltages V1 and V2, respectively. The high voltage controller 30 has a transistor Q2 receiving the voltage V1 output from the constant voltage power supply unit 20 as a collector terminal, a comparator 32 having an output terminal connected to the base terminal of the transistor Q2, and a capacitor C4 having one end grounded in parallel with the emitter terminal of the transistor Q2. Consists of The reference voltage Vref, which is a battery power source, is applied to the non-inverting input terminal of the comparator 32, and the divided voltage obtained by dividing the high voltage output Vout, which is the output of the high voltage generator 40, by a constant voltage, is applied to the inverting input terminal of the comparator 32. The comparator 32 compares the magnitude of the voltage between the reference voltage Vref and the divided voltage of the high voltage output Vout, amplifies the voltage difference, and applies an operating voltage to the base terminal of the transistor Q2. The transistor Q2 drops the voltage V1 output from the constant voltage power supply 20 in proportion to the magnitude of the operating voltage applied by the comparator 32. The higher the voltage of the high voltage output Vout, the greater the difference between the divided voltage of the high voltage output Vout and the reference voltage Vref in the comparator 32, so that the voltage drop of the voltage V1 applied to the transistor Q2 becomes larger. After that, the voltage of the dropped V1 is smoothed at the capacitor C4 and applied to the high voltage generator 40. The high voltage generator 40 includes an inductor L4 that receives a voltage applied from the high voltage controller 30, and a switching transistor Q4 having one end connected to the inductor L4 and the collector terminal grounded. In addition, the high voltage generator 40 includes an inductor L6 receiving the voltage V2 output from the constant voltage power supply unit 20, and a switching transistor Q3 having one end connected to the inductor L6 and a collector terminal grounded. A horizontal synchronization signal is applied to the base terminal of the switching transistor Q3. The horizontal synchronizing signal is a signal for synchronizing with a signal for horizontally scanning hot electrons in the cathode ray tube. The switching transistor Q3 is turned on by the application of the horizontal synchronizing signal, and the voltage V2 on the inductor L6 which is the primary side winding of the transformer T4 is induced to the inductor L7 which is the secondary side winding. One end of the inductor L7, the secondary winding of transformer T4, is grounded, and the other end is connected to the base end of transistor Q4 via a load resistor R1. Therefore, when the horizontal synchronizing signal is applied, the switching transistor Q3 is turned on and a current is induced in the transformer T4, which eventually turns on the switching transistor Q4. The switching transistor Q4 is configured in parallel with the retrace diode D3, the resonant capacitor C5, and the inductor L5 which is the primary winding of the transformer T3. When the switching transistor Q4 is turned on, the inductor L4, which is the primary winding of the transformer T2, is charged with the current applied by the high voltage controller 30. When the transistor Q4 is turned off, the current charged in the inductor L4 is discharged to the capacitor C5 and the inductor L5. . Capacitor C5 is charged by the discharge current of inductor L4 when transistor Q4 is turned off, but discharges the charging current to the ground terminal connected to the emitter terminal of transistor Q4 when transistor Q4 is turned on. Inductor L5 is charged by inductor L4 when transistor Q4 is turned off and discharges to the path of diode D3 when transistor Q4 is turned on. Transformer T2 configures the inductor L4 as the primary winding and the inductors L8, L9, L10, and L11 connected in series as the secondary winding. Diodes D4, D5, and D6 are connected between the inductors L8, L9, L10, and L11, and a diode D7 is connected to the output terminal of the inductor L11. The high voltage is induced in the secondary winding inductors L8, L9, L10, and L11 according to the change in the voltage of the inductor L4, the primary winding of the transformer T2, and the induced high voltage is charged in the high voltage charging capacitor C6 to output the high voltage Vout. This will occur. The high voltage is divided by the voltage divider resistors R2 and R3, and the divided voltage is applied to the inverting input terminal of the comparator 32 in the high voltage controller 30.

However, in the conventional high voltage generator as described above, a complex circuit is configured to generate a medium pressure in the constant voltage power supply unit 20 first, to adjust the medium pressure in the high voltage control unit 30, and to step up the medium pressure again in the high voltage generation unit 40 again. There was a problem that a configuration and a large number of devices are required. In addition, there is a problem that a lot of power is lost when the switching elements on / off.

Accordingly, an object of the present invention is to provide a device for generating a high voltage with a simple circuit configuration.

Still another object of the present invention is to provide a high voltage generating device capable of minimizing lost power.

In order to achieve the above object, the present invention is characterized in that a high high voltage having a high boost ratio is directly obtained by using a current resonance waveform in a switched mode power supply.

1 is a circuit diagram of a conventional high voltage generator

2 is a circuit diagram of a high voltage generation switching mode power supply according to an embodiment of the present invention;

3 is a signal waveform diagram of circuits of the high voltage generation switching mode power supply of FIG.

Hereinafter, exemplary embodiments of the present invention will be described in detail with reference to the accompanying drawings.

FIG. 2 is a circuit diagram of a high voltage generation switching mode power supply device according to an embodiment of the present invention, and FIG. 3 is a signal waveform diagram of circuits of the high voltage generation switching mode power supply device of FIG.

Referring to FIG. 2, the switching mode power supply apparatus according to an exemplary embodiment of the present invention is largely composed of a rectifying unit 10 for rectifying the AC input power Vin to DC and a high voltage generating unit 50 for generating a high voltage. The high voltage generator 50 outputs a high voltage by using the current ?? voltage resonator 52 for resonating the power applied from the rectifier 10 and the resonant current ?? voltage of the current ?? voltage resonator 52. 54, a voltage divider R2 and R3, a feedback circuit 56, an SMPS control unit 60, a first drive 62 and a second drive 64. The voltage dividing resistors R2 and R3 divide the high voltage output from the high voltage output unit 54. The feedback circuit 56 receives the divided voltage of the high voltage and applies the divided voltage level to the SMPS controller 60. Accordingly, the SMPS controller 60 applies an operating frequency according to the applied voltage to the first drive 62 and the second drive 64. The SMPS controller 60 controls the first drive 62 and the second drive 64 according to the operating frequency control of the feedback circuit 56. In response to the signal from the SMPS controller 60, the first drive 62 and the second drive 64 control the switching field effect transistors Q11 and Q12 on / off, respectively. Referring to the configuration of the switching mode power supply according to the present invention in more detail, the rectifier 10 is composed of a bridge diode circuit 12 and a capacitor C1 connected to the output terminal of the bridge diode circuit 12 and the other end is grounded, as in the prior art. The AC input power Vin is full-wave rectified in the bridge diode circuit 12, smoothed in the capacitor C1, and applied to the current-voltage resonator 52 in the high voltage generator 50 as the DC input power. The current-voltage resonator 52 may include a first N-channel field effect transistor Q11 (hereinafter, referred to as a first N-channel transistor) that receives a DC input power applied from the rectifier 10 as a drain terminal, and the first N-channel transistor Q11. A second N-channel field effect transistor (hereinafter referred to as a second N-channel transistor) Q12 connected to a source terminal and a drain terminal is included. The source terminal of the second N-channel transistor Q12 is grounded. The first N-channel transistor Q11 and the second N-channel transistor Q12 are configured in parallel with the retrace diodes D11 and D12, respectively. A capacitor C11, one end of which is grounded, is connected to the connection node 58 of each of the first and second N-channel transistors Q11 and Q12, and one end of the inductor L11 is connected. The other end of the inductor L11 is connected to a capacitor C12 having a larger capacity than the capacitor C11, and the other end of the capacitor C12 is grounded. In the high voltage output unit 54, the primary winding is composed of the inductor L11, the secondary winding is composed of the inductors L13 and L14 which are reverse to the current IL of the inductor L11, and the inductors L15 and L16 which are forward, and each inductor L13, L14, L15 and L16 is High voltage transformer T5 is connected in series. Diodes D14, D15, and D16 are connected between each inductor L13, L14, L15 and L16 to rectify the output of transformer T5. Diode D13 is connected in series at the output of inductor L13. The high voltage output unit 54 includes a high voltage charging capacitor C13 charged by a current induced in the inductors L13 and L14, and a high voltage charging capacitor C14 charged by the inductor L15 and a current induced in the L16 in series. have. Both ends of the capacitor C13 are connected to the organic current output terminal of the inductor L13 and the organic current input terminal of the inductor L14. Both ends of the capacitor C14 are connected to the organic current output terminal of the inductor L15 and the organic current input terminal of the inductor L16. The organic current input of inductor L16 is grounded. On the other hand, in the high voltage generator 50, the divider resistors R2 and R3 are connected in series between the output terminal of the high voltage output unit 54 and the ground terminal to divide the high voltage output Vout. The divided voltage of the high voltage output Vout by the voltage dividing resistors R2 and R3 is applied to the feedback circuit 56. The feedback circuit 56 adjusts the operating frequency of the SMPS controller 60 according to the divided voltage of the applied high voltage output Vout. The SMPS controller 60 operates the first drive 62 as a high signal and operates the second drive 64 as a low signal according to the operating frequency control of the feedback circuit 56. The first drive 62 and the second drive 64 apply operating voltages to the gate terminals of the first and second N-channel transistors Q11 and Q12 by the SMPS controller 60, respectively.

Hereinafter, the operation of an embodiment of the present invention will be described with reference to FIGS. 2 and 3. First, the first drive 62 controlled by the high signal of the SMPS controller 60 operates to turn on the first N-channel transistor Q11 during the period t0 to t1 as shown in FIG. 3. When the first N-channel transistor Q11 is turned on, the DC current I1 applied by the rectifier 10 is charged to the capacitor C11, the inductor L11 and the capacitor C12 through the first N-channel transistor Q11. Referring to FIG. 3, it can be seen that the input current I1 flowing during the period t0 to t1 gradually increases from 0 according to the current charging characteristic of the inductor. Thereafter, the first N-channel transistor Q11 is turned off during the period t1 to t2, and the second N-channel transistor Q12 is also turned off during the period t1 to t2. During the t1 to t2 sections in which the first and second N-channel transistors Q11 and Q12 are all turned off, the current I2 of the capacitor C11 charged during the t0 to t1 sections is discharged through the paths of the inductor L11 and the capacitor C12. This is because the capacitor C11 is set to a smaller capacity than the capacitor C12 and thus is charged only for a shorter time than the capacitor C12. The discharge current I2 generated during the t1 to t2 periods is represented by a half-cycle voltage resonance by an LC resonant circuit composed of a capacitor C11, an inductor L11, and a capacitor C12 in series. When the current charged in the capacitor C11 is completely discharged at time t2, the inductor L11 charged by the direct current input current I1 during the t0 to t1 section and the discharge current I2 of the capacitor C11 during the t1 to t2 section starts to discharge. The discharge in the inductor L11 is continued for the period t2 to t3 in FIG. 3, wherein the discharge current I3 is discharged in the forward path of the capacitor C12 and the diode D12. The discharge current I3 gradually decreases as shown in FIG. 3 and becomes almost zero at time t3. At time t3, the second N-channel transistor Q12 is turned on. The second N-channel transistor Q12 is turned on by the operation of the second drive 64 driven by the low signal of the SMPS controller 60. As shown in FIG. 3, the second N-channel transistor Q12 is turned on during the period t3 to t4, and the current I4 charged in the capacitor C12 is discharged through the path of the second N-channel transistor Q12 during the period t3 to t4. . The discharge current I4 flowing during the period t3 to t4 is opposite to the direction of the discharge current I3 of the input current I1 and the inductor L11, and the inductor L11 is charged by the discharge current I4. Referring to FIG. 3, it can be seen that the discharge current I4 flowing during the period t3 to t4 gradually increases in the negative direction of the input current I1 at zero. Thereafter, the second N-channel transistor Q12 is turned off during the period t4 to t5, and the first N-channel transistor Q11 is also turned off during the period t4 to t5. During the period t4 to t5 in which the first and second N-channel transistors Q11 and Q12 are both turned off, the discharge current I4 of the capacitor C12 is discharged through the paths of the inductor L11 and the capacitor C11. The discharge current I4 during the periods t4 to t5 is opposite to the discharge current I2 generated during the periods t1 to t2 and is in the form of a half cycle voltage resonance while charging the capacitor C11. Thereafter, when the capacitor C11 reaches the time t5 at which the capacitor C11 is fully charged, the inductor L11 charged by the discharge current I4 of the capacitor C12 during the period t3 to t4 and the period t4 to t5 starts to discharge. The discharge of the inductor L11 is continued for the period t5 to t6 in FIG. 3, wherein the discharge current I5 is discharged in the forward path of the diode D11. As shown in FIG. 3, the discharge current I5 gradually decreases in the negative direction of the input current I1, and becomes almost zero at time t6. At time t6, the first N-channel transistor Q11 is turned on again by the operation of the first drive 62 driven by the high signal of the SMPS controller 60, and the subsequent processes are repeated.

In the above processes, the waveform of the current IL flowing through the inductor L11 is represented by a sine wave having a period of t0 to t6 as shown in FIG. 3. Therefore, during the periods t0 to t3 in which the current IL of the inductor L11 is forward, current is induced in the inductors L13 and L14, which are the secondary windings of the transformer T5 of the high voltage output unit 54, as shown in FIG. Transformer T5 is for boosting, so the induced current charges capacitor C13 to the boosted voltage. During periods t3 to t4 in which the current IL flowing in the inductor L11 is in the reverse direction, current is induced in the inductors L15 and L16, which are the secondary windings of the transformer T5, as shown in FIG. The induced current charges capacitor C14 to a boosted voltage. The capacitors C13 and C14 used for charging the boosted voltage are connected in series to generate a high voltage output Vout of the double voltage, and the output current Io of the high voltage output unit 54 is each inductor L13 and L14 as shown in FIG. 3. , The sum of the currents induced in L15 and L16.

The feedback circuit 56 receiving the divided voltage of the high voltage output Vout adjusts an operating frequency according to the magnitude of the voltage applied to the SMPS controller 20 to stabilize the high voltage output Vout. The higher the voltage of the high voltage output Vout is, the shorter the operation frequency period of the SMPS controller 60 is, thereby shortening the operation cycles of the first and second drives 62 and 64 operated by the SMPS controller 60. Therefore, since the on and off cycles of the first and second N-channel transistors Q11 and Q12 controlled by the operations of the first and second drives 62 and 64 are shortened, the amount of current charged and discharged in each element is also reduced, resulting in high voltage output. The voltage at Vout is also lowered.

As described above, in the exemplary embodiment of the present invention, the high voltage output circuit of the double voltage is formed by the series connection of the capacitors C13 and C14, but the corresponding windings of the two capacitors are separately configured to obtain two high voltage outputs of the non-double voltage. It is also possible to configure a circuit for this. In addition, although one embodiment of the present invention has been described with respect to a high voltage generating device in a cathode ray tube such as a television or a monitor, the present invention can be applied to both a power supply circuit using a magnetron driving voltage or other high voltage.

As described above, in the present invention, since the high voltage ratio of the high boost ratio is directly obtained in the switching mode power supply, there is an advantage of enabling high voltage generation with a small component and a simple configuration. In addition, the loss power generated in the on / off period of the switching elements in the conventional high voltage generation process has the advantage that can be minimized by the zero voltage or zero current switching of the resonance current.

Claims (1)

In the switching mode power supply comprising a boost transformer for generating high voltage, A first switching element for supplying or interrupting DC power by a first drive signal; A second switching element connected between the first switching element and ground and supplying or blocking current through the first switching element by a second drive signal; A first capacitor connected between the contact point of the first switching element and the second switching element and ground; A first end of the transformer is connected to a contact point of the first switching element and the second switching element, and a second end of the second capacitor is connected to a first end of a second capacitor having a larger capacity than the first capacitor; The other end of the ground, First diodes connected to both ends of the first switching element in a direction of preventing input of the DC power source; A second diode connected to both ends of the second switching element in a direction of preventing input of the DC power source; Connecting the secondary winding of the transformer through third and fourth capacitors, Connecting the middle tab of the secondary winding of the transformer to the contact point of the third capacitor and the fourth capacitor, Connecting the first and second resistors for the partial pressure in parallel with the secondary winding of the transformer; Connecting a feedback circuit to a contact point of the first resistor and the second resistor to connect a power supply controller for generating two different frequencies according to the feedback voltage; A first drive circuit configured to output the first drive signal according to a first output of the power supply controller; And a second drive circuit configured to output the second drive signal according to a second output of the power supply controller.

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