JPS6362702B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention is a spectrum analyzer capable of highly sensitive C/N measurement of a spectrum over a wide frequency range, and particularly relates to an improvement of a local oscillator used in this spectrum analyzer.

<Background of the Invention> Conventionally, in order to measure the output signal of an oscillator, etc. with high sensitivity, the local oscillator that makes up the spectrum analyzer is configured with a high-frequency oscillator and a frequency divider to minimize noise. One idea is to frequency-divide the output of a high-frequency oscillator to increase its purity, and then feed the resulting output to a frequency converter.

An example is shown in FIG. This is performed by the high frequency oscillator 19, frequency dividers 21, 22, and amplifier 23.
It constitutes a local oscillator. 20 is a second local oscillator. 25 is a center frequency setter and an adder 26
The high frequency oscillator 19 is controlled through. 24 is a sawtooth wave oscillator that supplies a sweep signal to the adder 26 and the X axis of the cathode ray tube 27.

The high frequency oscillator 19 oscillates a high frequency of, for example, 2200 to 3700 MHz, and this output is divided into 1/3 by a frequency divider 21 using a tunnel diode, and further divided into 1/4 by a frequency divider 22 using a flip-flop. , gives a signal of, for example, 180 to 300 MHz to the mixer 13. In the mixer 13, this signal and, for example, 50Hz to 120M
Hz input frequency mixed with ₀ , for example 180MHz
Convert to the first intermediate frequency ₁ of . Furthermore, mixer 15
With this first intermediate frequency ₁ , for example 150M
Hz from the second local oscillator 20, and converts it into a second intermediate frequency ₂ of, for example, 30 MHz. This is amplified by a logarithmic amplifier 17, and a detector 18
Detects the wave and converts it to a DC signal and sends it to the Y of the cathode ray tube 27.
Supply to the shaft. Therefore, the frequency characteristics of the signal from the input terminal 11 are displayed on the cathode ray tube 27.

Now, if the delay time of the frequency divider can be ignored, the transfer function of the frequency divider with the frequency division ratio n is 1/n, so the transfer function of the circuit consisting of the frequency dividers 21 and 22 is 1/12. . Therefore, when the phase noise included in the signal from the high frequency oscillator 19 is ΔΘ, the phase noise included in the signal appearing at the output terminal of the frequency divider 22 becomes ΔΘ/12, which should be improved by about 22 dB.

However, in reality, there is a limit to improving the phase noise of the oscillator using this conventional method due to the phase noise of the frequency divider itself. For example, FIG. 2 shows the phase noise characteristics of the output of a frequency divider composed of flip-flops, where the horizontal axis represents the offset frequency m with respect to the oscillation frequency, and the vertical axis represents the ratio of the oscillation frequency signal to the noise. In Fig. 2, the phase noise near the oscillation frequency can be removed to some extent by configuring the high frequency oscillator 19 using a phase locking circuit and stabilizing its oscillation frequency, but the phase noise near the oscillation frequency can be removed to some extent. Since the gain of the synchronous circuit is lost, the phase noise is not improved, and in an actual circuit configuration, approximately -150 dBc/Hz is the phase noise of the frequency divider itself and cannot be removed.

<Object of the Invention> An object of the present invention is to provide a spectrum analyzer with low phase noise not only near the oscillation frequency but also far away from the oscillation frequency.

<Summary of the Invention> The present invention uses a phase-locked circuit to configure a frequency divider that divides an input frequency to 1/n, and a first local oscillator is provided by this frequency divider and a variable frequency oscillator. The oscillation signal from one local oscillator and the input signal are mixed to obtain the spectrum of the input signal.

<Embodiment of the invention> FIG. 3 shows an embodiment of the invention. Components in the figure that are the same as those in FIG. 1 are designated by the same reference numerals. A variable frequency generator 28 and a frequency divider 29 constitute a first local oscillator. The variable frequency generator 28 is, for example,
Generates frequencies from 2.4 to 4.4GHz. The frequency divider 29 is a frequency divider using a phase synchronization circuit, and divides the input frequency by, for example, 1/2 to generate a frequency of 1.2 to 2.2 GHz. In the mixer 13, the frequency from the frequency divider 29 and the input frequency ₀ of, for example, 10 kHz to 1 GHz are mixed and converted into an intermediate frequency ₁ of, for example, 1.2 GHz. Furthermore, in the mixer 15, this first intermediate frequency
₁ and the frequency from the second local oscillator 20 of, for example, 1.17 GHz, and convert it into a second intermediate frequency ₂ of, for example, 30 MHz. This is amplified by a logarithmic amplifier 17, detected by a wave detector 18, converted to a DC signal, and supplied to the Y-axis of the cathode ray tube 27 to display the frequency characteristics of the signal from the input terminal 11.

The frequency divider 29 includes a voltage controlled oscillator 31 and a multiplier 3.
4. Frequency conversion is performed using a phase synchronized circuit consisting of a phase comparator 33 and an adder 39.
Multiplier 34 receives frequency γ ₁ from isolator 32
is multiplied twice, and the phase comparator 33 compares the phase of this signal with the signal from the variable frequency generator 28. The frequency setter 38 is a pretuning voltage generator that receives the sweep signal from the sawtooth wave oscillator 24 and generates a voltage V _T corresponding to the oscillation frequency γ ₁ of the voltage controlled oscillator 31.
There is one that controls the voltage controlled oscillator 31 by adding V _T and the voltage from the phase comparator 33. Therefore, the oscillation frequency γ ₁ of the frequency divider 29 becomes γ ₁ =1/2γ ₀ .

Furthermore, the frequency divider 29 can also be configured as shown in FIG. This uses a circuit consisting of a frequency converter 35, an oscillator 36, and a phase comparator 37 instead of the circuit consisting of the transmitter 34 and phase comparator 33 shown in FIG. It is a device consisting of a mixer, or a multiplier and a mixer. This frequency converter 35 doubles the frequency γ ₁ of the isolator 32 and mixes this with the frequency γ ₀ from the input terminal 40,
An output with a frequency of γ ₀ −2γ ₁ is supplied to the phase comparator 37 . A phase comparator 37 compares the phases of the signal from the frequency converter 35 and the signal from the oscillator 36. The operations of the voltage controlled oscillator 31, frequency setter 38, and adder 39 are the same as in the case of FIG. Therefore, the oscillation frequency γ ₁ of the frequency divider 30 is γ ₁ =
It becomes 1/2 (γ ₀ − _S ). Here, the oscillation frequency _S of the oscillator 36 is set to be much smaller than γ ₀ .

Frequency setter 3 used for frequency dividers 29 and 30
8 may be constituted by a frequency/voltage converter, and may be configured to convert into the above-mentioned voltage V _T in accordance with the frequency from the variable frequency generator 28 or the input terminal 40. Furthermore, if the sweep frequency width is narrow, it may be set to a constant voltage. Further, a buffer amplifier, a coupler, or the like may be used instead of the isolator 32.

Next, another embodiment of the present invention is shown in FIG. This is a super-heterodyne type spectrum analyzer, and the variable frequency generator 28 is, for example, 2
A frequency of ~4 GHz is generated, and this frequency is sequentially divided by 1/2 through frequency dividers 29, 29', 29'', . . .
When the frequency is ₀ or 1 to 2 GHz, it is switched to the A terminal, and when ₀ is 0.5 to 1 GHz, it is switched to the B terminal, and the mixer 13 mixes the frequency from the input frequency ₀ selector switch 42, and passes the filter 14 to the frequency ₁ , for example MHz. Occur. This frequency is amplified by a logarithmic amplifier 17, detected by a wave detector 18, and converted into a DC signal.
7 and displays the frequency characteristics of the signal from the input terminal 11. Also frequency divider 2
9, 29', 29'', . . . may be replaced by a frequency divider 30 having the configuration shown in FIG.

<Operation of the Invention> FIG. 6 is a diagram for explaining the phase noise of the frequency divider according to the present invention, and parts corresponding to those in FIG. 4 are designated by the same reference numerals. Also mixer 43
and multiplier 44 correspond to frequency converter 35 in FIG. Here, ΔΘ _r1 is the phase noise caused by the oscillator 36, ΔΘ _r2 is the phase noise included in the signal from the input terminal 40, ΔΘ _V is the phase noise caused by the voltage controlled oscillator 31, ΔΘ _n is the phase noise caused by the multiplier 44,
It is assumed that V _od is the noise voltage caused by the phase comparator 37, V _oa is the noise voltage caused by the active loop filter, and V _ot is the noise voltage caused by the tuning port of the voltage controlled oscillator, which are respectively added to this circuit network through an adder. Now, let the transfer functions of the voltage controlled oscillator 31, multiplier 44, phase comparator 37, and adder 39 be Kv/S, n, Kφ, and A(S), respectively, and the open loop gain of this circuit is Go(S). Then, Go(S)=Kφ・A(S)・Kv/S・n. The phase noise appearing at the output terminal 41 due to each noise source is as follows.

ΔΘ _put,r1 = 1/n (1+1/Go(S))・ΔΘ _r1 ΔΘ _put,r2 = 1/n(1+1/Go(S))・ΔΘ _r2 ΔΘ _put,v =1/1+Go(S)・ΔΘ _v ΔΘ _put,n = 1/n (1+1/Go(S))・ΔΘ _n ΔΘ _put,od = 1/Kφ・n・(1+1/Go(S))・V
_od ΔΘ _put,oa = 1/Kφ・n・(1+1/Go(S))・V
_oa ΔΘ _put,ot = K _V /S·1/1+Go(S)·V _ot From this, the spectral density of the phase noise at the output terminal 41 is as follows.

SΔθ _put = (ΔΘ _put,r1 ) ² + (ΔΘ _put,r2 ) ² + (Δ
Θ _put,V ) ² + (ΔΘ _put,n ) ² + (ΔΘ _put,od ) ² + (Δ
Θ _put,oa ) ² + (ΔΘ _put,ot ) ² =1/n ² (1+1/Go(S)) {ΔΘ _r1 ² +ΔΘ _r2 ²
+ΔΘ _n ² }+1/Kφ ²・n ²・(1+1/Go(S)) ² {V
_ot ² +V _oa ² } + (1/1 + Go (S)) ² {ΔΘ _V ² + (K _V /S) ² V _o
t ² } In general, assuming that the phase noise ΔΘ _r2 included in the signal from the input terminal 40 and the phase noise ΔΘ _V caused by the voltage controlled oscillator 31 are large and other noises can be ignored, SΔθ _put = 1/n ² (1+1/ Go(S)) ²・ΔΘ _r2 ² +
(1/1+Go(S)) ²・ΔΘ _V ² = {1/n(1+Sτ)} ²・ΔΘ _r2 ² + {1/
1+1/Sτ} ²・ΔΘ _V ² ...(1) (However, τ=1/Kφ・A(S)・K(v)・n).

From equation (1), the range where the frequency deviation from the oscillation frequency is small, that is, the open loop gain Go
In a range where (S) is large, the influence of the phase noise of the signal from the input terminal 40 is large, and SΔθ _put {1/n(1+Sτ)} ² ·ΔΘ _r2 ² 1/n ² ·ΔΘ _r2 ² . Expressed in decibels, 10log(SΔθ _put ) = 10log(1/n ² ΔΘ _r2 ² ) = 10log(ΔΘ _r2 ² )−20logn｜dB In other words, phase noise near the oscillation frequency is included in the signal from the input terminal. 20logn phase noise
A signal improved by (dB) appears at the output terminal 41. Since n=2 now, the phase noise of the signal appearing at the output terminal is improved by 6 dB compared to the signal at the input terminal.

Furthermore, in a range where the frequency deviation from the oscillation frequency is large, that is, a range where the open loop gain Go(S) is small, the second term in equation (1) becomes dominant, and SΔθ _put {{1/1+1/Sτ} ² ΔΘ _v ² ΔΘ _v ² , ie the noise at frequencies far away from the oscillation frequency, is determined by the noise of the voltage controlled oscillator.

Figure 7 shows the phase noise characteristics of each device, and the horizontal axis is the offset frequency with respect to the oscillation frequency.
m, the vertical axis is the ratio between the oscillation frequency signal and the noise.
As the variable frequency generator 28, for example, a dashed line 61
As shown in FIG.
Use one with the phase noise characteristics shown in . The two-dot chain line 63 represents the phase noise characteristic of the voltage controlled oscillator 31 when the output of the voltage controlled oscillator 31 is doubled.
The output is 6dB worse. When the output of the variable frequency generator 28 is divided by the conventional frequency divider shown in FIG. 1, the phase noise of the variable frequency generator 28 is greater than the phase noise of the variable frequency generator 28 in the vicinity of the oscillation frequency, as shown by the dotted line 65, by the frequency division ratio. This is improved, but at a distance far from the oscillation frequency, it is determined by the phase noise of the frequency divider itself and tends to linger for a long time. By dividing the output of the variable frequency generator 28 by a frequency divider 29 or 30 including a phase synchronization circuit, the solid line 64
As shown in , the frequency division ratio is 1/2 near the oscillation frequency.
The noise characteristics far from the oscillation frequency are determined by the phase noise of the voltage controlled oscillator 31, compared to the phase noise characteristics 65 of a conventional frequency divider. Purity increases. Also variable frequency generator 2
By passing the output of 8 through a plurality of frequency dividers, for example, as shown in FIG. 5, the phase noise characteristics can be further improved.

Furthermore, by using the frequency divider according to the present invention, the frequency range from 10kHz to 1GHz is shown in FIG. 3, and from 0.25 to 2GHz by using three frequency dividers in FIG.
It becomes possible to measure the spectrum over a wide band. Further, by widening the oscillation frequency band of the variable frequency generator 28 to, for example, 4 to 8 GHz, it becomes possible to perform spectrum measurement in an even wider band.

<Effects of the Invention> As explained above, according to the present invention, the oscillation frequency from the variable frequency generator is frequency-converted using a frequency divider constituted by a phase-locked circuit, and the frequency is converted into the frequency of the input signal. By mixing the two to obtain the first intermediate frequency, it is possible to obtain a spectrum analyzer that can measure the phase noise of a spectrum with high sensitivity in a wide frequency range.

[Brief explanation of drawings]

FIG. 1 is a diagram showing a conventional spectrum analyzer, and FIG. 2 is a diagram for explaining the characteristics of a frequency divider used in a conventional spectrum analyzer.
FIG. 3 is a diagram showing one embodiment of the spectrum analyzer according to the invention, FIG. 4 is a diagram showing another embodiment of the frequency divider constituting the spectrum analyzer according to the invention, and FIG. 5 is a diagram showing another embodiment of the frequency divider constituting the spectrum analyzer according to the invention. Diagram showing another embodiment of the spectrum analyzer, No. 6
7 and 7 are diagrams for explaining the characteristics of the frequency divider constituting the spectrum analyzer according to the present invention. 11, 40: Input terminal, 41: Output terminal, 1
2: Attenuator, 13, 15, 43: Mixer,
14, 16: Filter, 17: Logarithmic amplifier, 1
8: Detector, 19: High frequency oscillator, 20: Second local oscillator, 23: Amplifier, 21, 22, 29, 3
0: Frequency divider, 24: Sawtooth wave oscillator, 25: Center frequency setter, 26, 39: Adder, 27: Cathode ray tube, 28: Variable frequency generator, 31: Voltage controlled oscillator, 32: Attenuator, 33 , 37: Phase comparator, 34, 44: Multiplier, 35: Frequency converter, 36: Oscillator, 38: Frequency setter, 42:
Changeover switch.

Claims (1)

[Claims] 1 A: a variable frequency generator; B: means for inputting a signal; b: a voltage controlled oscillator whose oscillation frequency is controlled in accordance with a control signal; c: multiplying the oscillation frequency of the voltage controlled oscillator. d) a phase comparator that compares the phases of the input signal and the multiplied frequency signal; e) a frequency setter that outputs a voltage corresponding to the oscillation frequency of the voltage controlled oscillator; and f) of the frequency setter. an adder that adds the voltage and the voltage of the phase comparator to output a control signal for the voltage controlled oscillator; g means for outputting a signal of the voltage controlled oscillator; and a frequency divider consisting of: A spectrum analyzer equipped with a first local oscillator.