JPS6361977A - Frequency modulation continuous wave radar system for measuring range - Google Patents

Abstract

(57) [Abstract] This bulletin contains application data before electronic filing, so abstract data is not recorded.

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a frequency modulated continuous wave type (FM/CW) radar system intended for range measurements, which comprises a linear frequency modulated microwave signal generator, a transmitting and receiving antenna, and a local oscillator. comprising derivation means for deriving a partial signal of the transmitted wave, called a signal, and a partial signal of the received echo wave, and for forwarding said two signals to the input of the mixer; The length of the transfer line is adjusted so that the interfering internally coupled signal has the same propagation time as the local oscillator signal to the mixer with the possibility of static adjustment of the phase between these two signals.

Generally, radars with a single antenna are used for long range measurements. In this case, the power is sent in a series of waves, which can be made independent of the radar's internal interference echoes. The present invention proposes to create a radar with a single antenna capable of measuring short ranges of the order of a few meters, and more specifically for board missiles (b
radio-altimetric probe that can be used for oardmissile
) and can measure ranges as small as 3 m.

The main disadvantages of radar or radio altimeter systems with two antennas (which are mentioned as technical problems in the background of the present invention) are as follows: the antenna cannot be integrated into the device; , equipped aircraft (
The problem is that these lines can be a source of error in range measurements, and must be connected by transmission lines that must be fixed on one side of the carrying aircraft and whose transmission delays must be calibrated. Moreover, the geometry of the antenna system is subject to imperfections when the distance separating the antennas is not negligible compared to the distance from the ground or the distance from the object to be detected, which is again a problem at small ranges. This results in a source of error. Finally, it is desirable to simplify antenna systems for purposes of compactness, installation on aircraft, and general economy.

The radar transmits asymmetric sawtooth waveforms with a constant positive slope for a given range, these sawtooths of period T are separated by (constant) level sections, i.e. these sawtooths have a constant positive slope of T T is greater than T no matter what
It is assumed hereafter that it is generated with a repetition period Tr which is r. Under these conditions, the general formula expressing the actual functioning mode of the radar is: ■ where f5 is the subtractive beat frequency between the transmitted wave and the received echo wave.
ncy), τ: Delay time between transmitted wave and received echo wave, ΔF: Transmitted signal or wobble frequency (wobul
frequency).

The actual range D is the following formula has a linear relationship with the delay τ.

By combining equations (1) and (2), the following basic equation is obtained.

Here, C is the speed of electromagnetic waves in the air.

The useful signal is created by a mixer that generates a subtractive beat between the local oscillators sampled from the transmitted and received waves.

If V. If L is the voltage of the local oscillator and VR is the voltage of the received wave, the beat output voltage V of the mixer is Vb − to −Vot・VR・s+n(2yr f B
t+ψ) (4), where ψ=2π・τ・Fa.

And F mi. is the minimum frequency of the transmitter and K is the mixer conversion coefficient. In the case of sawtooth modulation, the signal commonly used in radio altimeters is a sine wave train (sin wave train).
(usoidal train) whose periods T are sawtooth wave periods separated by (-4) level sections T, -T. In each level section, the mixer beat frequency V, is V,: K-VOL-
It is fixed at VR-s+n r (5).

It will be noted that the period of the beat signal corresponds to the wobble repetition period TR. Its frequency spectrum is therefore composed of frequency harmonics.

The spectrum of the beat signal can be calculated from its temporal shape by Fourier transformation. It represents the power density at the harmonics of the wobble frequency f. Starting from this calculation, by varying the various parameters, we can modify their evolution, especially in short ranges.
), it is possible to analyze the spectrum. Among several possible choices of radar functionality, the beat frequency of the signal f5 and the fixed tuning frequency f of the receiver.

It is preferable to choose to adjust it to . For calculation purposes, fbo is given as 25 kHz, for example. Analysis of the spectrum allows the following phenomena to be observed.

The number of sine wave periods n according to the above equation [4] between wobbles
When is a large number on the order of 10 or more, the envelope of the spectrum is distorted by the frequency fbo, where n = f, ·T (6) or n = τ ·AF (7), and the maximum value shifts towards lower frequencies. As a first approximation, the relative error of measurement in range D is equal to . F is the maximum frequency of the spectrum.

In order to keep the relative error of D within 10%, the number n needs to be greater than 3. In other words, this does not apply (Equation (7)
)), that is, if we combine equations (2) and (7),
Obstacle or target minimum range Dmi. This means that it must be equal to .

Apart from the above-mentioned drawbacks of two-antenna FM/CW radars, on the other hand, two-antenna radars easily satisfy the accuracy and sensitivity requirements imposed in the range 4.2 to 4.4 GHz reserved for radio altimeters. has been proven. In particular, a delay line placed in either the transmit channel or the receive channel solves the problem of accuracy at low altitudes because it artificially increases the range of the target. In this case, the use of an isolator reduces the internal B leakage without attenuating the dominant signal and allows measurements in the range from 0. With a single antenna, it is more difficult to solve the problem of self-dazzle noise. Over the same channel, the radar receives waves reflected by the antenna and useful waves coming from the target as echoes. Therefore, the use of isolators and delay lines is not possible without attenuating useful signals.

Therefore, detection of close targets with single antenna radar is an inherently very difficult problem to solve. As shown, applying equation (9) above yields 150 M
A frequency shift of Hz limits radar measurements to ranges greater than 3 m. To further reduce the minimum range, an attempt is made to increase the value of 75F, but the following two difficulties arise. That is, on the - side, it is difficult to obtain a frequency-linear source over a wide frequency deviation greater than 15QMHz. On the other hand, and this is a fundamental problem, the phase of the antenna's reflection coefficient is not linear over a wide passband. But to put it simply, 30
0M) obtained a reasonable linearity of the order of 10% over a deviation of 7JF, and with some types of antennas 40
It would seem possible to obtain good antenna matching over 0 MHz. So the minimum range can be lowered, at least theoretically to 1.5m with 300MHz AF.

As a result of the quasi-theoretical limitations mentioned above, several techniques are conceivable for creating single antenna radars.

For short range measurements, a known technique of radar known as "pseudo FM/CI'lJ" chops the transmitted signal according to the sampling theory and whose duration is, for example, 6.5 m for the range to be measured. It consists of transmitting a wave train that must be very short, less than 6 ns.It is difficult to create a switch for such fast microwave frequency signals, which is a challenge for this technology. This is a technical limitation.An FM/CW radio altimeter has already been proposed in which the transmitter and receiver function continuously using a common antenna.In this radio altimeter, the local signal of the receiver's radio frequency mixer is antenna S, W, R.

(Standing Wave Regillo)

This configuration is thus considerably simplified, but at the expense of a minimum measurable altitude that can reach 6 to 10 m, i.e. a value that is too high for the required application of the radar according to the invention. It is. Single-antenna FM/CI'l radars intended especially for short range measurements are also well known, in particular from French Patent No. 2.541465, which poses exactly the same technical problems as in this description. It's happening. In an FM/CWb-Goo device, if the transmitted and reflected signals are decoded on a common antenna, an interference signal is generated due to the coupling between the transmitter and receiver. This complex coupled signal is mainly due to partial reflections at the antenna, which has a finite and non-O magnitude reflection coefficient, and its directivity coefficient.
tof directivity) is the leakage signal introduced by the duplexer circuit, which is technically limited. This combined radio frequency signal, after demodulation in the input mixer of the receiver, creates an interfering signal at the input of the audio frequency amplifier of the receiver, which is located partially in the passband of the echo signal. , and has a significant amplitude. This limits the sensitivity of nearby object detection. To overcome this lack of sensitivity, it is known that static adjustments can be made, which consist of adjusting the length of the lines and possibly adjusting the phase shifter placed on one of these lines. Therefore, the phase of the local oscillation matches the phase of the main interference signal due to S, W, and R of the antenna. However, static adjustment is 1.
This is itself insufficient to achieve the desired accuracy for range measurements on the order of 5 m to 3 m.

In the above-mentioned French patent, in order to improve the sensitivity of the radar, an additional measurement is carried out after the input mixer of the receiving channel at a second channel connected by a level modulator to the frequency modulated signal generator of the transmitter. It consists of the introduction of an audio frequency amplifier with two inputs, the level modulator having a control input sensitive to an adjustable DC voltage signal.

According to the present invention, the above technical problem is solved in the FM/Cl1l radar specified in the first paragraph, in which a phase control loop including an amplifier and a low-pass filter is placed between the output of the above mixer and the local oscillator channel. the closed-loop cutoff frequency of said control system is just below the frequency of the useful subtraction beat signal between the transmitted and received waves provided at the output of the mixer; are solved separately by facts characterized by .

Carefully designed and arranged, such a control loop compensates in an instantaneous manner the variable phase difference that necessarily exists between some interfering signal, such as a mixer leakage signal, and the local oscillator signal. ing.

The following description, with reference to the accompanying drawings and given by way of example, will give a good understanding of how the invention may be embodied.

As we have seen above, it is not possible to introduce a delay line in a single antenna radar, which creates problems in the detection of close targets as the transit time of the reflected wave becomes very small. This problem, which is theoretically difficult, is illustrated by FIGS. 1 and 2, where the wobble frequency 71F is chosen equal to 150 MHz. In FIG. 1, the range D is made equal to 3 m (f b
= 25 kHz), the number n of sinusoids of the signal vb at the output of the mixer during the period T is equal to 3 in accordance with equation (7) above (see FIG. 1a). The maximum of this spectrum is obtained very close to the beat frequency f of 25 kHz, with the x shown in FIG. 1b. It will be noted that this is true for n greater than 3, i.e. for measurable ranges D greater than 3·m in the numerical application chosen as an example.

On the other hand, for nl smaller than 3 or D smaller than 3m, the spectrum is distorted and the measurement of D becomes inaccurate or impossible. In Figure 2, range D=
2m.

fb is equal to 25kHz and n=2.

not similar to If a section of the envelope surrounding a frequency f of 25 kHz is taken, the local maximum of the envelope is no longer obtained at a frequency of 25 kHz, but 22 kHz
, which would result in a range D error of 12% according to equation (8), assuming this maximum value is detected. For less than 2 sine waves, the spectral envelope is further distorted until it becomes a monotonically decreasing curve in the vicinity of D=1 m, after which the measurement of D has no further physical meaning. It will be noted that the value of T has almost no effect on the shape of the spectrum of signal v5.

In addition to the above-mentioned theoretical limitations, another difficulty in creating a single-antenna laser arises from the magnitude of the interfering signal, the total level of which is suitable to be below the level of the useful signal V,. A study of these interfering signals and their processing will now be explained with reference to the radar configuration of FIG. The portion of FIG. 3 represents a standard radar system for range measurements, including a transmission channel constituted by a voltage controlled oscillator (VCO) 1 connected to an antenna 2 by a circulator 3. VC low has a microwave frequency generator,
For example, this is 4225! AHz and 4375 MHz
between, i.e. at a center frequency of 4.3 GHz,
A positive asymmetric sawtooth waveform signal is generated which is linear in frequency for a period T in which two continuous sawtooth waves are separated by a sawtooth wave line -T. The receiving channel includes antenna 2, a transmitting and receiving antenna designed for this purpose to pick-amplify echo signals from the ground or target;
constituted by a circulator 3, one output of which is connected to a mixer 5 by a line 4,
The part of the transmitted wave emitted by is sampled by the amplifier 6 and transmitted directly to the second input of the mixer 5 by a line known as the local oscillator line 7 in standard radars. The mixer 5 is preferably a symmetrical mixer of the 4λ/4 type and its output, which is double output, is transmitted by two conductors 8, 9 to the negative and positive inputs of the differential amplifier 11. Output 1 of amplifier 11
2 is a source of an audio frequency signal S, which inter alia includes:
Frequency f, which is the subtracted beat frequency between the transmitted wave and the received wave
5 useful signals Vb. Output 12 is processing circuit 1
3, which performs range D measurements. Known means of modulating vCOI do not assist in understanding the present invention.
l and not shown. All that has been explained is that the frequency modulation must be obtained with good linearity.

Without special precautions, using the configuration described above, noise will be dominant in the signal S, especially at short ranges, and it will be impossible to measure range D for small ranges. In the radar shown in FIG. 3, the noise is due to interference coupling, and the following two are the main ones. That is,
The first, whose path is represented by dashed line 14, is caused by antenna reflection. -1QdB of antenna
Considering the reflection coefficient of , this interference coupling is -10dB-2
Equivalent to a transmitter/receiver coupling of 0βOgD. The second interference coupling results from leakage in the circulator 3. The equivalent coupling is equal to -25 dB - 20 βog D. The third one arises from leakage from mixer 5 by an equivalent coupling of -50 dB - 20 log D. Together these three interference combinations constitute what is now called the radar's self-dazzling noise. Each interferometric coupling produces a beat frequency whose path is due to a local oscillator signal represented by dash-dotted line 15 in FIG. 1, which signal has an instantaneous phase as follows.

5, where U is the reference index of the associated interferometric coupling signal, ALu represents the electrical length difference between the local oscillator and the associated interferometric coupling, and F is the instantaneous frequency of the transmitted signal.

The main difficulty in extracting a useful signal from the noise caused by the interference coupling at the output of the mixer 5 is the fact that the main interference coupling signals have mutually different electrical lengths, especially with respect to local oscillations. It is. A first means of implementing the invention is to correctly identify the main interference coupling path and to perform static correction adjustments.
the electrical length of these paths in order to equalize the paths of the local oscillator signals and to bring these interfering signals in phase with the local oscillator signals. However, this adjustment can only be performed for waves whose path is independent from that of the local oscillator. Therefore, the aforementioned third interference signal resulting from mixer leakage cannot be compensated for, although this latter signal fortunately has the lowest level of the three signals. Reference symbol β for local oscillation
. 1, and the line length adjustment that is FCP for statically compensable interference coupling naturally takes into account the different microwave frequency components encountered in the different paths, each of these components having a measurable electrical length. , and added to the actual transmission line length along the associated path. In this way, the level of the antenna reflected signal is set to -35clB-20A.
og D, and the circulator leakage signal to the value -5
It is possible to drop to 0 dB −20 βog D.

Figure 4 shows the Fresnel diadem.
agram) illustrates the various interferences between the wobbles and the evolution of the useful beat signal in the form of a vector that can be moved more or less with respect to the local oscillation vector 16 taken as a reference vector. Starting from the top of the fixed vector 16, a vector 17 is formed representing the antenna reflection and circulator leakage. Static adjustment causes vector 17 to remain fixed relative to vector 16 during wobble, ie the resultant vector 18 of these two vectors also remains fixed. It is also preferable to try to align the two vectors 16 and 17. To do this in addition to static adjustment, an adjustable phase shifter can be introduced in the receive channel 4 or in the local oscillator channel 7. This phase shifter is represented by the dashed line at 20 in FIG. 3 in the local oscillator channel, and it can be created by a fixed and adjustable capacitor. It is necessary to consider the equivalent electrical length of the phase shifter 20 during static adjustment.

Starting from the top of vectors 17 and 18 in FIG. 4, other vectors such as 19 and 21 are shown symbolizing other interferingly coupled signals that cannot be phased with respect to the local oscillator signal, i.e. In particular, the lead or lag with respect to a local oscillation signal, such as a leakage signal from the mixer 5, is symbolized. During the wobble, the vertex of the vector 19 advancing with respect to the local oscillation traces a curve 23, while
The vertex 22 of the lagging vector 21 describes a curve 24. Point 22 represents the end of the resultant vector, from which a vector representing the useful signal 25 is drawn, which has a lower amplitude with respect to the previous vector. During the wobble,
Vector 25 with peak 26 describes several rotations around point 22. The projection of point 260 onto axis 27 perpendicular to vector 16 provides the variation of the sought voltage V,. The motion of points 22 and 26 can even be better represented by introducing a time axis aligned on vector 16.

In the plane of coordinate axes 27 and 28, points 22 and 26 are curved by solid line 29 and dotted line 31, respectively.

Without interference coupling, the curve 31 would be a sine wave along the time axis Ot. Attenuating the disturbing effects of interferential coupling is equivalent to approaching this theoretical sine wave as closely as possible. To do this, we keep the resultant vector 18 fixed and 5
and aligned on vector 16 as described above, but other measures are needed for vectors such as 19 and 21. Reducing the amplitude of the interference vector is a preferred element for implementing the invention. In fact, the level of the useful signal represented by vector 25 in FIG.

It can be calculated from the received power P.

at the knee G: Antenna gain σ0: Reflectance of the ground /l: Transmission wavelength It is.

A useful signal is typically at a level of -60 dB-20 βog D. If a comparison is made with the levels already mentioned above for the main interfering signal, it is possible to improve the signal-to-noise ratio S/N from -50 dB to -25 dB after static adjustment, but this It has been noted that this is insufficient and that achieving a S/N ratio of >+10 dB is adequate. As shown, the spectral envelope of the interfering signal after static adjustment is represented by the dotted line 33 in FIG.
If, then, only the spectral lines adjacent to the beat frequency f, are considered, the S/N ratio is approximately OdB,
This is a desirable property if there is appropriate filtering of the signal.

The invention also results from introducing a dynamitter reduction of the interference coupling in the form of a phase control between the outputs 8, 9 of the mixer 5 and the phase shifter 35, which is preferably inserted into the local oscillator channel. Resolved technical issues. It is a problem of attenuating the effects of interfering signals without interfering with useful signals.

Let 15 be the equivalent electrical length of the useful coupling produced by the reflection of the wave on the ground. The phase shift depends on the mixer type to the value λ/2 or to the value 0, and the phase shift introduced by the useful signal is not modified, ie it becomes.

For this purpose, after amplification and filtering, the audio signal at the output of mixer 5 controls a phase shifter 35 .

For example, the dual outputs 8 and 9 of the mixer 5 are the two outputs of the operational amplifier 36.
one input, and its output is a low-pass filter 37
connected to. This control makes it possible to obtain a sensitivity in which the sensitivity of the mixer, which depends on the beat frequency of the audio signal, is improved especially at low values of f. Low pass filter 37
The filtering performed in 1 consists only of passing useful signals at frequencies lower than the beat frequency fb. In particular, the frequency,f,is a fixed value,f,. (example
(for example, 25 kHz), the cutoff frequency fc of the low-pass filter 37 is f. In this way, quasi-instantaneous compensation
tantaneous compensat 1on)
has been obtained for the jamming effects of interfering signals that cannot be compensated for statically. Referring to FIG. 4, where the instantaneous combination of the just-mentioned interference signals is defined as 38, everything is as if the resultant vector 18 was always equal to the vector 38.
This will greatly improve the signal-to-noise ratio.

In Figure 4, this means that the origin of vector 18 is from 0' to 0.
′ is 5. For dynamic compensation,
The S/N ratio can be increased to values on the order of 10 dB as desired. Amplifier 36 in FIG. 3 may be a simple operational amplifier that also includes the filtering function of low pass filter 37. This operational amplifier has, for example, an open loop, a cutoff frequency of 10 Hz, and a gain of 1001 B. mixer 5
Taking into account the conversion factor and phase shifter 35, the closed loop cutoff frequency is 20 kHz. This filter is a single pole filter, giving a gain characteristic of 201B/octave,
This gives a control loop with good stability. Such control does not affect the useful signals, the loop is closed only for interfering signals, and as the beat frequencies of the interfering signals become lower, i.e. the beat frequency at which the level of these interfering signals is highest It has a large effect on the value of . Since the phase shifter 35 can compensate for the residual delay of the internal echoes, the phase shift α between the combination of the interfering voltage signal (i.e., vector 46 in FIG. 4) and the local oscillator signal is the field of action of the phase shifter. )
It should be noted that it is necessary to remain within. This restriction is basically satisfied for the static adjustment described above.

Other weaker interference waves generally cannot create large phase shifts among the total interference waves. In practice, phase shifter 35 can be made by a single diode, and the phase change of vector 46 is of the order of π/2. If this phase change has the value π/
If more than 2 is required, it is necessary to use a phase shifter that can operate for such increased phase changes.

The beat frequency f of the useful signal is set to a fixed value fbo 1
It will also be noted that it should not be maintained.
``Some altimeters have frequency shift AF and wobble run.
The slope p of the eve p is a constant pitch. these conditions
1 Teha' = "-to frequency f b is the range D to be measured
+,: Directly proportional. According to the above criteria for the operation of the filter 37, the cutoff frequency f of the filter
It is necessary to adjust each frequency c to one frequency f. In practice, such control of the filter 37 is carried out by the automatic search unit of FIG.
t) 4Q, which can be performed by a systematic test of increasing or decreasing the value of the slope p, in particular to lock on to the range D searched during the switch-on of the radio altimeter.
(FIG. 5 shows the operation of the radar according to the invention. Four curves are shown, which represent the level in dB as a function of the range D on a logarithmic scale.

The dotted curve 41 represents the self-dazzling noise of the radar with only static compensation (adjustment), the dotted curve 42 is the self-dazzling noise curve awaiting static and dynamic compensation, and the dashed-dashed curve [4 represents the radar self-dazzling noise] represents the thermal noise level of The solid curve 43 between curves 41 and 42 represents the lowest useful word level. The signal-to-noise ratio S/N, which is the difference in vertical coordinates between curves 42 and 43, increases by 3 as D changes from 100 m to 3 m.
0 (changes from iB to 10dB.

FIG. 6 shows a second embodiment of the invention, in which a 3 dB coupler 48 is used. Surface elements that are the same as in FIG. 3 have the same reference symbols. 3dB coupler 48 is the third
Circulator 3 and coupler 6 in the figure: Thus, the given functions are integrated. The sixth diagram includes three isolators: one isolator 49 in the transmit channel and one isolator 51 in the local oscillator channel.
, and one isolator 52 in the receive channel of the useful signal. It will be noted that these isolators are particularly useful for improving sensitivity in high ranges, but are not essential for low ranges. Again an optional adjustable phase shifter 20 is shown here in the receive channel. Local oscillation channel 7 is 3dB
Between the coupler 48 and the mixer 5, preferably a symmetrical 4λ/4 type mixer, a 12 dB coupler 53 serving as an attenuator and a phase shifter 54 for controlling the phase of the local oscillation constituted by a varicap diode; and preferably includes an isolake 51. The control loop includes an operational amplifier 55 with microwave frequency decoupling constituted by a capacitor 56 and an inductance 57.
Contains.

In the radar configuration of FIG. 6, additional interference coupling must be considered. That is, it is a reflection of the transmitted signal on the 12 dB coupler 53, whose path is defined as 58. This interfering signal can be compensated for statically since its path is essentially independent of the local oscillator channel. In FIG. 4, it is represented by the dotted vector 59, and the reflected interference coupling of the antenna is represented by the dotted vector 17'.
It is represented by For example, the main coupling of the radar in FIG. 6 is as follows.

21 dB for antenna reflection two reflection coefficient -15 dB.

The reflection coefficient of the 12 dB coupler is -21 dB for the reflection coefficient of -15 dB, and the leakage of the 3 dB coupler 48 is -25 de.

Mixer leakage and receive isolator reflections: 45 dB0 Static reduction consists of line length adjustment proceeded by two dominant -21 dB couplings.

It is preferred that the antenna 2 is placed very close to the 3 dB coupler 48 and that it includes a matching, ie a line placed in front of the antenna, allowing equalization of the delays of these two reflections. The lengths of the two interference paths 14 and 58 are then adjusted to the local oscillation path (path 15). If the phase shift of the resultant vector 17 with respect to the vector 16 is too high at this stage of static adjustment, a phase shifter 20 is introduced, its equidistant length is measured and the length of lines 4 and/or 7 is accordingly corrected and then adjusted so that the relative phase between the two signals at the input of mixer 5 is brought to a value of the order of 5 to 10 degrees. After these adjustments, dynamic compensation can be performed in a manner similar to that described above with reference to FIG. After dynamic compensation, in both cases the level of the interfering signal is reduced to the value -55 dB - 401 OgD, which produces a signal-to-noise ratio S / N = 6 dB + 20 βog D. The dominance of the local oscillator over the combination of interference vectors gives a fairly constant power to the mixer 5 diode, so broadband matching of the mixer, such as a 10 dB variation in the antenna reflection, will therefore result in only a 3 dB variation in the total power. .

Preferably, a paricamp diode 54 is placed in series with the local oscillator transmission line where variations in capacitance result in large phase shifts. This position can be defined by the phase of the reflection coefficient of the isolator 51. The phase shifter therefore operates at approximately 90°, which allows compensation of large external echoes such as those caused by water droplets on the antenna. As for the operational amplifier 55, it also has the role of a filter. Its gain is 100 dB at 10 Hz in open loop and decreases by 20 dB/octave at higher frequencies.

For the radars of FIGS. 3 and 6, a very compact microwave frequency unit is obtained in short lines through the use of printed circuit technology. Once printed circuit cards are obtained after adjusting the lengths of the lines 4, 7 as described above, it is possible to mass-produce them with good accuracy.

The antenna can also be a printed antenna, such as antenna type 122 from the French company T, RoT.

For example, considering the expansion of Teflon glass, the electrical length of the IJ tube wire is 50Ω micros (50Ω micros) in the temperature range from 50℃ to 75℃.
Stable at a minimum of 1% over °C. If this uncertainty is considered over a 50 cm local oscillator line and assuming stable antenna reflections, the corresponding change in electrical length is 5 ma+, which is only 15° for a transmit frequency of 4.3 GHz. This results in a phase shift.

The present invention is a missile-like "intelligence".
It is preferred for applications in NTSC weapons where little space is available and the use of a single antenna instead of two antennas is desirable. For this application, the device according to the invention is preferably used in a radio altimeter capable of measuring altitudes between 3 m and about 1000 m.

This device can also be used as a radar. This is because the missile has a single antenna that makes it easy to spot and detect. In this latter case, the received power is derived from the following equation:

In this 5, δ is equivalent radar area (equivalent rad
area).

The radar according to the invention can detect targets with an equivalent area of 1 m2, but up to a maximum range of 100 m, beyond which thermal noise predominates over self-dazzling noise. To measure higher ranges, a high gain antenna or high transmit power is required.

For example, by increasing the value of the wobble frequency AF to 300 MHz, and for example its reflection coefficient is 400 MHz.
T, R, T with good matching over the range.

By choosing an antenna that has linear phase over a wide passband, such as our 140 type antenna,
It is possible to extend the range of the radar according to the present invention to a range of 3 meters or less.

(Summary) This radar is used to sample a local oscillator signal and an echo signal and to send these two signals to the input of a mixer (5) which gives at its output a subtracted beat signal between these two signals. includes a linear frequency modulated microfrequency signal generator (1), a transceiver antenna (2) and deflection means (3, 6; 48). The line length is adjusted to create a static adjustment of phase between the local oscillator signal and the interfering antenna reflected signal. According to the invention, an amplifier (36) and a low pass filter (37)
A phase control loop comprising a phase control loop is connected between the output of the mixer and the input of a phase shift circuit (35; 54) placed in the local transmission channel. The closed control loop cutoff frequency is just below the frequency of the subtracted beat signal.

F! , +/CW radio altimeter.

[Brief explanation of the drawing]

FIG. 1a shows the waveform of the useful output signal of the mixer shown in FIGS. 3 and 6, and the associated spectrum is shown in the same figure. Figure 2 shows the first range for the lower range D of the radar system.
A similar curve is shown in the figure. FIG. 3 is a block diagram of a first embodiment of a radar system according to the invention. FIG. 4 is a Fresnel diagram which allows the operation of the radar to be explained. FIG. 5 is a diagram showing the behavior of the radar as a function of range D. FIG. 6 is a block diagram of a second embodiment of a radar system according to the invention. 1... Voltage controlled oscillator (VCO) 2... Transmitting/receiving antenna 3... Circulator 4
...Line 5...Mixer 6...Coupler 7...Local oscillation line 8.9...Conductor or output 11...Differential amplifier 12...Output 13...
・Processing circuit 14...Dotted line or interference path 15...-Dot-dashed line or path 16...Local oscillation vector or fixed vector 17
, 17'... Vector 18... Composite vector 1
9, 21... Vector 20... Phase shifter 22.
...Vector vertex 23.24...Curve 25...
・Useful signal or vector 26... Point or peak 27.28... Coordinate axis 29... Solid line 31... Dotted line 33...
・Dotted line 35... Phase shifter 36... Operational amplifier 37... Low pass filter 38...
Interfering signal or vector 40... automatic search unit y) 41.42.43.
44...Curve 46...Vector 48.
...3dB coupler 49, 51. 52... Isolator 53... 12dB coupler 54... Phase shifter 55... Operational amplifier 56... Capacitor 57... Inductance 58... Interference path 5
9... Vector patent applicant NV Philips Fluiranpenfabriken

Claims (1)

[Claims] 1. Frequency modulation continuous wave type (F
M/CW) radar system, which includes a linear frequency modulation microwave signal generator, a transmitting/receiving antenna, a partial signal of the transmitted wave called a "local oscillation signal", and a partial signal of the received echo wave. , and derivation means for forwarding said derived signal to the input of a mixer, wherein at least the internal interference coupled signal due to antenna reflections is localized to the mixer by the possibility of static adjustment of the phase between these two signals. In those where the length of the transfer line is adjusted to have the same propagation time as the oscillating signal, a phase control loop containing an amplifier and a low-pass filter is placed in phase with the output of the above mixer and the local oscillator channel. connected between the control input of the shift circuit and characterized in that the closed-loop cutoff frequency of said control system is just below the frequency of the useful subtraction beat signal between the transmitted and received waves applied to the output of the mixer. radar system. 2. The radar system according to claim 1, wherein said deriving means is constituted by a coupler for local oscillation and by a circulator for deriving received echo waves. 3. The radar system according to claim 1, wherein the derivation means is constituted by a 3 dB coupler and the local oscillation channel includes a signal attenuation coupler, wherein the static adjustment is performed on the transmission signal on the attenuation coupler. A radar system characterized in that it responds to interference coupling caused by reflection. 4. Radar according to any one of claims 1 to 3, comprising an adjustable phase shifter in one of the two signal channels connecting said derivation means to said mixer. system. 5. Any one of claims 1 to 4, wherein the mixer is a symmetrical 4λ/4 type mixer having two outputs, and the phase shift circuit is constituted by a varicap diode. Radar system described in. 6. The radar system according to any one of claims 1 to 5, wherein the amplifier and the low-pass filter are both constituted by operational amplifiers. 7. comprising automatic means for adjusting said cutoff frequency f_c as a function of the range D to be measured, which controls said low-pass filter; The radar system described in one. 8. Claim 8, characterized in that said automatic means for adjustment are themselves controlled by an automatic search device intended to achieve the locking of said radar system in at least one search range D. Radar system according to item 7. 9. Claim 1, characterized in that the line transmitting microwave frequencies is created using printed circuit technology.
Radar system according to items 8 to 8.