JPS63159779A - Ultrasonic position detector - Google Patents

Abstract

PURPOSE:To improve distance resolution and to improve direction measurement accuracy by compressing pulses of a chirp signal made into a complex number by the best filter. CONSTITUTION:Wave receivers 2a and 2b are arranged nearby so that the phase difference at the center frequency of a received arrival wave is <=180 deg.. Further, frequency mixers 4a1 and 4a2, and 4b1 and 4b2, a local oscillator 5, a 90 deg. phase shifter 6, and LPFs 7a1 and 7a2, and 7b1 and 7b2, a 90 deg. phase shifter 6, and two orthogonal detecting circuits composed of LPFs 7a1 and 7a2, and 7b1 and 7b2 make chirp signals received by the receivers 2a and 2b into complex numbers respectively. Those chirp signals are compressed in terms of pulse by optimum filters 8a and 8b. The complex number output signals which are compressed in terms of pulse by the optimum filters 8a and 8b are used to measure the phase difference based upon the arrangement of the receivers 2a and 2b to find the ultrasonic arrival direction and find signal power from pulse-compressed complex number output signals, thereby detecting the position of an ultrasonic sound source.

Description

Detailed Description of the Invention [Industrial Field of Application] The present invention relates to an ultrasonic position detection device used for example on a ship, and more specifically, to an ultrasonic position detection device such as a transponder by measuring the phase difference of an ultrasonic reception signal. The present invention relates to a position detection device that determines the direction of a sound source.

[Conventional technology]

In the conventional ultrasonic position detection device called the ultra-short baseline force method, which measures the phase difference between adjacent receivers, it is necessary to measure the phase difference of the ultrasonic burst signal emitted from the transponder within the pulse time. be. Therefore, in order to obtain a highly accurate phase difference, it is necessary to increase the pulse time of the burst signal and perform sufficient averaging processing. As a result, the distance resolution deteriorates, making it difficult to accurately locate the transponder.

[Problem that the invention seeks to solve]

The present invention aims to improve the above-mentioned drawbacks of conventional ultrasonic position detection devices, improve distance resolution and direction measurement accuracy, and detect the position of an ultrasonic sound source such as a transponder with high precision.

[Means for solving problems]

An ultrasonic position detection device of the present invention for solving the above problems will be explained with reference to FIG. 1 corresponding to an embodiment. As shown in the figure, the ultrasonic position detection device of the present invention has a first receiving wave 112a and a second receiving wave that are arranged closely so that the phase difference in the center frequency of the received arriving wave is within 180 degrees. A wave receiver 2b, a first wave receiver 2a, and a second wave receiver 2
First orthogonal detection circuits 4aI, 4a2, 5, 6, 7a, 4aI, 4a2, 5, 6, 7a, .
7a2 and the second quadrature detection circuit 4b, 4b2, 5,
6Φ7bl·7b2, and a first optimal filter 8a and a second optimal filter 8b that pulse-compress the complex chirp signal.

[Effect]

From the complex number output signal pulse-compressed by the optimal filter 8a and the optimal filter 8b, the receiver 2a and the receiver ? The phase difference based on the arrangement of b can be measured. The direction of arrival of the ultrasonic wave is determined from the phase difference, and at the same time, the position of the ultrasonic sound source is detected by determining the seven-point power from the pulse-compressed complex output signal.

〔Example〕

Examples of the present invention will be described in detail below.

FIG. 1 is a block diagram of an embodiment of an ultrasonic position detection device to which the present invention is applied, which is installed on a ship. In the figure, 1 is the bottom of the ship, 2a and 2b are receivers, 3a and 3b are receiving amplifiers, 4a1, 4a2, 4b, and 4b2 are frequency mixers, 5 is a local oscillator, and 6 is a 90 degree (tc/2) shifter. Phase box, 7a1, 7a2, 7b1 and 7b2 are low pass filters, 8a and 8b are optimal filters, 9a1.
9a2, 9b, and 9b2 are multipliers, lOa and 10b are adders, 11 is a JAN-1 function calculator, 12 is a direction calculation circuit, 13a and 13b are squarers, 14 is an adder, 15 is a square root calculator, 16 is a display circuit.

Receivers 2a and 2b are installed on the bottom l of the ship with an interval d, and a linearly frequency-modulated chirp signal arrives from the direction of the arrow in the figure at an incident angle α as a response signal from a transponder, etc. . Signal S received by receiver 2a
The signal Se received by A and the receiver 2b is as follows, taking into account the phase delay and lead.

SA =A cos(2πfot+w gt2+θ)
--------(1) SB = A cos(2wfot
+w gt2+θ+ξ)-(2) where A is the amplitude,
rO is the center frequency, and the end is the frequency sweep rate (Δf/T),
Δf: frequency width, T is pulse width, θ is fixed phase, and 4 phase differences.

The phase difference ξ is based on the ultrasonic arrival direction α and the receiver spacing d, and if input is the wavelength at the center frequency to, the following relationship is established.

dsinα ξ=2π□ ・・・・・・(3)λ Therefore, the direction of arrival of the ultrasonic wave α is the modified input α=sin−1 of the first-order equation (3) if the phase difference between the signal S^ and the signal SB is obtained. (□ξ) ・・・・・・
・(02π d can be obtained. However, in order to eliminate ambiguity in phase difference detection, it is necessary to satisfy the phase difference condition 1ξ1−π. For this, the propagation distance difference d sin α must satisfy the following condition. There must be.

Input1dsinα1ku−・・・・・・(
5) From this, it can be seen that the spacing between the receivers becomes less than half a wavelength, making it possible to downsize the receiver.

Next, the signal S^ from the receiver 2a expressed by equation (1) is amplified by the reception amplifier 3a, and converted into a complex signal by known orthogonal detection. First, the 1 gin 2πfat signal generated by the local oscillator 5 and the input signal are sent to the frequency mixer 4.
a2 and pass it through a low-pass filter 7a2 to obtain the imaginary component QRA, and on the other hand, the output cos2πfat signal of the 90 degree phase shifter 6 and the input signal are sent to the frequency mixer 4.
By multiplying by a1 and passing through a low-pass filter 7al, a real component IR8 is obtained. As a result, the orthogonal detection output is as follows.

Real component: IRA=cos(πgt2+(?) −
... m (13) imaginary component = QRA = 5in (πJL
t2+θ) -------(7) For example, the imaginary component QR^ of this orthogonal detection output has a waveform as shown in FIG. be done.

The internal configuration of the optimal filter 8a is as shown in FIG. 20+ and 202 in the figure are a type of delay memory generally called DELTIG; 21.・21212・21
3 and 214 are multipliers, 221 and 222 are adders, and 231
・232 is an integrator. The operation of the optimal filter is equivalent to performing cross-correlation processing and provides the maximum signal-to-noise ratio at its output. Optimal filter 8b
has a similar configuration.

The input signal xR of the optimal filter is considered as a complex number I R+ j QR, and the cross-correlation function R(τ) with the reference signal Xr=Ir+JQr is determined by the following equation.

x■° indicates the conjugate complex number of Xr, and indicates the delay time.

Now, if we expand equation (8), we get becomes.

The configuration of the optimal filter shown in FIG. 3 executes equation (I3). The DELTIGs 201 and 202 temporarily store the input signals, and the multipliers 21 and 212 and 213
- 21a multiplies the reference signals TT and QT, adders 221 and 222 add the multiplication results, and integrator 23+ integrates the first term of equation (9) to obtain the real part correlation result IC. Output. Integrator 232
(9) Integrate the second term of the second term, and get the imaginary part correlation result Q
Output c.

Here, the real part 1y of the reference signal ST is cos(πμj2)
, the imaginary part Qy is 5 inches (πμi2), and when inputting IRA shown by equation (8) to the real part IR of the optimal filter, and inputting QR^ shown by equation (7) to the imaginary part QR, the optimal filter 8a is The output will be as shown below.

・・・・・・(10) ・・・・・・(1) Expressing equations (10) and (11) together as a complex number, RA (?) = ICA (?) + j QCA
(τ) at this optimal filter output.

The amplitude term l expressed as 5in(πΔfτ)/πΔfτ
Ra(τ)1 is shown in FIG. 2(C), and the phase term 7krg(RA(τ)) represented by a π radius τ200 is shown in FIG. 2(d).

It can be seen from this that the pulse width is compressed to approximately 1/Δf, and that the pulse width has no relation to the original pulse width and is determined only by the frequency sweep width Δf. The compression ratio between the input and output of this optimal filter is TΔf, and the law of conservation of energy inevitably increases the amplitude, resulting in an improvement in the signal-to-noise ratio of V/T-ζ]. Incidentally, in the pulse compression result shown in FIG. 3, unnecessary side lobes are generated on both sides of the center pulse, but this can be reduced by weighting the chirp signal waveform on the frequency axis.

On the other hand, the signal Se from the receiver 2b expressed by equation (2) is amplified to an appropriate size by the receiving amplifier 3b in the same way as the signal Sa, and is then amplified to an appropriate size by the frequency mixers 4b18 and 4b2.
The signal is orthogonally detected by the low-pass filter 7b and 7b2 to obtain the following orthogonally detected output.

The imaginary component QR[l of this orthogonal detection output is shown in FIG. 2(b) in correspondence with the signal QR^ expressed by the above equation (7).

When this pulse is compressed by the optimal filter 8b, the output becomes: Real part: ......(15) Imaginary part: ......(io) Summarizing equation (15) and (1B), Rs(τ)
= Ice (() + j Qce (?), and the amplitude product of this optimal filter output is IRB (?)
1 is shown in FIG. 2(e), and the phase term Arg(Re(τ)) represented by πgτ2+θ+ξ is shown in FIG. 2(f).

Since the information necessary for detecting the ultrasound arrival direction α is the phase difference ξ occurring in the signals between the receivers, the following complex number fundamental theorem Ae-j'eBej'1 = ABej('z-II
) "Using (18), find the phase difference ξ by the conjugate complex product of equations (12) and (17). That is, RA#s Rs = (Ica-jQc^) (Ice
+jQce)=IcAIce+QcAQce+j(I
cAQce-QcAIcs) (19a) From this, the phase difference ξ can be more easily determined and becomes a constant value as shown in FIG. 2(h). Multipliers 9al, 9a2, 9b in FIG.
+9 b2 and adder 10a@10b are (19a)
The adder 10a outputs the real part and the adder tob outputs the imaginary part. And jan
The -1 function calculator 11 extracts the phase difference ξ by the transformation shown by equation (20), and the primary direction calculation circuit 12 calculates the equation (4) to determine the ultrasound arrival direction α. Further, the power I RA''R[l I of the arriving wave signal can be determined by the following equation, and has a waveform as shown in FIG. 2(g).

IR^”RB 1 Furthermore, since the signal-to-noise ratio is at its highest at the peak of this pulse-compressed waveform, the phase difference ξ is at the point indicated by the circle in Fig. 2 (h). Most accurate.

The calculation of equation (21a) is performed using the squarers 13+ and 13 in FIG.
2, the adder 14 and the square root calculator 15 output the 0 display circuit 16 as the power of the input signal to the display circuit 16. Based on this, the position of the ultrasonic sound source, such as a totance bonder, is displayed on a CRT using a B-scope or the like that displays distance on the vertical axis and direction on the horizontal axis.

In the above embodiment, two-dimensional position detection using two receiving systems has been described, but in the same way, another receiver is arranged orthogonally to the receiver rows 2a and 2b, and ultrasonic waves arrive. If the direction is detected, three-dimensional position detection is also easily possible.

It is also possible to apply the present invention to a fish school distribution measuring device that detects the swimming position of a school of fish by transmitting a chirp signal using a transmitter and capturing echoes reflected from the school of fish. In this case, distance resolution is improved due to the pulse compression effect of the chirp signal, making it easier to separate and identify between fish schools or fish bodies, and it is possible to output scattered intensity with less influence of interference.

〔Effect of the invention〕

As explained above, the ultrasonic position detecting device of the present invention performs pulse compression using a chirp signal to increase distance resolution, and at the same time improves the signal-to-noise ratio by the compression gain, so that the direction of arrival of the ultrasonic wave is determined by the compressed pulse. Since the phase component can be determined more accurately than the phase component of , there is an advantage that highly accurate ultrasonic position detection is possible.

[Brief explanation of the drawing]

Fig. 1 is a block diagram showing an embodiment of the ultrasonic position detection device of the present invention, Fig. 2 is a time chart diagram for explaining its operation, and Fig. 3 is a block diagram showing an embodiment of the optimal filter. It is. i,,,,bottom 2a/2b 0. Receiver 3a
・3 b,...,, receiving amplifier 4 al-4al・4 b+・4 b29. , Frequency mixer 5 , Local oscillator 6. ,,,,9
0 degree phase shifter 7 al-7al・7 bB7 b2 , , low-pass filter 8a・8b optimal filter 9 a+・9 a2-9 bl・9 b2. -0 multipliers 10a-10b, Adder 11...tanl function calculator 12, Direction calculation circuits 13a and 13b, Squarer 14, Adder Vessel 15. ,,, Square root operation base 16...Display circuit 20+, 202, =- delay memory 211, 212, 213, 21a-160 multiplier 221
・222・・・Adder 231・232 ,,,,, ii Divider Figure 2 37 (Voluntary) Procedural Amendment January 1986 61 Commissioner of the Japan Patent Office Kuro 1) Akio Tono 1, Case Indication of 1985 Patent Application No. 310% 2+ No. 2, Name of the invention Ultrasonic position detection device 3, Relationship with the case of the person making the amendment Patent applicant address 5-1 #il, Shimorenjaku, Mitaka City, Tokyo Title of the invention
(433) Japan Radio Co., Ltd. 4, Agent 160 Address Azalea Building, 2-42-13 Kabukicho, Shinjuku-ku, Tokyo Telephone 232-east Name (
8B10) Patent attorney Yoshio Komiya 5, "Detailed explanation of the invention" column of the specification to be amended (1) Page 10 of the specification, lines 15 to 17, "Real part: ...... (15) J ``Real part: ...... (15) J (2) Same < Page 10, lines 18 to 20, ``Imaginary part: ...... (1G) J'' Imaginary part: ......(1B)J (3) The same <12 pages, 18 lines r13+J is r
13aJ, (0 same < 13 pages 1 line r132J is corrected to r13b).

Claims (1)

[Claims] 1. First and second receivers placed in close proximity such that the phase difference in the center frequency of the received incoming wave is within 180 degrees, and for converting the chirp signal received by each of the receivers into a complex number. first and second quadrature detection circuits;
It includes first and second optimal filters that pulse-compress the complex chirp signal, and measures a phase difference based on the receiver arrangement from the complex output signal pulse-compressed by the first and second optimal filters. An ultrasonic position detecting device characterized in that the ultrasonic sound source position is detected by simultaneously obtaining the direction of arrival of the ultrasonic wave and obtaining the signal power from the pulse-compressed complex number output signal.