JPS6113434B2 - - Google Patents
Info
- Publication number
- JPS6113434B2 JPS6113434B2 JP55090278A JP9027880A JPS6113434B2 JP S6113434 B2 JPS6113434 B2 JP S6113434B2 JP 55090278 A JP55090278 A JP 55090278A JP 9027880 A JP9027880 A JP 9027880A JP S6113434 B2 JPS6113434 B2 JP S6113434B2
- Authority
- JP
- Japan
- Prior art keywords
- frequency
- filter
- intermediate frequency
- output
- video
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
Links
- 230000010355 oscillation Effects 0.000 claims description 4
- 239000000203 mixture Substances 0.000 claims 1
- 238000010586 diagram Methods 0.000 description 9
- 238000000034 method Methods 0.000 description 2
- 230000035945 sensitivity Effects 0.000 description 2
- 238000010897 surface acoustic wave method Methods 0.000 description 2
- 230000005540 biological transmission Effects 0.000 description 1
- 239000003990 capacitor Substances 0.000 description 1
- 238000001514 detection method Methods 0.000 description 1
- 230000006866 deterioration Effects 0.000 description 1
- 230000005236 sound signal Effects 0.000 description 1
- 238000011144 upstream manufacturing Methods 0.000 description 1
Classifications
-
- H—ELECTRICITY
- H04—ELECTRIC COMMUNICATION TECHNIQUE
- H04N—PICTORIAL COMMUNICATION, e.g. TELEVISION
- H04N5/00—Details of television systems
- H04N5/44—Receiver circuitry for the reception of television signals according to analogue transmission standards
- H04N5/60—Receiver circuitry for the reception of television signals according to analogue transmission standards for the sound signals
- H04N5/607—Receiver circuitry for the reception of television signals according to analogue transmission standards for the sound signals for more than one sound signal, e.g. stereo, multilanguages
Landscapes
- Engineering & Computer Science (AREA)
- Multimedia (AREA)
- Signal Processing (AREA)
- Television Receiver Circuits (AREA)
Description
【発明の詳細な説明】
本発明は、インターキヤリヤ方式の音声多重テ
レビジヨン受信機に関し、特にバスによる音質の
劣化を改善することを目的としたものである。DETAILED DESCRIPTION OF THE INVENTION The present invention relates to an intercarrier audio multiplex television receiver, and is particularly aimed at improving the deterioration of sound quality caused by buses.
第１図に従来のインターキヤリヤ方式音声多重
テレビジヨン受信機の要部ブロツク構成図を示
す。同図において、端子１から入力された高周波
入力は、混合器２で局部発振器３の出力と混合さ
れ、中間周波信号に変換される。その中間周波信
号には映像搬送波と音声搬送波が含まれ、共に映
像中間周波増幅器４で増幅される。その出力は映
像検波端子５に与えられるとともに、音声検波器
６に加えられる。音声検波器６では、映像搬送波
と音声搬送波の差の周波数（4.5MHz）を中間周
波数とする音声中間周波信号が得られる。この音
声中間周波信号は音声中間周波増幅器７で増幅さ
れ、周波数弁別器８で周波数弁別される。その出
力は多重復調器端子９を経て多重復調器（図示せ
ず）で復調される。 FIG. 1 shows a block diagram of the main parts of a conventional intercarrier audio multiplex television receiver. In the figure, a high frequency input from a terminal 1 is mixed with the output of a local oscillator 3 in a mixer 2 and converted into an intermediate frequency signal. The intermediate frequency signal includes a video carrier wave and an audio carrier wave, both of which are amplified by the video intermediate frequency amplifier 4. The output is given to the video detection terminal 5 and also added to the audio detector 6. The audio detector 6 obtains an audio intermediate frequency signal whose intermediate frequency is the difference frequency (4.5 MHz) between the video carrier wave and the audio carrier wave. This audio intermediate frequency signal is amplified by an audio intermediate frequency amplifier 7 and subjected to frequency discrimination by a frequency discriminator 8. The output is demodulated via a multiple demodulator terminal 9 by a multiple demodulator (not shown).
以上のインターキヤリヤ方式では、映像搬送波
と音声搬送波の差の周波数成分（4.5MHz）を音
声中間周波とするので、局部発振器３の出力周波
数が変動しても、音声中間周波数は変動しないと
いう特長をもつが、その反面、音質を害するバズ
が本質的に発生するという欠点がある。バズと
は、映像信号成分が音声チヤンネルに混入するた
めに生ずる妨害のことである。 In the intercarrier method described above, the frequency component (4.5MHz) of the difference between the video carrier wave and the audio carrier wave is used as the audio intermediate frequency, so even if the output frequency of the local oscillator 3 changes, the audio intermediate frequency does not change. However, on the other hand, it has the disadvantage that it inherently generates buzz that impairs sound quality. Buzz is an interference caused by the mixing of video signal components into the audio channel.
バズの本質的な発生原因として、テレビジヨン
受信機の映像中間周波選択度特性の影響がある。
映像中間周波選択度特性は、テレビジヨン映像電
波が残留側波帯伝送を行つているため、第２図に
示すように映像搬送波の付近では、傾斜特性（ナ
イキストスロープ）を持つている。このため、搬
送映像信号の上下側帯波間にレベル差が生じ、第
３図のベクトル図に示すように映像搬送波に位相
変調が発生する。この位相変調分が音声中間周波
信号に混入し、バズ妨害となる。ナイキストスロ
ープによる位相変調分Φは次式で与えられる。 The essential cause of buzz is the influence of the video intermediate frequency selectivity characteristics of television receivers.
The video intermediate frequency selectivity characteristic has a slope characteristic (Nyquist slope) near the video carrier wave, as shown in FIG. 2, because the television video radio wave performs vestigial sideband transmission. Therefore, a level difference occurs between the upper and lower sidebands of the carrier video signal, and phase modulation occurs in the video carrier wave as shown in the vector diagram of FIG. This phase modulation component mixes into the audio intermediate frequency signal, resulting in buzz interference. The phase modulation amount Φ due to the Nyquist slope is given by the following equation.
Φ＝tan-1（ｍβＳｉｎ２πｖｔ／１＋ｍｃｏｓ２π
ｖｔ）
〔ラジアン〕 ……(1)
ここで、v
：映像変調周波数（KHz）
ｍ：v成分による振幅変調度（Ｏ＜ｍ＜１）
α：映像搬送波付近の特性の傾斜（１／KHz）
β：αv
そして、Φによる等価FM分Δは、β≪１と
すると、
Δ＝ｍ（ｍ＋ｃｏｓ２πｖｔ）／（１＋ｍｃｏｓ２
πｖｔ）２ vβ……(2)
となり、Δの最大値は、２πvｔ＝πの時
で、
Δnax＝ｍ／１−ｍvβ＝ｍ／１−ｍv ２α……(3)
となる。すなわち、音声中間周波信号は、映像信
号により最大周波数偏移がΔmaxの周波数変調
を受けたものと等価となる。Φ=tan -1 (mβSin2π v t/1+mcos2π
v t) [Radian] ...(1) Where, v : Video modulation frequency (KHz) m: Amplitude modulation degree by v component (O<m<1) α: Slope of characteristics near the video carrier wave (1/KHz) ) β:α vAnd the equivalent FM component Δ due to Φ is, assuming β≪1, Δ=m(m+cos2π v t)/(1+mcos2
π v t) 2 v β...(2), and the maximum value of Δ is when 2π v t=π, Δ nax = m/1-m v β=m/1-m v 2 α... (3) becomes. In other words, the audio intermediate frequency signal is equivalent to a signal subjected to frequency modulation with a maximum frequency deviation of Δmax by the video signal.
本発明は、この周波数変調による妨害を減少さ
せようとするものである。 The present invention attempts to reduce the interference caused by this frequency modulation.
本発明の実施例を第４図に、そして、その動作
を解析するためのモデルを第５図に示す。 An embodiment of the present invention is shown in FIG. 4, and a model for analyzing its operation is shown in FIG.
第４図において、１〜９は第１図の従来例と同
一の部分を示し、その動作も同様である。ただ
し、３は電圧制御型の局部発振器となつている。
１０は後述の位相特性をもつフイルタ、１１は局
部発振器３に含まれる発振周波数可変用の可変リ
アクタンス素子に電圧を印加するための電圧供給
回路である。 In FIG. 4, numerals 1 to 9 indicate the same parts as in the conventional example of FIG. 1, and their operations are also the same. However, numeral 3 is a voltage-controlled local oscillator.
Reference numeral 10 denotes a filter having a phase characteristic to be described later, and reference numeral 11 denotes a voltage supply circuit for applying a voltage to a variable reactance element for varying the oscillation frequency included in the local oscillator 3.
第５図のモデルにおける端子１２は第４図の端
子１に対応し、ここに与えられる高周波入力は
iなる周波数偏移をもつ音声搬送波を含んでい
る。周波数加算器１３は第４図の混合器２に対応
し、ここで上記周波数偏移ｉに、負帰還周波数
β（Vo＋Vn）が加算される。１４は音声搬送周
波数付近でμ１なる周波数偏移の利得（すなわち
入力信号の周波数が変化すると出力信号の位相が
偏移するような特性を有しているフイルタにおけ
る入力周波数対出力位相偏移量特性の傾き）をも
つフイルタで、第４図のフイルタ１０に対応す
る。周波数加算器１５では、前記の式(2)で示され
た周波数偏移をｎとして表わして加えてある。
１６は周波数弁別器８に対応し、その感度はμ２
である。１７は第４図の電圧供給回路１１と局部
発振器３を含めてなる負帰還回路に対応する回路
であり、その感度はβである。 Terminal 12 in the model of Fig. 5 corresponds to terminal 1 of Fig. 4, and the high frequency input given here is
It contains an audio carrier wave with a frequency shift of i . The frequency adder 13 corresponds to the mixer 2 in FIG. 4, and here a negative feedback frequency β (Vo+Vn) is added to the frequency deviation i. 14 is a frequency deviation gain of μ 1 near the audio carrier frequency (i.e., the amount of input frequency versus output phase deviation in a filter that has a characteristic that the phase of the output signal shifts when the frequency of the input signal changes) This filter corresponds to the filter 10 in FIG. 4. In the frequency adder 15, the frequency shift expressed by the above equation (2) is expressed as n and added.
16 corresponds to the frequency discriminator 8, whose sensitivity is μ 2
It is. 17 is a circuit corresponding to the negative feedback circuit including the voltage supply circuit 11 and local oscillator 3 shown in FIG. 4, and its sensitivity is β.
第５図で、μ＝μ１・μ２とすると、
Vo＝μ（ｉ＋βVo）
Vn＝μ２ｎ＋μβVn
であるから、
Vo＝μ／１−μβ×ｉ≒−１／β×ｉ ……(4)
Vn＝μ２／１−μβ×o≒−μ２／μβ×ｎ……(5)
（ただし、μβ≫１とする。）
すなわち、負帰還がかかつていない場合の妨害
電圧μ２ｎに比し、本発明による構成では、妨
害電圧Vnはμβ分の１に減少している。 In Figure 5, if μ=μ 1・μ 2 , then Vo=μ(i+βVo) Vn=μ 2 n+μβVn, so Vo=μ/1−μβ×i≒−1/β×i ……(4 ) Vn=μ 2 /1−μβ× o ≒−μ 2 /μβ×n……(5) (However, μβ≫1.) In other words, the disturbance voltage μ 2 n when there is no negative feedback In contrast, in the configuration according to the present invention, the disturbance voltage Vn is reduced by a factor of μβ.
次に音声搬送周波数付近でμ１なる周波数偏移
の利得をもつフイルタ１４の構成法について述べ
る。第６図で、フイルタ１９への入力e1を、
e1＝Asin（ωcｔ＋Δωｄ／ｐsinpt） ……(6)
出力e2を、
e2＝Asin（ωcｔ＋μ１Δωｄ／ｐsinpt） ……(7)
とする。ここで、
Ａ：振幅
ωc：音声搬送周波数
Δωd：入力周波数偏移
ｐ：音声信号周波数
である。フイルタ１９への入力周波数ω１と、出
力周波数ω２は、
ω１＝ωc＋Δωdcospt ……(8)
ω２＝ωc＋μ１Δωdcospt ……(9)
であり、その出力信号の位相をΦとすると、ωc
は一定、Δωdはｔの関数であるが故に、
ｄω２／ｄω１＝ｄ／ｄω１・ｄΦ／ｄｔ＝ｄ／ｄｔ・
ｄΦ／ｄω１＝μ１……(10)
したがつて、
ｄΦ／ｄω１＝μ1t＋γ（γ：積分定数） ……(11)
すなわち、遅延時間ｄΦ／ｄω１が周波数偏移
の利得μ１と時間ｔとの間に式(11)の関係をもつフ
イルタを構成すればよい。このようなフイルタは
表面弾性波フイルタで得ることが考えられる。 Next, a method of constructing the filter 14 having a gain of a frequency shift of μ 1 near the audio carrier frequency will be described. In Fig. 6, the input e 1 to the filter 19 is expressed as e 1 = Asin (ω c t + Δω d /psinpt) ...(6) The output e 2 is expressed as e 2 = Asin (ω c t + μ 1 Δω d / psinpt) ...(7). Here, A: amplitude ω c : audio carrier frequency Δω d : input frequency deviation p: audio signal frequency. The input frequency ω 1 to the filter 19 and the output frequency ω 2 are ω 1 =ω c +Δω d cospt (8) ω 2 =ω c +μ 1 Δω d cospt (9), and the output signal thereof is Let Φ be the phase of ω c
is constant and Δω d is a function of t, so dω 2 /dω 1 = d/dω 1・dΦ/dt=d/dt・
dΦ/dω 1 = μ 1 ...(10) Therefore, dΦ/dω 1 = μ 1 t+γ (γ: integral constant) ...(11) In other words, the delay time dΦ/dω 1 is the frequency deviation gain μ What is necessary is to construct a filter having the relationship shown in equation (11) between 1 and time t. It is conceivable that such a filter can be obtained by a surface acoustic wave filter.
以上の説明から明らかなように本発明は次のよ
うな優れた特長を有するものである。 As is clear from the above description, the present invention has the following excellent features.
(1) 周波数偏移の利得がμ１なるフイルタを、映
像搬送波付近に傾斜特性をもつ映像中間周波増
幅器の前段に設け、音声周波数弁別器の出力に
よつて制御される局部発振器の出力を、上記フ
イルタの入力側に負帰還することにより、イン
ターキヤリヤ方式にもかかわらず、バズ妨害を
減少させることができる。(1) A filter with a frequency shift gain of μ 1 is provided upstream of a video intermediate frequency amplifier having a slope characteristic near the video carrier, and the output of the local oscillator controlled by the output of the audio frequency discriminator is By providing negative feedback to the input side of the filter, buzz interference can be reduced despite the intercarrier system.
(2) 上記フイルタは前記式(11)で示す特性を有して
おればよい。これを表面弾性波フイルタで実現
すれば、インダクタンスやキヤパシタや能動素
子が不要となるので、容易に小型化、無調整化
し得る。(2) The above-mentioned filter should just have the characteristic shown in the above-mentioned formula (11). If this is achieved using a surface acoustic wave filter, inductance, capacitors, and active elements are no longer required, making it easy to downsize and eliminate adjustment.
第１図は従来のインターキヤリヤ方式音声多重
テレビジヨン受信機の要部ブロツク図、第２図は
テレビジヨン受信機の映像中間周波選択度特性
図、第３図は映像中間周波選択度特性の影響によ
つて生ずる位相変調のベクトル図、第４図は本発
明の実施例の要部ブロツク図、第５図は同本発明
の実施例の動作を説明するためのモデル図、第６
図は本発明に用いるフイルタの入出力の関係を説
明するためのブロツク図である。
２……混合器、３……局部発振器、４……映像
中間周波増幅器、６……音声検波器、７……音声
中間周波増幅器、８……周波数弁別器、１０，１
４，１９……フイルタ、１１……電圧供給回路。
Figure 1 is a block diagram of the main parts of a conventional intercarrier audio multiplexing television receiver, Figure 2 is a diagram of video intermediate frequency selectivity characteristics of the television receiver, and Figure 3 is a diagram of video intermediate frequency selectivity characteristics of the television receiver. FIG. 4 is a block diagram of the main part of the embodiment of the present invention, FIG. 5 is a model diagram for explaining the operation of the embodiment of the present invention, and FIG. 6 is a vector diagram of phase modulation caused by the influence.
The figure is a block diagram for explaining the input/output relationship of the filter used in the present invention. 2...Mixer, 3...Local oscillator, 4...Video intermediate frequency amplifier, 6...Audio detector, 7...Audio intermediate frequency amplifier, 8...Frequency discriminator, 10, 1
4, 19...filter, 11...voltage supply circuit.
Claims (1)
特性を有している映像中間周波増幅器と、前記映
像中間周波増幅器の前段に設けられｄΦ／ｄω１＝μ1t
＋ γ（ただし、ｄΦ／ｄω１はフイルタの群遅延時間、ω
１ はフイルタへの入力周波数、Φはフイルタの出力
信号の位相、μ１はフイルタにおける入力周波数
対出力位相偏移量特性の傾き、ｔは時間、γは定
数）なる特性を有するフイルタと、前記映像中間
周波増幅器の出力側から得られた音声中間周波信
号の周波数を弁別する周波数弁別器と、前記周波
数弁別器の弁別出力によつて発振周波数が制御さ
れ前記映像中間周波増幅器の傾斜特性によつて生
じる映像中間周波信号の位相変調成分を補償する
ことのできる発振出力を発生する局部発振器と、
前記フイルタの入力側に設けられ前記局部発振器
の発振出力と高周波入力とを混合して前記フイル
タに入力する混合器とを備えた音声多重テレビジ
ヨン受信器。[Scope of Claims] 1. A video intermediate frequency amplifier whose frequency characteristic at a video carrier frequency has a slope characteristic, and a video intermediate frequency amplifier provided at a stage before the video intermediate frequency amplifier, dΦ/dω 1 =μ 1 t
+ γ (however, dΦ/dω 1 is the group delay time of the filter, ω
1 is the input frequency to the filter, Φ is the phase of the output signal of the filter, μ 1 is the slope of the input frequency vs. output phase deviation characteristic in the filter, t is time, and γ is a constant); a frequency discriminator for discriminating the frequency of the audio intermediate frequency signal obtained from the output side of the video intermediate frequency amplifier, and an oscillation frequency controlled by the discrimination output of the frequency discriminator and according to the slope characteristic of the video intermediate frequency amplifier. a local oscillator that generates an oscillation output capable of compensating for the phase modulation component of the video intermediate frequency signal that occurs when
An audio multiplex television receiver comprising: a mixer provided on the input side of the filter for mixing the oscillation output of the local oscillator and a high frequency input and inputting the mixture to the filter.
Priority Applications (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP9027880A JPS5715580A (en) | 1980-07-01 | 1980-07-01 | Voice multiplex television receiver |
Applications Claiming Priority (1)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| JP9027880A JPS5715580A (en) | 1980-07-01 | 1980-07-01 | Voice multiplex television receiver |
Publications (2)
| Publication Number | Publication Date |
|---|---|
| JPS5715580A JPS5715580A (en) | 1982-01-26 |
| JPS6113434B2 true JPS6113434B2 (en) | 1986-04-14 |
Family
ID=13994044
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| JP9027880A Granted JPS5715580A (en) | 1980-07-01 | 1980-07-01 | Voice multiplex television receiver |
Country Status (1)
| Country | Link |
|---|---|
| JP (1) | JPS5715580A (en) |
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPH0318787A (en) * | 1989-06-15 | 1991-01-28 | Matsushita Electric Ind Co Ltd | Seismographic device |
-
1980
- 1980-07-01 JP JP9027880A patent/JPS5715580A/en active Granted
Cited By (1)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| JPH0318787A (en) * | 1989-06-15 | 1991-01-28 | Matsushita Electric Ind Co Ltd | Seismographic device |
Also Published As
| Publication number | Publication date |
|---|---|
| JPS5715580A (en) | 1982-01-26 |
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