JPS6020787A - Digital phase control system of brushless motor - Google Patents

Abstract

PURPOSE:To accurately control the rotation of a brushless motor by setting the timing of an exciting pulse so that the symmetrical relationship is held before and after the center of the time when the rate of change becomes zero in the zone that the induced voltages of the excited phase and the rate of change of the torque become minimum. CONSTITUTION:The output of a speed detector 6 is compared by a phase comparator 19 with the phase of a reference pulse from 12-stage frequency divider 17, the logic sum of the output and the output pulse-width-modulated are taken by an OR gate 28, further the logic product of the output of a starter 21 is taken by an AND gate 29, and inputted to a distributor 23. Further, the rotor position is detected by Hall elements H1, H2, the outputs are inputted through waveform shapers 24, 25 to the distributor 23. The timing of generating the exciting pulse at every one phase is set to hold the symmetrical relationship before and after the center of the time when the rate of change becomes zero in the zone that the induced voltages of the excited phase and the rate of change of the torque become minimum.

Description

[Detailed Description of the Invention] Technical Field The present invention relates to a digital phase control method for brushless motors that is extremely effective against noise and drift, and also enables highly accurate constant speed drive.
The present invention relates to a digital phase control method that suppresses the occurrence of rotational unevenness even when rotating at high speeds such as 10,000 rotations, making it possible to control rotation with extremely high precision.

BACKGROUND ART Brushless motors are motors that rotate at high speed and at a constant speed in complete synchronization with a reference clock, and are used to drive VTRs, printers, and other devices.

FIG. 7 is a partially cutaway plan view showing an example of the internal structure of the branless motor. In the drawings, l is a magnetic circuit component, /A is its magnet rotor, /B is a stator, /C is a gap, C is a winding drive part, '3 is a rotor position detection part, and t is a bearing part.

The left side shows the rotation axis, and the second side shows the speed detection section.

This figure 1 is a %G phase co-pole brushless motor (BLM).
In the case of , the magnetic circuit component / is attached to the axis of one rotation, and the magnet rotor /A and % are magnetized in the axial direction.
Stator/B of phase slot.

and a gap /C formed between the magnet rotor /A and the stator /B.

The winding drive unit that generates the rotational force has a coil for each phase and a coil for each phase. These daphase coils are driven by seven or more excitation pulses given to each phase from a distribution circuit (not shown), and the current position of the rotor A is detected by a rotor position detection section 3.

The rotor position detection unit 3 has a mechanical angle of “rorl rad”.
It is composed of one Hall element arranged in a shifted positional relationship.

The speed detection f%6 is composed of, for example, a reluctance type speed detector using a multipolar magnet, and performs highly accurate speed detection.

In addition, a high-precision bearing is used for the bearing part, and good rotation of the rotating shaft S is guaranteed.

FIG. 2 is an example of a drive circuit. In the drawings, LA
-LD is the excitation winding of BLM, QA~QD are drive transistors, DA-DD are backflow prevention diodes, R/~R
2 and RA-RD are resistors, C is a capacitor, P is the collector terminal of transistor QA, and ΦA to ΦD are excitation pulses of each phase.

Conventional rotation control of the BLM shown in FIGS. 1 and 2 was performed in an analog manner.

FIG. 3 is a block diagram showing a conventional servo control system. In the drawing, ■ is the reference voltage, and θ is the angular velocity (rad
/3ec ], 7 to /4 are control system processing circuits, and 1
0 and // indicate the speed control system % /2 to /6 indicate the decoy control system. Note that 1 m is the number of slits in the encoder.

In the conventional servo control system, as shown in Fig. 3,
The rotational speed and phase difference of the BLM were detected in parallel, and the speed deviation signal and the phase deviation signal were converted into a combined analog signal for control.

The speed control system and the phase control system are run in parallel.
The l'i control method is especially effective in rotation control where starting characteristics are a problem and torque fluctuations are involved.

That is, the speed deviation and P L L (Phase
If the rotation is controlled based on phase correction by LockedLoop) control, rotation at a desired speed can be obtained.

However, in the case of analog processing like this.

For example, when controlling the cylinders of a VTR, the circuit size becomes large and there are problems with noise resistance, drift, etc.

In addition, there is a system for performing such servo control digitally.
The t11 method is also known.

For example, a signal proportional to the rotational speed (speed signal) is digitally extracted from the rotor position detection signal.

Constant speed driving is performed by controlling the pulse width of the excitation pulse using this signal.

According to such a control system, a speed signal can be obtained by simply providing a position detecting means, so there is an advantage that a means for detecting the number of revolutions (velocity) per revolution is not required. However, if there is magnetization unevenness in the rotor, a detection error will occur in the position detection signal itself, and even if it is digitally processed, the error will remain as it is, making it impossible to perform accurate rotation control. It's inconvenient.

Purpose Therefore, the digital phase control method for brushless motors of the present invention solves these inconveniences in the rotation control of BLM used in conventional servo systems, and improves noise resistance and drift by using simple control means. It is an object of the present invention to provide highly accurate rotation 101'' control that is extremely effective and has extremely little rotational unevenness.

The digital phase control method of the brushless motor of this invention is particularly effective for applications such as polygon mirrors in laser printers, where starting characteristics are not a problem and torque fluctuations are small. However, it enables high-precision, constant-speed rotation with almost no rotational unevenness and no risk of being affected by uneven magnetization of the rotor.

Structure: Therefore, in the brushless morph dissicle position control method of the present invention, the induced phase of the excited phase 1. , 27'/ where the rate of change of IL pressure and torque is minimum
11 (where n is the number 8 of the motor) The timing of the excitation pulses is set so that a symmetrical relationship is maintained around the time when hj and the rate of change is zero in time. There is.

11th. FIG. 1 is a block diagram showing an example of the configuration of a control circuit used when implementing the brushless morph digital phase control method of the present invention. The symbols in Figure mJ are the same as in Figures 1 and 2, /7 is a /co-stage frequency divider, /g is a waveform shaping circuit, /q is a phase comparator, U0 is a pulse width modulator, 21 is the starting circuit, 2 is the /phase unipolar drive circuit, 3 is the distribution circuit, 2 and 2 are the waveform forming circuits 1.2 and 2, 27 is the impark, 2g is the OR gate circuit, and Cobb is the AND The gate circuit is shown in red, and θ1 is the frequency of the reference clock output from the two-stage frequency divider/7.

θ is the frequency of the speed detection pulse, θ. K7θ8 indicates a phase detection pulse having a pulse width corresponding to the deviation between the frequency θ□ of the reference clock and the frequency 02 of the speed detection pulse, and K7θ8 indicates a control signal output from the pulse width modulator λθ. In addition,
H/ and H2 are Hall elements for detecting the position of the rotor.

As already explained, in the D-phase BLM, the G-phase winding L
A-LD is provided, and a magnetic roller/A
Hall elements H/ and H are arranged with a mechanical angle of πl/, 2 Crad).

The signals output from the Hall elements H/ and H2 are shaped into square wave pulses by the waveform shaping circuits 2 and J4, respectively, and are given to the distribution circuit 13 along with the inverted signals via imparks 27 and 26. .

The distribution circuit 3 is provided with an encoder that encodes these signals and generates an encoded output, and a logic circuit that performs AND processing on this encoded output and a control signal inputted via the AND gate circuit 2q. hand,
An excitation pulse is output to the drive circuit 22 shown in detail in FIG. 2, which drives the excitation coils LA-LD of the BLM.

At startup (rotation speed = θ), the output of the pulse width modulator θ is KTθ. is in the state of 1H level, which is applied to the AND gate circuit Utah via the OR gate circuit 2g.

When the %BLM power supply is turned on, the starting circuit 2/2 is activated and the output is set to ``H'' for a certain period of time.

Therefore, the output of the AND gate circuit 29 is also given to the distribution circuit 3 at 1H'. The control signal of this 1H level is
An excitation pulse is generated by AND processing with the output of the encoder in the distribution circuit 3.

Note that this excitation pulse is given to the % drive circuit 2 as a logic level signal.

With this excitation pulse, the drive circuit here is connected to the da-phase winding LA-
By driving either/phase of the LD, the BLM starts rotating.

The rotation of the BLM is detected by two Hall elements H/ and H2 that detect the rotor position, and the encode output changes depending on the input to the distribution circuit 23, and the excitation phase is sequentially switched, and the rotation of the BLM is detected by the two Hall elements H/ and H2 that detect the rotor position. Speed also increases.

If the gain characteristics of the control system are sufficient without the pulse width modulator 20, the speed of the BLM will increase and the frequency 02 of the output pulse of the speed detector B will also increase at the same time, and the phase comparator/ When entering the phase detection range 9, the 1-phase control system performs a synchronous pull-in operation to rotate the BLM synchronously with respect to the frequency θ of the reference clock. Furthermore, the phase detection range is as follows.

If the period T of the reference clock is λπ [rad], then the range is approximately π (rad).

In such a speed range, a digital phase comparator/
The phase detection pulse θ8 from q becomes a pulse signal with a pulse width that accurately detects the deviation between the frequency 01 of the reference clock and the frequency 02 of the waveform-shaped output pulse of the speed detection unit B. That is, this phase detection pulse θ8 is a signal having the same period as the period T of the reference chlorane and a pulse width corresponding to the deviation between the frequency θ and 02.

However, in general, this phase detection pulse θ. With a pulse width of , it is not possible to increase the speed of the BLM to the specified speed. Therefore, this pulse θ is adjusted by the pulse width modulator θ. Further widen the pulse width of 7 to 5 output pulses.
θ. to occur. In the circuit of FIG. 7, for reasons to be explained in detail later, the control signal rd8 output from the pulse width modulator λθ is supplied to the distribution circuit 3 via the OR gate circuit g. There is.

When the speed of the BLM continues to increase and the output frequency 02 from the speed detection unit B matches the frequency θ of the reference clock, the BLM rotates in complete synchronization with the reference clock and rotates at a specified rotation speed.

Figure S shows 1l of the control circuit in Figure 7 during synchronous rotation.
1 is a time chart of ilJ M'l system signals. The code found in each signal waveform ζζ corresponds to the code position in the %G diagram. Incidentally, the deviation between the output of the phase comparator/9, ie, θ, and θ2 during this synchronous rotation (steady rotation) is called a steady phase deviation.

As described above, since the BLM performs constant speed rotation with high precision, it is an extremely effective control means when used in this state, that is, when the starting characteristics are not much of a problem and torque fluctuations are small.

Therefore, for such purposes of use, it is possible to control the rotation with extremely high precision using only a single phase control (PLL) system without using a 5-speed control system. The digital phase control method of this invention is
Focusing on these points, the present invention aims to further suppress the occurrence of rotational unevenness.

Here, to express the operation of the BLM using a mathematical formula, the induced voltage ke of the BLM is: ke=Kecosθ・−(1) Ke: Maximum value of the induced voltage constant of the BLM CV Sec rd−′] Also, the torque kt of the BLM is kt-KtCO5θ (2) Kt: Maximum torque constant value of BLM CKg f cm A-”) The terminal voltage V of the motor is.

i, L: Excitation current of i phase [A/phase] Ra: Winding resistance of /phase [/phase] L8: Inductance of /phase [H/phase] Furthermore, the fluctuating torque TT generated by the motor is.

'fT' Ktsi @ cos θ = J - ω1 Tf ... (4) t J: Rotor and load inertia [Kg f ''cm Sec 2] D two viscous load (Kg f 0m/rad/Beo
]Tf: Friction torque [Kg f cm ).

becomes. This equation (5) also holds true when other phases are excited. Note that from Equation 1 (1), KtCO3θ=kt.

From this equation (5), it is clear that when considering the conditions for rotating the BLM with the minimum excitation current, it is sufficient to use the section where kt is maximum for each phase.

In the case of the /phase unipolar drive of the BLM with the phase co-pole, each phase shares the excitation of π/a (rad), so
Ki-! If the magnet is excited in the cη section, maximum efficiency can be obtained in this /phase unipolar drive.

Next, a case where a disturbance torque exists and TT in the above equation (4) fluctuates at the angular frequency ω' will be described.

Note that only the varying torque will be considered here.

In this case, equation (4) becomes: TT= TPcosω′t”' Kt iB cosθ
...(6) TP: Can be expressed as the maximum value of fluctuating torque.

FIG. 5 is a time chart for explaining the relationship between the input timing of the control signal, the generation timing of the excitation pulse, and the torque fluctuation during steady rotation operation in the digital phase control system of the present invention. In the drawing, %T0 is /
Occurrence times of the encoded outputs of the phases, t, and t, and t,
is the non-occurrence time of the excitation pulse, and t3 is the input period of the control #4 and represents the output time of the excitation pulse.

Torque fluctuations in this case occur at a specific period determined by the bearings used in the BLM.

In addition, the collector waveform ke of the /phase is obtained by AND processing the output of the encoder in the distribution circuit 23 and the control signal.
It changes as shown in Fig. 6.

With reference to FIG. 6 and equation (6), the timing of generating excitation pulses using the digital phase control system of the present invention will be explained.

First, by subtracting the excitation current la of the /phase from Equation 1 (6), it becomes.

This formula (Formula (3) that expresses the terminal voltage V of the motor in front of the force)
By substituting into , the following is obtained, and when this equation (8) is further converted into an equation for the angular velocity ω of the BLM, the following is obtained.

Considering this equation (9), Kecos θ and K
It turns out that if the magnetization is performed in the "n-[rad]" section of the section where l cos θ is maximum, the waveforms of ke and kt are in phase and the rate of change is small, so it is possible to reduce fluctuations in the angular velocity ω of the BLM. Ru.

To explain this relationship with reference to Figure 6, which shows a time chart during steady rotation, the excitation pulse for each phase can be expressed as % B
Induced voltage ke of LM (torque k of BLM in in-phase relationship)
(The same applies to t) The section where the rate of change is the minimum (Figure 6 shows the collector waveform of the driving transistor, so the excitation pulse has the opposite polarity and corresponds to the maximum section of the waveform. ) and the generation time T of the encode output in the distribution circuit 23. It is only necessary to generate them in an accurate manner.

In other words, with respect to the encode output generation time TO, there is a temporally symmetrical relationship centered around the point in time when the rate of change of ke becomes zero;
+ + 12+-2t+ +3ts , -α0,
And ideally.

If the excitation pulse 1B is generated so that t, =: t24 tv2 '''αD, it is possible to remove the influence of disturbance torque and significantly suppress one-rotation unevenness. ffl+ is mechanical angle.

In order to generate excitation pulses at such timing, the encode output during steady rotation as shown in Fig. 6 must be k.
It is necessary to set the mounting positions of Hall elements H/ and H2 so that they correspond exactly to the section where the rate of change of the waveform is minimum.

The time chart in FIG. 6 is for the case where 7 rotations of the da-phase BLM are driven with 72 excitation pulses, and 3 excitation pulses are given for each phase.

The number of excitation pulses is not necessarily three per phase, and may be an even number instead of an odd number. The number of excitation pulses varies depending on the rotational state of the driven body, etc., and may be set to an optimal number so that unevenness in one rotation is mutually canceled out.

By the way, in order to generate the excitation pulse at the timing shown in FIG. 6, the control signal KTθ8 with a pulse width corresponding to one phase deviation is sent to the distribution circuit, 2.
It is necessary to input it to 3.

Various methods are possible as a signal processing method for this purpose, but in the control circuit shown in FIG. θ8 and the control signal Krθ from the pulse width modulator 20. Input this into the OR gate circuit 2g,
Or is being processed.

FIG. 7 is a time chart showing one embodiment of the control signal correction method used in the present invention.

In the case of pulse width modulating the comparison output θ8 of the phase comparator /q, in the conventional control system, the control signal KTθ8 was generated simply by a pulse width modulator θ consisting of an integrating circuit or the like.

In this case, 7θ0 at the output of the modulator 20 is the comparison output θ, as shown in FIG. There will be a delay with respect to

This delay in the control signal Krθ8 causes uneven rotation of the BLM, so in the digital phase control system of the present invention, one correction method is to perform OR processing using the OR gate circuit 2g and control this OR processed signal. Use as a signal.

If the OR-processed control signal is input to the distribution circuit 23, the excitation pulse can be generated at the timing explained in connection with FIG. This eliminates rotational unevenness, making it possible to control rotation with extremely high precision.

FIG. 3 shows the detection signal and pulse signal θ detected by the speed detection unit B in the digital phase control method of the present invention.
2 is a time chart showing the relationship with 2.

In the GII control circuit in Figure y, the speed detection pulse θ2 output from the waveform shaping circuit ig is shaped into a square wave pulse by a zero-cross amplifier, etc., and compared in phase with the reference clock. Detection accuracy is improved. Note that it is not necessary to shape the waveform into a square wave, and it is sufficient to shape the waveform into an appropriate pulse signal.

Such waveform shaping is also performed on the outputs of the Hall elements H/ and I(, 2) by the waveform shaping circuit J5 and unit.

Therefore, in the control circuit shown in Figure 1, all subsequent operations are processed digitally, and are excellent in noise resistance, drift, etc.

The following FIG. 9 is a measured waveform diagram showing the relationship between energization timing and rotational unevenness using the digital phase control method of the present invention.

In this experiment, the deviation between the center of the encode output and the center of the control signal on the time axis t is defined as the energization timing θ, as shown at the bottom of the figure.

The occurrence of rotational unevenness was measured in the range of θ=-π to π.

As a result, this energization timing θ is different from θ and −π (=
It was confirmed that rotational unevenness is extremely small for π), and relatively high for θ=-"roya"ro.

This Figure 9 shows the relationship between the encode output and the control signal, but since the excitation pulse is generated by AND processing these two signals, the generation timing of the excitation pulse is the same as the generation timing of the control signal. I can think.

In the case of θ-O, where the incidence of rotational unevenness is low, the distribution circuit 3 is similar to the case explained in connection with FIG.
The mutual timing between the encode output and the control signal in the diagram is symmetrical in the drawing (temporally symmetrical) with respect to the center point of the encode output, which corresponds to the time when the rate of change of ke becomes zero. A control signal is input due to the following relationship.

Further, when θ=-π, four control signals are provided, but in this case as well, the control signals are input in a symmetrical relationship on the drawing with respect to the center point of the encoded output.

On the other hand, the occurrence rate of uneven rotation is particularly high in 1π, π, and
In this case, the correlation between the encoded output and the control signal is θ=
The relationship is not maintained like θ or 1π.

In other words, when the energization timing θ is close to O or -π, in other words, the control signal has timings that are temporally symmetrical around the point in time when the rate of change of the induced voltage ke becomes zero (the center point of the encoded output). When the input value is , the rotational unevenness will be significantly reduced, and if this relationship deviates, the rotational unevenness will gradually increase.

It can be seen that when θ=-1 or -, which is significantly deviated, rotational unevenness increases significantly and is affected by disturbance torque.

Effects Therefore, according to the digital phase control method of the brushless morph of the present invention, since all processing is performed using digital signals, it is excellent in noise resistance, drift, etc., and can be started by simply using a simple control means using only the phase control system. It is possible to perform high-precision rotation control with significantly less unevenness per rotation in high-speed, constant-speed rotations such as 70,000 rotations in drive systems where the characteristics do not pose much of a problem and torque fluctuations are relatively small. You can get an excellent wide effect.

[Brief explanation of drawings]

Fig. 1 is a partially cutaway plan view showing an example of the internal structure of a brushless motor, Fig. 2 is an example of a drive circuit, Fig. 3 is a block diagram showing a conventional servo control system, and Fig. 3 is a diagram showing the invention. Figure S is a block diagram showing an example of the configuration of a control circuit used when implementing the digital phase control method of a brushless motor. Fig. 6 is a time chart for explaining the relationship between the input timing of the control signal, the generation timing of the excitation pulse, and the torque fluctuation during steady rotation operation in the digital phase control method of the present invention, and Fig. 7 is used in the present invention. FIG. 3 is a time chart showing an example of a control signal correction method. FIG. 3 is a time chart showing the relationship between the detection signal and pulse signal detected by the speed detection section in the digital phase control method of the present invention. FIG. FIG. 3 is a measured waveform diagram showing the relationship between energization timing and rotation unevenness using the digital phase control method of the present invention. In the drawing, B is the speed detection section, /7 is the 72-stage frequency divider,
7g is a waveform shaping circuit, /9 is a phase comparator, 20 is a pulse width modulator, 2/ is a starting circuit, 2.2 is a /phase unipolar drive circuit 1.23 is a distribution circuit, 2g and 2shio are waveforms A shaping circuit is shown. Procedural amendment (1) Kazuo Wakasugi, Commissioner of the Japan Patent Office1, Indication of the case 1983 Patent Application No. 1283oo2, Name of the invention Digital phase control system for brushless motors 3, Person making the amendment Relationship to the case Patent applicant 1-3-6 Nakamagome, Ota-ku, Tokyo (67/l) Ricoh Co., Ltd. 4, Agent 6, Contents of amendment 1) Changed "Ttrolla IA" on page 8, line 11 of the specification to "ttrolla IA" ” he corrected. 2) Page 9, lines 10 to 13 are corrected as follows. When the voltage drop across the BL and M current detection resistors R3 is greater than the constant voltage (reference voltage) set in the startup circuit 21, the startup circuit 21 operates. At this time, the on (“H”) and off (“L”) signals of the activation circuit 21 are given to the distribution circuit 23. Correct the fluctuation 1 to Luk J to be the "torque generated by the motor."that's all

Claims (1)

[Claims] A brushless motor, a speed detection means, a phase comparison means for comparing the phase of an output from the speed detection means and a reference pulse, and a control signal modulated to a pulse width corresponding to the comparison result. a distribution circuit that generates excitation pulses through logical processing of a control signal output from the pulse width modulation means and a signal encoded with the output from the rotor position detection means; and this distribution circuit. Brushless morph 1 consisting of a drive circuit that applies a drive current to the windings of a brushless motor using excitation pulses from
The lj13 control circuit is provided with waveform shaping means for shaping the output from the speed detection means into a pulse, and waveform shaping means for shaping the output from the rotor position detection means into a nodle. The generation timing of each excitation pulse is set at 2% (however, n
is the number of motor phases), and the digital phase control method is characterized in that the output is symmetrical around the point in time when the rate of change becomes zero.