JPS60200791A - Speed controller of induction motor - Google Patents

Abstract

PURPOSE:To preferably control a vector without interference between the secondary magnetic flux and a torque current by limiting a torque component command value to a reference value or lower determined by the input voltage of an inverter. CONSTITUTION:A voltage detector 111 and a command value limiter 112 are provided to limit a torque voltage component command value (29a)Vqs* to a range that an exciting current in response to an exciting current component command value (231)Ids* can be always flowed to a motor 6. Thus, even if a power source voltage to be supplied to an inverter 5, i.e., the output voltage Vdc of a converter 2 decreases to the prescribed value or lower, the generated torque of the motor 6 is limited, but the control of noninterference between the secondary magnetic flux and the torque current of the motor 6 is maintained, no vibration occurs in the motor 6, but the torque in response to the limited torque current component command value (27a)iqs* is generated.

Description

DETAILED DESCRIPTION OF THE INVENTION [Technical Field of the Invention] The present invention relates to an improvement in a device for vector control of the speed of an induction motor.

[Prior art]

One type of control for induction motors is vector control in which the primary current of the induction motor is controlled as a vector quantity.

What are the various types of vector control force formulas? However, there is a method that linearizes the torque transfer function of an induction motor by considering the leakage inductance of the induction motor, which can obtain control performance equivalent to that of a DC motor even in a transient state. PowerElectr
onics Conference Collected Papers 19B3.
The engineering n-5 posture of Electric
Published by Cal Engineers of Japan,
H, Sugimoto outside "Theory and C"
haracteristi-cs of a New
Engineering Motor Drive! Yoy
stemHavingLinearTransfa
r Function”.

An example of applying this method to an elevator is shown in Figures 1 to 23.
This will be explained using figures.

In Fig. 1, R, S, and T are three-phase AC power supplies, and (2) is a converter (second figure),
(3) is a smoothing capacitor that smoothes the output of converter (2), (4) is a voltage detector consisting of a resistor connected to both ends of smoothing capacitor (3), and (5) is a voltage detector that is connected to both ends of smoothing capacitor (3). A PWM inverter (Fig. 2) is connected to a transistor and a diode, and generates variable voltage/variable frequency AC outputs (5a) to (5C) using a pulse width modulation (PWM) method, and (6) is an inverter ( 5
) three-phase induction motor for hoisting, driven by the output of
(7) is a speed detector that is directly connected to the electric motor (6) and emits a speed signal 9r (7a) proportional to the rotational speed of the electric motor (6);
(8) is the drive sheave of the hoist driven by the electric motor (6), (9) is the main rope wrapped around the sheave (8), and 00 is connected to one end of the main rope (9). elevator car, 0υ
is the counterweight 9 connected to the other end of the main rope (9), @~0
→ is a current transformer (IQ
are a sine wave signal (,:!1a) and a cosine wave signal (
21b) and input the current feedback signals (12a) to (14a).
) on the coordinate axis rotating in synchronization with the angular velocity ω of the secondary magnetic flux vector of the electric motor (6) (15a)
,' and the three-phase converter into torque current component signal (15b)/
Coordinate conversion circuits A (Fig. 11) and O consisting of two-phase conversion circuits
Q is a divider (e.g. ANALOG DEV, TCES
Co., Ltd., ADS33), (17) is a coefficient multiplication circuit (Fig. 13) that multiplies the input by a coefficient and generates a slip frequency signal (1'7a,), (18) is a speed signal (7a), and the gain p ( The forward rotation amplifier (corresponding to the number of pole pairs of the motor (6))
(Fig. 6), θ is an adder (Fig. 1r7) that adds the slip frequency signal (1'7a) and the output of the normal rotation amplifier 0 to generate a synchronous angular velocity signal (19a), and slope is the synchronous angular velocity signal (19
a) to obtain the phase angle signal of the secondary magnetic flux vector (ZOa
) (Fig. 19), (2υ is the phase angle signal "J
(2oa) and the corresponding sine wave signal (2oa) is input.
1a) and a function generator (
Fig. 14), (A) is a subtracter that subtracts the excitation current component signal (15a) from the excitation current component command value - and generates the deviation signal 'fc (Fig. 18), and (C) is composed of a delay/lead circuit. (24a) is the excitation voltage component command value, and (c) is the speed command value @ to the speed signal (Fig. 20). 7
(a) is a subtracter (Fig. 18) that subtracts and generates a deviation signal, and (a) is a speed control circuit (Fig. 20) that controls the output of the subtractor (which is composed of a delay/lead circuit) to zero. ), (ZVa) is the torque current component command value, @ is the torque current component signal (15
b) and a torsion current component control circuit (Fig. 20), which is configured with a delay-lead circuit and controls the output of the subtractor to zero (Fig. 20). ), (29a) is the torque voltage component command value, and (29a) is the excitation voltage component command value (24a) and torque voltage component command value (z9a) by inputting the sine wave signal (21a) and cosine wave signal (zlb). ) to the primary voltage command value (30a) for each of the three phases.
(30c) This is a coordinate conversion circuit B (FIG. 12) consisting of a two-phase/three-phase conversion circuit.

In Figure 2, (2 people) are power running converters connected to AC power supplies R, S, and T and forming a three-phase full-wave rectifier circuit by thyristors (2AA) to (2AF), (2AAg) to
(2AFg) is the gate of thyristor (2AA) to (2AF), (2B) is the gate of thyristor (2BA) to (2BF)
A three-phase full-wave rectifier circuit is formed by the AC power supplies R and S on the AC side.

A regenerative converter (2BAg) whose DC side is connected to the DC side of the power converter (2 people)
zBFg) is the gate of the thyristor (2BA) to (2BF), and the converter for power and regeneration (2A), (2B)
A converter (2) is configured. (4a) is the output of the voltage detector (4), and (5A) to (5F) are transistors that configure the inverter (5), two of each. Three sets of devices connected in series are connected in parallel. (5
Aa) ~ (5Fa) are transistors (5A) ~ (5F
), the bases (5G) to (5L) are diodes connected in parallel to transistors (5A) to (5F), respectively, and AC outputs ( 5a) to (5C) are emitted. The glue is the synchronous angular velocity signal (19a) and the voltage detector output (
4a), the firing signals (32a) to (3at) corresponding to the synchronous angular velocity signal (19a) are generated, and the firing signal (32
a) to (32f)f: Thyristor gate (2A
ignition signals (32i) to (3
2t) respectively in thyristor game) (2BAg) ~ (
2BFg), (c) inputs the -th' voltage command values (30a) to (30C) and sends base drive signals (3sa) to (3sf) to the transistor bases (5Aa) to (5Fa), respectively. Figures 3 to 7 show the configuration of the gate circuit 0, which is the base drive circuit that supplies J.
(Figure 5), (G) is the normal rotation amplifier (Figure 16), ⑼ is the operational amplifier, (36a) is its output, and (B) is the output (3) when the input reaches a positive predetermined value. A comparator that becomes rHJ (
(Fig. 21), Zeng outputs (3) when the input reaches a predetermined negative value.
Comparator where 8a) becomes rHJ (Fig. 22), R4-R4
is resistance.

'The heavy pressure detector output (4a) is the synchronous angular velocity signal (19a)
), that is, when the voltage of the smoothing capacitor (3) is lower than the voltage command value, the added value becomes negative, but it is inverted by the operational amplifier (to) and the output (36a) becomes positive. Therefore, the output (3'7a) of the comparator (b)
is "H", and the output (38a) of the comparator (c) is rLJ. In the opposite case, that is, when the voltage of the smoothing capacitor (3) is higher than the voltage command value, the added value becomes positive, but the output (36a) of the operational amplifier (g) becomes negative. Therefore, the output (3'7a) of the comparator (b) is rL
J, the output (3sa) of the comparator (c) becomes rHJ. This output (3'i'a).

(38a) is used to operate either the power converter (2 people) or the regeneration converter (2B).

In Figure 4, 0 is an inverting amplifier with a gain of -1 (Figure 15)
, (to), 0η are switching elements that become conductive when the input A becomes rI (J) (for example, HARRiS, HA
201) and (6) are operational amplifiers, (42a) is their output, R-R is a resistor, and C1 is a capacitor.

When the B signal (3'7a) is rHJ, the signal (36a)
is reversed and passes through the switch element -, and the signal (3Sa) becomes "
When the signal (36a) is "H", the signal (36a) passes through the switch element 0]) as it is. These signals are processed by an operational amplifier (6),
Gain and phase compensation is performed by resistors R5 to R and capacitor C1.

In Figure 5, (51) is a transformer, (52) is a transformer (51)
) is connected to the secondary side of the bridge-connected rectifier circuit, (
53) is a Zener diode connected to the DC side of the rectifier circuit (52), (54) is a capacitor, (55) is an inverting amplifier with a gain of -1 (Figure 15), (56) and (57) are Operational amplifier, (5B) and (59) are diodes to limit the negative side voltage, (60) and (61) are transistors, (60a) and (60b) are their collector outputs, R10""'21 is a resistor, +V is a semiconductor positive power supply, and -V is a negative power supply. Note that the gate circuit 02 of FIG. 2 is provided with three sets of the circuits of FIG. 5 for R-phase to T-phase.

Transformer (51), rectifier (52), Zener diode (
53).

The circuit composed of the capacitor (54) and resistors R1o and R11 is the thyristor (2AA) to (2AF) in Figure 2.
, (2BA) to (2BF) is a circuit that generates a power supply synchronous voltage for controlling the firing angle. For example, by applying R-phase and T-phase line voltages to the transformer (51), R-phase thyristors (2, AA), (2AD), (2
A synchronous voltage is obtained to control the firing angles of BA, ) and (2BD). The rectifier circuit (52), the Zener diode (53), and the capacitor (54) generate a nearly triangular voltage, which serves as a reference value and is used as a reference value for the operational amplifier (54).
6) , supplied to (5-1).

An operational amplifier (56), a circuit consisting of resistors R12, R13, and a negative power supply -■, and an operational amplifier (57),
A circuit consisting of a resistor R14゜RI5 and a positive power supply +■ has a biased voltage connected to an operational amplifier (56
) and (57), each comparator has a hysteresis width. Therefore, for example, if the signal (42a) has a positive value and exceeds the predetermined value of the triangular voltage, the output of the operational amplifier (56) will be r
Becomes HJ. At this time, the output of the operational amplifier (5'7) is "L". Conversely, when the signal (42a) has a negative value and exceeds the predetermined value of the triangular voltage, the operational amplifier (42a)
The output of 57) becomes rHJ. At this time, the operational amplifier (
The output of 56) is rLJ. Operational amplifier (56
) becomes [J, the transistor (60) is turned on and the output (6Oa) reaches zero voltage. - Since the transistor (61) is non-conducting, the output (61a) becomes a positive voltage.

In Figures 6 and 7, (63) to (70) are diodes, (71) to (14) I''i pulse transformers, (75) to
(78) is a capacitor, (79) to (82) are transistors, and R2□ to R31 are resistors. Note that although the figure shows only the R phase, the S phase and T phase are also configured in the same way.

When the signal (3'i'a) is rHJ, that is, the voltage of the smoothing capacitor (3) is lower than the voltage command value (during power running), the transistors (79) and (eo) are conductive, and the pulse transformer (71) is turned on. ), a positive voltage is applied to one end of the negative side of (72). And transistor output (6oa
) becomes zero voltage, current flows through the - secondary side of the pulse transformer ('y1) and the diode (63), so a pulse voltage is generated on the secondary side, and the thyristor (2AA)
is conductive.

At this time, since the transistor output (61a) has a positive voltage, no current flows to the negative side of the pulse transformer (72), no pulse voltage is generated on the secondary side, and the thyristor (2AD) is not conductive. In this way, the row converters (two) operate and increase the voltage of the smoothing capacitor (3). Further, when the signal (38a) is rHJ, the pulse transformer (73) or the pulse transformer (74) is operated by the transistor outputs (60a) and (61a), and the thyristor (2BA) or the thyristor (2BD) is operated.
) conducts. In this way, the regeneration converter (2
B) operates and reduces the voltage across the smoothing capacitor (3).

FIGS. 8 and 9 show the configuration of the base drive circuit. In FIG. 8, (84) is a triangular wave signal (s4a)' with a constant frequency sufficiently higher than the frequency of the AC power supplies R, S, and T. il-
Oscillating triangular wave generator (Figure 9), (85A) to (B1
0) compares input A and input A2, and input A1≧input A2
When the output is rHJ, the input A is a comparator (Fig. 23) where the output is rLJ when the input is A2, (86A)
~(86C) is a two-phase distributor, (86AA) ~(86AC
) is NOT gate, (86AD) is resistor, (B6AE)
is a capacitor, (86AF) and (86AG) are AN
This is the D gate.

The comparator ('55A) compares the -th voltage command value (3oa) and the triangular wave signal (84 & ), and calculates the -th voltage command value 130a.
) is greater than the triangular wave signal (84a), the output is rHJ
Therefore, the output (85Aa) of the comparator (85A) has a waveform as shown in FIG. NOT gate (86A
By the operation of A) to (86AC), when the output (s5Aa) is rHJ, the output (33a) of the AND gate (s6AF)
) becomes rnJ, and the output (33d) of the AND gate (86AG) becomes rLJ. Still, when the output (85Aa) is rLJ, the output (86ALi') of AND game)
33a) becomes rLJ, and A N D game) (86
The output (33d) of AG) becomes [J. That is, the transistors (5A) and (5D) of the inverter (5) are made conductive alternately. The same goes for the two-phase dividers (8eB) and (sac), and the outputs (3sb) and (33e) alternately connect transistors (5b) and (5E)
33c), (33f) The transistor (5C
) and (5F) are made conductive alternately. In this way, a voltage in which a sine wave is modulated into a triangular wave is applied to the motor (6).

In Figure 9, (84A) is an AC power supply that emits a sine wave AC with a constant frequency that is sufficiently higher than the frequency of AC power supplies R, S, and 'l', (84B) and (84C) are Zener diodes, and (84D). ) is a capacitor, (84F), (84F
) is resistance.

The maximum value of the sine wave alternating current of the alternating current power supply (84A) is limited by the zener diodes (s4B) and (sac).

This is delayed by a delay circuit with a large time constant consisting of a capacitor (84D) and a resistor (84E), thereby obtaining a triangular wave signal (84a).

FIG. 11 shows the configuration of the coordinate conversion circuit A. In the figure, the inverting amplifiers (FIG. 16) and (SSC) to (88E) are inverting amplifiers (15th and 88E) with gains of 11, -1, and -τ, respectively. (Fig. 17), (89A) to (89C) are adders (Fig. 17),
(89D) is the subtractor (Fig. 18), (9OA) ~ (9
0D) is a multiplier (for example, ANALOG DEVICE
Company S, AD533).

Excitation current component value 9 (f5a) and torque current component signal (
x5b) and the current feedback value 'i3' (12
The following relationship exists between a) to (14a).

Here, ids e excitation current component (15a)'qa +
+) Luk current component (15b) tag~iw: electric motor - next'
Ichiryu (12a) to (14a) coordinate conversion circuit A is for calculating this.

FIG. 12 shows the configuration of an example of coordinate conversion circuit B, in which (9
1A) to (91D) are multipliers similar to the multiplier (9oA), (92A) and (92B) are subtracters (Fig. 18), (
92C).

(92D) is an adder (Fig. 1'7), (94A), (9
4B) is an inverting amplifier (FIG. 15) with a gain of m-.

The excitation voltage component command value (24a), the torque voltage component command value (29&), and the -th voltage command value (30a) to (30
0) has the following relationship.

Here, ■Second to V: Secondary voltage command value (30a) to (3
0c) ■:, : Excitation voltage component command value (24a) VZ, : To#ri7! The pressure component command value (29a) in the coordinate conversion circuit B) is used to calculate this.

FIG. 13 shows the configuration of the coefficient multiplication circuit μ, in which (9
6) and (97) are operational amplifiers, and R35 to R40 are resistors (R3B=R39).

The slip frequency signal (1'7a) pω6 is calculated as follows.

Here, p: number of pole pairs of the electric motor (6) ωa: slip angle frequency tic (17a) Rr secondary resistance value of the secondary electric motor 6) Lr: secondary inductance value of the electric motor (6), that is,
The input from the divider @ is multiplied by Rr/Lr and inverted to become a negative value. Furthermore, this is inverted by the operational amplifier (97) and becomes a positive value, resulting in a slip frequency signal "(1'7
a) is output.

Figure 14 shows the configuration of the function generator ■υ, and in the figure (98
) is an A/D converter (for example, BURRBRO) that inputs the phase angle signal (ZOa) and converts it into a digital value.
WN Co., ADC80) and (99) are cosine ROMs (for example, UNTIL Co., 12716) in which the value of coF3g for each phase angle θ is stored as a digital value; The sine RO
M. (101).

(102,) is D that converts digital value to analog value
/A converter (for example, BURRBROWN, DA, C
80).

The value of cos θ for the phase angle θ represented by the phase angle signal (ZOa) is obtained from cosine ROM (99) and
The values of θ are respectively read from the sine ROM (100) and converted into analog values by the D/A converters (101) and (102) to become a cosine wave signal (21b) and a sine wave signal (21a).

The configurations of other elements are shown in FIGS. 15 to 23. In the figure, A, A, , A2... are inputs, B is output, and P.

P, , P2... are operational amplifiers, R,, R2...,
1,2r r -. - is a resistor, C is a capacitor, D is a diode, and Z is a Zener diode.

FIG. 15 shows an inverting amplifier.

2 B=--A R.

Therefore, if R1-R2, then B=-A.

FIG. 16 shows a normal rotation amplifier.

FIG. 17 shows an adder.

B=A, +A2+A3... becomes.

FIG. 18 shows a subtracter.

B-/42- AI becomes.

FIG. 19 shows an integrator.

FIG. 20 shows a delay/lead circuit.

Therefore, R,=R2,R2C=T,,(R2+R2
□) C=T2 FIGS. 21 and 22 show comparators, and bias voltages determined by resistors R2 and R4 are applied to the operational amplifier P, respectively. Assuming that this value is e, in FIG. 21, when A≧e, and in FIG. 22, when A≦−〇, the outputs become B and FH, respectively.

FIG. 23 is also a comparator, and when A1 and A2, the output B becomes [J, and when A, (A, .), the output B becomes rLJ.

First, an outline of vector control will be described according to FIG.

The equation of state for an induction motor is d-q rotating with an angular velocity ω
In the (excitation component-torque component) coordinate system, the d-axis and q-axis components of the -order flow and the secondary current d
If the axis and q-axis components idr l'qr are respectively state variables and the d-axis and q-axis components Vd, .

Here, R: Primary resistance of the induction motor Rr: Secondary resistance L8 as above Secondary inductance Lr: Secondary inductance as above M: Mutual inductance between the secondary winding and secondary winding p: Number of counter poles as above ωr: Actual angular velocity of the rotor P=d/clt: Differential operator σ: Leakage coefficient 2 expressed by the formula 2 L, L, °°°■ Similarly, the generated torque T of the induction motor. is expressed by the following formula.

...■ However, λdrrλ9r represent the d-axis and q-axis components of the secondary magnetic flux, respectively, and are expressed by the following equation 0.

As can be seen from equations 0 and 2, equation 0 is nonlinear in that it includes the angular velocity ω of the secondary current vector and the rotor angular velocity ω1 in the state matrix. In addition, equation (2) is nonlinear in that it includes the product of two state variables. Therefore, if this continues, it will be difficult to achieve good speed control.

By the way, the principle of vector control is that the primary current to be supplied to the induction motor is regarded as a vector quantity on the coordinate axes (d-q axes) that rotate in synchronization with the secondary magnetic flux vector, and this -
By decomposing the secondary flux vector into a component parallel to the secondary magnetic flux vector (i.e., excitation current component) and a component perpendicular to the secondary magnetic flux vector (i.e., torque current component), and controlling each independently, the secondary The objective is to perform non-interfering control of magnetic flux and torque. When controlling the excitation to a constant value, that is, when id, = d, 8 (constant value), the above objective is achieved by controlling the d-axis component 1dr of the secondary current vector to be zero. be done. In other words, 'de "■ds (constant value) ...■id,=O・
...■ Under the conditions, the equation of state of the induction motor, equation 0, and the generated torque, equation 2, are each linearized as follows.

...■ "' ...■ To knee pM1dII'qr""p mutual deiqs Also, at this time, λdr=M6...λ9r-〇, and the secondary magnetic flux rotates in synchronization with the d-axis Becomes a vector.

By the way, the conditions of Equations 0 and 0 are as follows: The primary current 1ds + 'qB & rotor angular velocity ω of the induction motor is the control amount, and the angular velocity ω of the secondary magnetic flux vector and the -order voltage d-axis component "'!++ , are satisfied by controlling according to the 0 equation and the [phase] equation, respectively.

Vd,, = R, ■: 6-ωσLi 10K (engine: n −i
ds) ... [Phase] qe However, pω6 in the formula 0 represents the slip frequency, and ω:8 in the formula 0 represents the excitation current command value (constant value).

Next, we will explain the specific operation of the vector control described above in Figures 1 to 2.
This will be explained with reference to Figure 3. This example uses a pwu inverter (5
) is used to control the excitation of the induction motor (6) to a constant value. However, the first and second terms on the right side of the above [phase] equation are omitted because the gain K of the third term is sufficiently large.

The excitation current component command value @engine da is input to the subtracter (a), and the excitation current component signal ' (15a) 1d@, which is emitted from the coordinate conversion circuit A (15), is actually the motor (6
) and the deviation is output. This deviation becomes the excitation voltage component command value (24a) V: through the excitation current control circuit (c), and is input to the coordinate conversion circuit B).

The speed command value (e) ω: is input to the subtracter (a) and compared with the speed signal (7a) ω1 from the speed detector (7),
The deviation is output. This deviation becomes the torque current component command value (27a) via the speed control circuit and is input to the subtracter (8). Here, coordinate transformation circuit A (1
Torque current component signal (15b) emitted from e,
.

The deviation is outputted and sent to the torque voltage component command value (29a) V* to the torque current component control circuit).
and is input to the coordinate conversion circuit B(1).

6 - F, divider 0, and coefficient multiplier circuit a output a slip frequency signal (1'7a) pω8 according to the second term of the above equation 0, which is outputted to adder 4 as a speed signal (7a) ω. , is the induced lightning obtained by multiplying the polar logarithm by p! It is added to the electrical angular frequency pω of the IJ machine (6) to form a synchronous angular velocity signal (19a,) of the secondary magnetic flux vector. This is integrated by an integrator (a) to produce a phase angle signal (20a) θ of the secondary magnetic flux vector. A sine wave signal (21a) and a cosine wave signal (zlb) corresponding to this phase angle are extracted by the phase angle generator Q) and input to the coordinate conversion circuit B).

coordinate conversion circuit B) are manually operated (24a), (29a), (
21a).

(21b) ft converted to primary voltage command value (30a)
~ (30c) is output to operate the base drive circuit 03 of the PWM inverter (5), and the base drive signal (33a) is output.
~(:ysf) is given to the inverter (5), and the well-known PW
M control is performed. On the other hand, the synchronous angular velocity signal -5+(19a
) is input to the gate turn @0→. The gate circuit (a) is the voltage detector output (4a) and the synchronous angular velocity signal (19a)
, it detects whether the electric motor (6) is running in power mode, and when it is running in power mode, it gives ignition signals (32a) to (s2f) to the converter for power mode (2 people), and when it is in regenerative mode, it gives ignition signals (32a) to (s2f) to the converter for power mode (2 people). Signal (3
2g) to (3Zt) are given to the regeneration converter (2B). As a result, the voltage across the normal temperature capacitor (3) is changed, and well-known PAM (Pulse Amplitude Modulation) control is performed. Then, the AC output (5a) from the inverter (5)
) to (5C) are detected by current transformers (2) to 0, and current feedback values g(12a) to (14a) are detected by coordinate conversion circuit A (1
51.

In this way, the exciting current component signal (15a) i of the primary current, which is a direct current, and the torque current component signal (45b)
) 19 and 19, each has an independent negative feedback control system, and has a predetermined excitation current component command value (a) d*, a speed command value (a) ω: and a speed signal (7a) ω, When a torque current component command value (2'7a) corresponding to the deviation is given, each current control system performs appropriate gain compensation and phase compensation for the current deviation to generate The excitation voltage component command value (24a) V6 and the torque voltage component command value (29a) V: are used as manipulated variables to control the -primary current vector of the motor (6) to match the primary current reference vector. This enables highly accurate speed control with excellent responsiveness. In addition, the slip frequency signal (1'7
a) pω is calculated using the -order current feedback signal of the motor (6), and according to this, the instantaneous value of the -order voltage of the motor (6) is given, so that it is transiently equivalent to a DC machine. This enables highly accurate vector control that satisfies high performance. Therefore, even in elevator control where emphasis is placed on transient response, it is possible to achieve suitable control performance.

It is assumed that the excitation current component in response to the current flow reliably flows through the motor 6). Therefore, there is plenty of AC power.

Due to a cause such as a voltage drop in S and T, the output voltage of the converter (2) decreases, causing the motor (6) to change the excitation current command value;
If the excitation current component corresponding to 8 cannot be caused to flow, the vector control condition expressed by equation 0 collapses.

As a result, it has been found through simulations that vibrations occur in the torque generated by the electric motor (6), which significantly impairs the riding comfort of the 00.

[Summary of the invention]

This invention aims to improve the above-mentioned problems by limiting the torque component command value according to the AC power supply voltage or the input voltage of the inverter9. It is an object of the present invention to provide a speed control device for an induction motor in which vector control is reliably performed by components following their command values, and stable motor torque can be generated.

[Embodiments of the invention]

An embodiment in which the present invention is applied to elevator speed control will be described below with reference to FIGS. 24 and 25.

In Fig. 24, (111) is a voltage detector (Fig. 25) that detects the output voltage of converter (2) (input voltage of inverter (5)) ■4° and emits the corresponding voltage signal (lla). , (112) is the torque voltage component command value (
29a) ■: Command value limiting circuit that limits 8 (Fig. 25)
(UNI2a) is its output.

In Figure 25, the voltage detector (11 units) is the resistor (1 unit lA)
~(llIc) and an isolation amplifier (I
IID), the output voltage of converter (2) ■,
. After dividing by a predetermined voltage division ratio, the output voltage ■,. A positive voltage signal (xlla) corresponding to f,'fk. The command value limiting circuit (112) includes a negative power source -■ and a resistor (11
2A), (112B), a negative predetermined voltage ■5 setting circuit, an operational amplifier (112c), and a diode (l
12D) and resistors (112F) to (112H), an operational amplifier (112I) and a resistor (Uni 2
J)~(112x,), operational amplifier (112H~(112Q), s diode (11
2R), (112S) and a resistor (112T).

Next, the operation of this embodiment will be explained.

Converter output voltage■,. is a voltage signal (1, 11a) corresponding to the output voltage ■4° by the voltage detector (111).
is converted to This voltage signal (1x1a) is subtracted by a predetermined negative bias voltage (2) in the subtracting circuit, and the sign is inverted and the gain is adjusted, and the operational amplifier (112C) generates a negative limiting reference voltage. Therefore, the output of the operational amplifier (112) becomes a positive limiting reference voltage. Note that the operational amplifiers (1xz+a) and (112N) function as comparators, and the operational amplifiers (112P) and (112Q) function as buffer amplifiers.

Now, if the torque voltage component command value (29a) V'' is positive a and is lower than the above limit reference voltage, the voltage applied to the inverting input terminal (-) of the operational amplifier (112M) is the non-inverting input. Since it is lower than the positive limit reference voltage of the terminal (+), the operational amplifier (112M) is set to the command value (29a).
vo Yoji also outputs a high positive saturation voltage. Therefore, the diode (112R) is in a reverse blocking state,
The operational amplifier (112M) cannot conduct any output current. - On the other hand, since the positive voltage applied to the inverting input terminal (-) of the operational amplifier (112N) is higher than the negative limit reference voltage of the non-inverting input terminal (+), the operational amplifier (x12m) is in negative saturation. Outputting voltage. Therefore, the diode (xlz
s) is also reverse blocked, and the operational amplifier (112N) cannot conduct current. Therefore, the command value (29a) V”s is not affected by these comparators, and the operational amplifier (
1U2Q), and outputs (112a)v" as it is.
□G. That is, V;g-V9:. The same applies when the torque voltage component command value (29a) V'' is negative and is lower than the limit reference voltage by m.

Next, assume that the command value (29a) v* is positive and attempts to exceed the limit reference voltage of 9 units. Since the positive voltage applied to the inverting input terminal (-) of the operational amplifier (UNI 2M) tends to become higher than the positive limiting reference voltage applied to the non-inverting input terminal (+), the operational amplifier (UNI 2M) l121A) attempts to output a negative saturation voltage, the diode (l12R) is forward biased, and the resistor (uni2T) and diode (11
2R)'! i--and sinks current from the operational amplifier (112P). As a result, a voltage drop occurs due to the resistor (112T), and the voltage applied to the non-inverting input terminal (+) of the operational amplifier (112Q) decreases to the limit reference voltage. Therefore, the output (xx
2a) V ”l is suppressed below the limit B reference voltage.Similarly, the command value (2 wings)■
0 is negative, and if an attempt is made to exceed the limit reference voltage, the negative voltage applied to the inverting input terminal (-) of the operational amplifier (112N) will become the negative voltage applied to the non-inverting input terminal (+). Since it tries to be larger than the limiting reference voltage, the operational amplifier (112N) outputs a positive saturation voltage, the diode (1128) is forward biased, and the diode (n2
S) and the operational amplifier (112P) through the resistor (112T)
). As a result, an operational amplifier (l12Q
The voltage applied to the non-inverting input terminal (+)K of ) is the output of the operational amplifier (112P), that is, the negative command value (29a)
V'', the negative market voltage becomes smaller due to the heavy pressure drop that occurs across the resistor (112T) due to the above current, and the operational amplifier (112N
) is suppressed below the negative limit reference voltage at which O is balanced. In this way, the torque current component command (W, (2'7
a) Torque voltage component command value (2:9a) V corresponding to V
” is q8 9θ Always limited to the range in which the excitation current according to the excitation current component command value (a) ■; That is, the output voltage V of the converter (2) is the AC power supply R, S.

Even if the voltage of T drops below a predetermined value due to a drop in the voltage of T, etc., the generated torque of the electric motor (6) will be limited;
Non-interfering control of the secondary magnetic flux and torque current of the electric motor (6) is maintained, and the electric motor (6) is controlled to a limited torque current component command value (27a) i; without causing vibration.

It will continue to generate torque according to the This is a great advantage, especially when riding comfort in the car QO is important, such as in an elevator.

26 and 27 show another embodiment of this invention.

FIG. 26 is the same as FIG. 24 except that the command value limiting circuit (112) shown in FIG. 27 is used, and the limiting reference voltage is the output voltage of the voltage detector (ill). The excitation current component command value is controlled by 8.

FIG. 27 is the same as FIG. 25 except that a divider (112U) is inserted between the subtraction circuit and sign inversion circuit of FIG. 25, and the output voltage of the voltage detector (111) is .

The ratio of the excitation current component command value (a) and the excitation current component command value (a); The operation is similar to that in FIG. 25.

This example is suitable for variable control of the excitation current component command value (A) ■: 8. For example, in order to make the command value (E): 8 equivalent to 1 field weakening in a DC machine. When the value is set to a small value, the limit reference electric current component is set to a value relatively higher than the normal level, so that the limit on the torque current component is relaxed.

In this way, even when variably controlling the magnetic flux, the limiting reference voltage is appropriately lowered and the limiting reference voltage is controlled to the optimum limiting reference voltage without prolonging the time it takes for the acceleration/deceleration of car ao to reach an appropriate value. Note that in each of the above embodiments, the torque voltage component command value (29
a) Although the one that limits V'' is shown, it is possible to give B a similar function by limiting the torque qI+current component command value (27a) i''.

Further, in each of the above embodiments, the excitation current component and the torque current component are controlled respectively, but the present invention is also applicable to a structure in which the secondary magnetic flux and the torque current component are controlled respectively. In this case, the magnetic flux command value λdr is used instead of the excitation current component command value (c) in FIG. 26, and the excitation current component signal (15a) is set to the 0 using the d-axis components λ, r of the secondary magnetic flux obtained through the first-order lag circuit with the following time constant, and the subtractor (A) that calculates the deviation between the magnetic flux command value λdr and the d-axis components λ, , lr of the secondary magnetic flux. All you have to do is make it work.

〔Effect of the invention〕

As described above, in this invention, AC power supply voltage is converted into variable voltage/variable frequency AC power via a converter and an inverter, and the converted AC power is supplied to an induction motor, and torque component command value and magnetic flux command value or excitation current component command value are The speed of the electric motor is controlled in accordance with the above, and the torque component command value is limited to below a reference value determined by the AC power supply voltage or the input voltage of the inverter.

As a result, even if the voltage supplied to the inverter decreases for some reason, it is possible to perform good vector control and generate stable motor torque without causing interference between the secondary magnetic flux and the torque current.

[Brief explanation of drawings]

Fig. 1 is a block circuit diagram showing a conventional induction motor speed control device, Fig. 2 is a circuit diagram of the converter and PWM inverter portion of Fig. 1, and Figs. 3 to 7 are circuit diagrams of the gate circuit of Fig. 2. Circuit diagram, Figure 8 is a block circuit diagram of the base drive circuit in Figure 2, Figure 9 is a circuit diagram of the triangular wave generator in Figure 8,
10 is an explanatory diagram of the operation of FIG. 8, FIG. 11 is a block circuit diagram of coordinate conversion circuit A in FIG. 1, FIG. 12 is a block circuit diagram of coordinate conversion circuit B in FIG. 1, and FIG. FIG. 14 is a block circuit diagram of the function generator shown in FIG. 1, and FIGS. 15 to 23 are circuit diagrams of other elements. Fig. 16 shows a normal rotation amplifier, Fig. 17 shows an adder, Fig. 18 shows a subtracter, Fig. 19 shows an integrator, Fig. 20 shows a delay/lead circuit, Figs. 21 to 23 show a comparator, and Fig. 24 25 is a block circuit diagram showing an embodiment of the speed control device for an induction motor according to the present invention, and FIG.
26 is a block circuit diagram showing another embodiment of the present invention; FIG. 27 is a circuit diagram of the voltage detector and command value limiting circuit shown in FIG.
This figure is a circuit diagram of the command value limiting circuit of FIG. 26. In the figure, R, S, and T are three-phase AC power supplies, and (2) is a
(5) is a PWM inverter, (6) is a three-phase induction motor, (7) is a speed detector, and QQ is a coordinate conversion circuit A1 (
A) is a subtracter, (C) is an exciting current component control circuit, (C)
is a subtractor, (b) is a speed control circuit, (a) is a subtractor, (
4) is a torsion current component control circuit, (G) is a coordinate conversion circuit B, (111) is a voltage detector, and (112) is a command value limiting circuit. In addition, the same reference numerals in the figures indicate the same or equivalent parts. 21 ga zIl Fig. 1-2 0 Fig. 13 Fig. 14 Fig. I Fig. 15 Fig. 16 Fig. 17 Fig. 18 Fig. 19 Fig. 70 Fig. 21 Fig. 22 Fig. 23 Fig. 25 m L-- ,,.゛/12 Procedural amendment (voluntary) 1. Display of the incident! ly Gansho 59-57876 No. 2,
Title of the Invention Relationship between the speed control device 3 for an induction motor and the person making the correction f'1 Patent Applicant Address Taku Tochiyo "11-ku Hinouchi 2-chome 2-3 C)" Name ( 60],) Nisubishi Electric Co., Ltd. representative +
1. ．． +Jinhachi Department 4, Agent 5, Subject of amendment (1) Detailed explanation of the invention in the specification (2) Figure 9 of the drawings 6, Contents of amendment (1) rADS33J in the second line of page 4 of the specification Corrected it to rAD533J. 1 (2) On page 18, line 10 of the same page, replace rB-iJ with “B
=N” and corrected. [3] In the second line of page 22, the phrase ``Example of speed'' should be corrected to ``Velocity ω,''. (4) In the drawings, Figure 9 will be corrected as shown in the attached appendix. . 7. One or more drawings showing Figure 9 after the revised list of attached documents

Claims (1)

[Claims] (1) Convert AC power supply voltage to DC via a converter, convert it to variable voltage/variable frequency AC power via an inverter, supply it to the car drive induction motor, and generate the torque component command value and magnetic flux command. A voltage detector that detects the AC power supply voltage or the input voltage of the inverter, and an output of the voltage detector that detects the torque component command value. 1. A speed control device for an induction motor, comprising a command value limiting circuit that limits the command value to below a reference value determined by the following.