JPS596454B2 - Proximity switch oscillation circuit - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to an oscillation circuit for a high frequency oscillation/loss detection type proximity switch.

High-frequency oscillation/loss detection type proximity switches utilize the fact that the high-frequency loss of the detection coil increases when a nearby object such as a metal object comes close to the detection coil. The detection operation is performed by attenuating the amplitude of this oscillation circuit or stopping oscillation due to the approach of a nearby object.

This high-frequency oscillation/loss detection type oscillator circuit (1) is capable of oscillating over a wide frequency range, (2) the oscillation amplitude changes sharply in response to changes in the loss of the detection coil, and (3) the detection coil is as long as possible. 2) The oscillation frequency can be easily determined; and 4) The loss value of the detection coil at which the oscillation width begins to attenuate or oscillation stops can be easily determined. It is desired that the oscillation width (both ends of the detection coil) be as large as possible.

From the viewpoint of satisfying each of these points to some extent and having a simpler circuit configuration, the circuit shown in FIG. 1 has been proposed as a circuit configuration suitable for IC implementation.

To explain this circuit, the transistor Trlld
The emitter follower is connected, and its output current is ■.

is transistor Tr2. A feedback current is fed back as 1F through a current mirror circuit made up of Tr3.

This feedback current IF is supplied to a resonant circuit consisting of a detection coil L and a capacitor C.

The detection coil L has two terminals, and one end is connected to the OV side of the power source.

If an electric anchor vT is generated in this parallel resonant circuit (tank circuit), the emitter follower circuit constituted by the transistor Tr1 causes the collector of the transistor Tr1 to have approximately VTJ.

(Re is the value of the resistor R8 connected to the emitter of the transistor Trl).

flows. This current■.

A current ■F approximately equal to .times.F flows into the collector of the transistor Tr3 by the action of the current mirror circuit and is supplied to the parallel resonant circuit.

Current fed back to parallel resonant circuit via current mirror circuit■
Since F can be adjusted by the value of resistor R8, the magnitude of the feedback current can be varied by resistor R8.

That is, the strength of oscillation can be varied by changing the resistance Re.

Also, the oscillation frequency is determined by the resonant frequency of the parallel resonant circuit, ignoring the phase delay in the emitter follower circuit and the current mirror circuit, and is determined by the four inductances of the detection coil L and the capacitor C.
determined by the capacity of

The circuit shown in FIG. 1 has an extremely simple circuit configuration and is particularly suitable for use as an IC as an oscillation circuit for a proximity switch.

However, there are drawbacks as described below.

The base DC bias of transistor Tr1 is low, so the base input voltage vT and collector current ■.

Even if the relationship with is not linear and vT is sinusoidal, Ic
1d distorted waveform, and as a result, the feedback current ■F also becomes a dramatically distorted waveform.

When the feedback current IF has a distorted waveform in this way, the only component that can excite the parallel resonant circuit is the fundamental wave component, so the ability of the oscillation circuit is evaluated by the magnitude of the fundamental wave component IFI of IF with respect to VT. There must be.

In other words, the negative conductance gosc of the active circuit viewed from both ends of the parallel resonant circuit is gosc=IF'1/VT=Io]/'VT (where I
.

1 is expressed as the fundamental wave component of the collector current Ic of the transistor Tr1.

If the value of this negative conductance gO8e remains constant with respect to the magnitude of the amplitude VT, the change in the oscillation amplitude will be steep with respect to the change in the loss of the detection coil L.

If the base DC bias of transistor Tr1 is inappropriate, the distortion component will increase according to the amplitude VT, and the fundamental wave component will decrease, so gosc will depend on the magnitude of the amplitude VT, resulting in a sharp oscillation width. I can't get any change.

FIG. 2 shows data showing the results of investigating how the characteristics of gosc with respect to the amplitude vT change when a direct current bias voltage Eb is applied to the amplitude VT.

When the bias voltage Eb is 0, gos c is extremely non-linear, and even if the bias voltage Eb is applied to compensate for the base-emitter voltage VBE of the transistor Trl, it is still insufficient and non-linear. characteristics have been shown to appear.

From FIG. 2, it can be seen that in order to eliminate the dependence of gosc on the voltage VT and obtain a linear characteristic, a bias voltage Eb that is at least equal to the total oscillation width vT is required.

In view of the above, an object of the present invention is to provide an oscillation circuit for a proximity switch that is improved so that the oscillation amplitude changes sharply with respect to a change in the loss of the detection coil.

That is, according to the present invention, the above object is achieved by inserting a level shift circuit between the column resonant circuit and the base of the transistor Tr1.

This level shift circuit needs to be inserted in series with the parallel resonant circuit on the base side of the transistor Tr1 in order to shift a constant DC level with respect to the amplitude vT generated at both ends of the parallel resonant circuit.

Furthermore, this level shift circuit must not add any extra impedance to the parallel resonant circuit (as seen from the parallel resonant circuit), and it is also necessary that the shift level does not vary with the amplitude vT. be.

Each example will be described below.

FIG. 3 shows the first embodiment, in which a resistor RB is inserted between one end of the parallel resonant circuit and the base of the transistor Tr4. The current {circle around (2)}B of the constant current circuit is made to flow through this resistor RB via a current mirror circuit consisting of Tr5.

A voltage drop of IBXRB occurs in this resistor RB, and the DC level is shifted by this voltage drop.

Here, transistor Tr4. The reason for providing the current mirror circuit consisting of Tr5 is as follows.

In order to increase the amplitude vT of the parallel resonant circuit, it is necessary to maintain a wide dynamic range of the transistor Tr1.

Therefore, the base bias voltage of the transistor Trl needs to be high.

If the amplitude VT changes significantly, it will have an adverse effect on the constant current circuit and cause a current drift, so a current mirror circuit was provided to eliminate this.

FIG. 4 shows a second embodiment. In this figure, an energy diode zD is inserted between one end of the parallel resonant circuit and the base of the transistor Trl, and a transistor Tr4. Tr5
The current IB from the constant current circuit via the current mirror circuit consisting of
is supplied to obtain a shift voltage of the Zener voltage vz.

FIG. 5 shows a third embodiment.

In this figure, one end of the parallel resonant circuit and the transistor T
A plurality of (n) Darlington-connected transistors T r 6 + T r 7 between the base of rl
+T r 8m... is inserted and a transistor T is inserted into this.
r4. The current ■B from the constant current circuit is supplied through a current mirror circuit consisting of Tr5.

Thus, in the Darlington circuit nXVBE(VBE
obtains the shift voltage of the base-emitter path of one transistor.

Note that in order to obtain the desired shift voltage by connecting four diodes in series when there is no suitable Zener diode, use the sixth diode.
As shown in the fourth embodiment shown in the figure, each base and collector may be short-circuited and connected in series.
A problem arises when the

That is, when an IC is constructed by integrating planar bipolar transistors, a parasitic diode is generated between the collector of the NPN transistor and the P substrate.

The parasitic diode Ds of the transistor Tr8 closest to the parallel resonant circuit is shown in FIG. As shown by the front line, it is connected between the collector of the transistor Tr8 and the power supply Ov.

Therefore, in the negative half cycle of the parallel resonant circuit, both sides of the parallel resonant circuit are short-circuited by this diode DS and the base-emitter path of the transistor TrB.
It will be clamped at a certain voltage.

Therefore, even if the base bias voltage of the transistor Tr1 is increased to widen the dynamic range of the base voltage of the transistor Trl, the oscillation width VT will not become large due to clamping on the parallel resonant circuit side.

Therefore, when applying a desired shift voltage using a large number of transistors, the circuit configuration shown in FIG. 5 is superior.

FIG. 1 is a circuit diagram showing a fifth embodiment.

In place of the Zener diode zD shown in FIG. 4, a constant voltage circuit consisting of a transistor Tr6 and resistors R1 and R2 is inserted.

FIG. 8 shows a sixth embodiment of the transistor Tr6. T
A constant voltage circuit is constituted by r7 and resistor RH+R2.

In either case of FIG. 7 or FIG. 8, when the voltage generated across the resistor R2 attempts to exceed the base-emitter voltage VBE (approximately 0.6 V) of the transistor Tr6, the transistor Tr6 becomes conductive.

It works to reduce the collector-emitter voltage and exhibits constant voltage characteristics.

In the case of Fig. 7, the constant voltage characteristics are somewhat rounded due to the current flowing through the resistors R1 and R2, but in the case of Fig. 8, the current flowing through the resistors R1 and R2 is supplied from the power supply side, so it is constant. It is characterized by a small rounding of voltage characteristics.

Note that the constant voltage circuit is not limited to those shown in FIGS. 7 and 8, and other circuits may also be used.

Although each embodiment has been described above, it is also possible to mix the level shift circuits shown in each embodiment and connect them in series.

Furthermore, other level shift circuits, except for the circuit shown in FIG. 3, have constant voltage characteristics by themselves, and do not necessarily need to supply a constant current, and may be an unstabilized current source.

The level shift circuits shown in each of the above embodiments all serve to shift the voltage generated in the parallel resonant circuit in a direct current manner, and it is desirable that the internal four impedance in the alternating current state is O.

If an alternating current internal impedance exists, the voltage generated across the parallel resonant circuit will be divided by the input impedance seen from the base side of the transistor Tr1, and the voltage applied to the base of the transistor Trl will be an attenuated voltage. It becomes.

In particular, when implementing an IC, the more complex the level shift circuit becomes, the more parasitic capacitance increases, and the internal 1
The impedance increases, which is inconvenient.

In such a case, a phase shift will occur at the input and output ends of the level shift circuit.

In order to improve these problems, a bypass capacitor C is connected in parallel to the level shift circuit, that is, between one end of the parallel resonance circuit and the base of the transistor Tr1, as shown in FIG.
Connecting B becomes extremely effective.

[Brief explanation of drawings]

FIG. 1 is a circuit diagram showing an oscillation circuit of a proximity switch that has been proposed in the past, and FIG. 2 is a graph showing the relationship between negative conductance gosc and amplitude VT.
5, 6, 1, 8 and 9 respectively show the first, second, third, fourth, fifth, sixth and seventh embodiments of the present invention. It is a circuit diagram. L: Detection coil, C: Capacitor forming a parallel resonant circuit together with the detection coil L, Trl: Transistor connected as an eminter follower, Tr2. Tr3...
- Transistors forming a current mirror circuit constituting the feedback circuit, Tr 4 r Tr 5...Transistors forming a current mirror circuit for supplying N current to the level shift circuit.

Claims (1)

[Claims] 1. A PU consisting of a transistor connected as an emitter follower, a current mirror circuit for feeding back the output current of this transistor, and a detection coil and a capacitor to which the current fed back by the current mirror circuit is supplied. An oscillation circuit for a proximity switch comprising a resonant circuit, a current source for supplying a base current of the transistor, and a base bias level shift circuit connected between the base of the transistor and one end of the parallel resonant circuit. 2. The oscillation circuit for a proximity switch according to claim 1, wherein a capacitor is connected in parallel to the base bias level shift circuit.