JPS5885628A - Delta modulator - Google Patents

Abstract

(57) [Summary] This bulletin contains application data before electronic filing, so abstract data is not recorded.

Description

DETAILED DESCRIPTION OF THE INVENTION The invention consists of a cascade arrangement formed by a difference generator, a loop filter, a two-level quantizer, a clocked pulse-controlled sampler, and a feedback path from the output of this sampler to the input of the difference generator. , a delta modulator with a feedback loop to another power source supplying the signal to be coded comprising a difference generator sequence and a loop filter each independent of the other.

Delta modulators or delta-sigma modulators of this type are generally known. The loop filter of such a modulator is usually in the form of a single or double integrator.

It has already been proposed to form a loop filter, also commonly referred to as an over-sampled delta modulator, as a high-order filter to remove voice-band Asonization noise.

SUMMARY OF THE INVENTION An object of the present invention is to provide a delta modulator configured such that a loop filter is suitable for the signal to quantization noise ratio on the output side of the delta transformer.

The present invention consists of a difference generator, a loop filter, a lambda level generator, a clock pulse controlled sampler, and a feedback path from the output of this sampler to the power of the difference generator.
In a delta mod*a, in which the sequence of difference generators and the loop filter are each independent, the loop filter , the sampler's I
The phase transition of the feedback loop caused by the delay independent of the N wave number is within the frequency range (in this case independent of the sampler frequency delay and T is the sampling period), a minimum phase network with a phase characteristic that is refilled to Ft /100 with a margin (ffl) of It is characterized in that it is made to be constant.

The invention is based on the recognition that a delta modulator or a delta-sigma transformer P can be represented by a linear feedback network comprising a time delay device (T) and a noise source (N). The time delay in this case is represented by the time delay introduced by the sampler and the parasitic delay in the electronic components between the outputs of this sampler. The noise source is Aiko noise P generated by the sampler itself. The reason is that the output signal of the emulator is sampled at clock instants.

The optimal filter for a linear feedback network is a minimum phase filter. According to the knowledge based on the present invention, the optimal loop filter for a delta modulator or delta-sigma modulator is:
This is a minimum phase network that is compatible with the phase shift maintaining the stability of the feedback loop.

Therefore, the sum of the phase from the time delay device (T) and the phase of the loop filter must not exceed t-1100.

The present invention will be explained with reference to the drawings.

A signal to be encoded is supplied to the input terminal / of the delta modulator shown in No. 1 (2). This input terminal is terminated with a difference generator. The output terminal of the difference generator is connected to a two-level quantizer 3.
Connect to. The output terminal of the encoder 3 is connected to a clock pulse-controlled sunblader, at the output of which a coded input signal is produced in the form of an l-bit code and this signal is supplied to the output terminal j and to the return path 6. This feedback path B, which comprises a loop filter 7 (transfer function F (W) ), converts a portion of the output signal of the sunblader into a t-p wave and supplies it to the second input terminal of the difference generator 2 .

By arranging this loop filter 7 between the difference generator and the two-level encoder 3, the delta-sigma modulator shown in FIG. 2 is formed.

In practice, the loop filter 7 is a single or double integrator.

The present invention combines the delta-sigma q adjuster shown in FIG.
is at least approximately represented by the linear feedback network shown in the figure, namely a feedback network comprising a loop filter 7 (transfer function F (W) ), a delay element t (delay tT), an amplifier (gain 50) and a dose M10. This was achieved based on the recognition that it is possible.

The delay caused by the delay element t is the sunblader delay f:1
2 and 10 represent the amplification noise (N) applied to the output signal of the obfuscator 3 by the sunblader. The reason is that the output signal is sampled at clock instants.

Assuming a delay τ-T (where T is the sampling period), the fourth Ig:J song all shows a change in phase and (mfR) frequency. e = T/T-K/r-co. lol
m I! +872 or 72 of the sampling frequency aJB
At the frequency of , the phase becomes -E.

According to the invention, the loop filter 7 argF (ω)
It has a phase characteristic according to a curve B showing a change in .

The phase characteristics of both the loop filter 7 and the delay element r are represented by the curve iIO. In the frequency range from zero to ω-ω0, the overall phase has a margin (m) of approximately -E. The stability of the feedback loop is established under specific conditions by appropriately determining the cage of m. The characteristic argy(ω) is shown in FIG. 5 on a vs. Wits wave scale.

For the feedback loop in Figure 3, the phase transition of the feedback loop is t
- of the frequency passing through roo (Figure 1!O).

Then the loop gain is correctly l. The following equation holds true for gain A6.

"e"" 1IF(cu) I IN - of the wave number. (Figure q, song 11B), when the phase of the loop filter is zero, the following equation holds true.

A, -IR,F (ω0) Here, 1(6F(ω.) represents the actual W& part of F (aJ) in the case of solid-liquid @ω=ω.

song! The reason why the phase change according to lB is optimal is shown below.

L ωtoω. When argF(ω) is larger than zero, the loop filter has a differential characteristic and can reduce the difference between IF(ω)1 at the baseband frequency and IF(m)l at the frequency eIJo. can.

2. aJ<ω. If the margin m is made larger than necessary for stability, then at low frequencies.

IF (aI) l and IF (aI) l at high frequencies
〇Reducing the difference mentioned in Reasons 1 and 2 above will reduce the signal-to-noise ratio at the baseband frequency, and therefore result in suboptimal results.O Phase characteristics shown in Figure 6 is rmT/2. Fig. 3 shows the change in argBr'(ω) of the optimal loop filter in the case of m-0. This characteristic −@D is the song 1lE when τ−T1m≠0 (
This corresponds to song 11B in Figure 4].

This curve has a sampling frequency tn s = 2
E/'[' is m- at the frequency at which both the phase shift of the loop filter and the phase shift due to the delay τ are greater than tro'y. 0 can therefore be derived from the point of view that it needs to be at least twice as large as 0. Therefore, the upper limit of the frequency range in which the loop filter refills its phase shift to 1roo depends on having the lowest value of - E/τ or ω
-ff/T. The phase of the loop filter for τ〉T changes according to the curve MB (Fig. 4') and τ〈T
The phase of the loop filter for the case s changes according to the song s, for example the song 11D (see figure) for the case τ-V2, in which case a bend occurs at a frequency tn -7r/T with a phase value other than zero.

Song 1 [In order to use a minimum phase loop filter that changes according to D, as shown by the dotted line, argt+'(all is ω
-rr/T, 11r(Φ)1 in the region of 7r/T changes at a value higher than li'(c
o) Significantly different from 1. Therefore, the feedback loop attenuates the signals at frequencies near the peak −/r/T, thereby satisfying the requirement that the sampling frequency mm m fJ/T be at least equal to twice the highest dominant signal frequency. can be satisfied.

We also confirmed that the signal-to-quantization noise ratio of the modulator in the frequency band from zero to Φ is expressed by the following equation.

S/, −(3A”/8aIoT2)Cf(Oo-mB
11 + H (al month d pass here H (al) −F (a
l) eXP (-jτ)/II' (aI□) l
Oyo(ftm.

are the lowest values of frequencies ω-1r/T and ω-ff/T, T&
Well, it's the sampling period.

Figure 7 shows the bit rate fk) and the possible wavenumber band fB −
The S/N characteristics calculated as a function of the ratio to the upper limit of alB/2ff are shown.

Curve parameters a τ-T/72, m-O b r-'r/g, mo, 17rCτ-T72
．． The loop filter is an ideal integrator d τ−'1”, m
O e rmT, m-5o, 17r f τ 1.25T, [0-0, LA” beep) Speed is iaoM'o/s, video bandwidth is 5 MH
2(fb/fB暉gg), the curve f is S of naH
/N ratio, which is expressed for a sinusoidal signal without noise weighting. This takes into consideration the dirty video signal and noise weight.1. What if! In this case, the ideal amplitude characteristic of the loop filter (song @f in Fig. 7) is shown on a logarithmic frequency scale by the song sG in Fig. r.Actual approximation of the ideal characteristic Curve the Bode diagram of a
ll (indicated by

FIG. 9 shows seven examples of loop filters with phase characteristics H. This filter 7 has three RC times Wsm/1.
It is constructed by cascading arrangement of /l and /J and two separation amplifiers lda and /j. The transfer function of this filter has the following poles and zeros.

Pole Zero 1, OMHz (2X) 15MH2 (!X)6
, liMHz fOMH2B

[Brief explanation of the drawing]

1st fi! J is a block circuit diagram showing the W configuration of the delta modulator, FIG. S is a block circuit diagram showing the configuration of the delta-sigma modulator, and FIG. 3 is the delta-sigma modulator 1I shown in FIG.
Figure S is a diagram showing the phase characteristics related to the feedback network shown in Figure 3. Figure S is a diagram showing the phase characteristics of the minimum phase loop filter. A characteristic diagram showing the characteristics and amplitude characteristics on a logarithmic scale. Figure 4 shows the signal-to-noise ratio when the optimal loop phi (T) and phase margin (m) are different when the values of slow & ff1 (T) are different. Characteristics and bit speed characteristics Figure 82 is a characteristic diagram showing the ideal amplitude characteristics of the loop filter in a specific case and its approximate value, Figure 9 is a block circuit showing 7M of the loop filter. It is a diagram. l...input terminal, 2...difference generator, 3...2-level encoder, da...clock pulse control sampler,!
...m million terminals, 6...feedback path, 7...loop filter,! ...Delay element, 9...Amplifier, 10...
・Noise source, //, /J. 13...80 circuit network, #l, /j...separation amplifier. - to t/f. FIG. 8 Procedural Amendment September 16, 1980 1. Indication of the Case 1981 Patent Application No. 181286 2. Name of the invention Delta Modulator 3. Person making the amendment Relationship to the case Name of patent applicant N.・B. Phillips Fluiran Penfabriken Specification/1 Title of the invention Delta modulator 2, Claims 1 Difference generator, loop filter, two-level spoiler, clock pulse control sampler, and this sampler and a feedback loop to another power supplying the signal to be encoded comprising a cascade arrangement formed by a feedback path from the output of the difference generator to the input of the difference generator, the sequence of difference generators and the loop filter respectively In an independent delta modulator, the loop filter is modified so that the phase shift of the feedback loop caused by the frequency-independent delay of the sampler is limited by the lowest frequency of the frequencies ω-E/T and ω-E/τ. In a defined frequency range (here, the delay independent of the sample frequency and T is the sampling period), the margin (ffl) is approximately 1lrO.
1. A delta modulator, characterized in that the minimum phase circuit network has a phase characteristic such that the phase characteristic is refilled with zero, and the phase of the loop filter is constant above the cutoff frequency. 3. Detailed description of the present invention for providing the signal to be encoded, comprising a cascade arrangement formed by a quantizer, a clock pulse controlled sampler, and a feedback path from the output of this sampler to the input of a difference generator. This relates to a delta modulator G having a feedback loop to human power, and in which the sequence of difference generators and the loop filter are each independent. Delta modulators or delta-sigma modulators of this type are generally known. The loop filter of a modulator is usually in the form of a single or double integrator. It has already been proposed to form a loop filter, also commonly referred to as an oversampled delta modulator, as a high-order filter to eliminate voice band quantization noise. An object of the present invention is to provide a delta modulator configured such that a loop filter is suitable for the signal to quantization noise ratio on the output side of the delta modulator. The present invention provides a signal to be encoded comprising a cascade arrangement formed by a difference generator, a loop filter, a two-level attenuator, a clock pulse controlled sampler, and a feedback path from the output of this sampler to the input of the difference generator. In a delta modulator in which the difference generator sequence and the loop filter are each independent, the feedback loop caused by the frequency-independent delay of the sampler is A frequency range in which the phase shift is upper bounded by the lowest frequency G of frequencies ω-A''/T and ω-E/τ (here, the delay is independent of the frequency of the sampler, and T is the sampling period). ), the margin (m) is approximately /!r00
The present invention is characterized in that the minimum phase network has a phase characteristic such that the loop filter is refilled, and the phase of the loop filter is constant above the cutoff frequency. The invention is based on the recognition that a delta modulator or a delta-sigma modulator can be represented by a linear feedback network comprising a time delay device (τ) and a noise source (N). . The time delay in this case is represented by the time delay caused by the sampler and the parasitic delay in the electronic elements between the input and output of this sampler. Also, the noise source refers to quantization noise generated by the sampler itself. The reason is that the output signal of the numerator is sampled at clock instants. The optimal filter for a linear feedback network is a minimum phase filter. According to the present invention, the optimal loop filter for a delta modulator or delta-sigma modulator is a minimum phase network whose phase shift is also compatible with preserving the stability of the feedback loop. Therefore, the sum of the phase from the time delay device (τ) and the phase of the loop filter must not exceed /iroo. The present invention will be explained with reference to the drawings. The input terminal / of the delta modulator shown in FIG. 1 is supplied with the signal to be encoded. This input terminal is terminated with a difference generator 2. The output terminal of the difference generator 2 is connected to a two-level quantizer 3. The output terminal of the quantizer 3 is connected to a clock pulse-controlled sample plug, which produces at the output of this sampler a coded human input signal in the form of an l-bit code, and this signal is supplied to the output terminal S and to the return path B. . Loop filter 7 (
This return path B, which has a transfer function F(ω)), supplies a part of the output signal of the sunblader as a p-wave to the second input terminal of the difference generator 2. By disposing this loop filter 7 between the difference generator 2 and the two-level modulator 3, a delta-sigma modulator shown as 1 in FIG. 2 is formed. In practice, the loop filter 7 is a single or double integrator. The present invention combines the delta-sigma modulator shown in FIG.
The linear feedback network shown in the figure includes a loop filter 7 (transfer function F(ω)), a delay element l (delay amount τ), an amplifier 9 (
This is based on the recognition that the noise source 10 can be represented at least approximately by a feedback network comprising a gain Ae) and a noise source 10. The delay caused by the delay element t represents the delay of the sampler l, and the noise source lθ represents the quantization noise (N) applied to the output signal of the quantizer 3 by the sampler t. The reason is that the output signal is sampled at the clock time. Given the delay .tau.-T (where T is the sampling period), curve A in FIG. 3 shows the change in phase and (circulation) frequency. ω-ff/T = K/τ-ω. At a frequency of -ωS/2, that is, /2 of the sampling frequency S, the phase becomes -E. According to the present invention, the loop filter 7 has a phase characteristic that follows a curve B representing a change in argF(ω). The phase characteristics of both the loop filter 7 and the delay element r are represented by a curve C. In the frequency range from 0 to 100, the total phase is approximately -E in margin (m). The amplitude characteristic IF(ω)I and phase of the optimal minimum phase loop filter when =T is determined by appropriately determining the value of m so as to establish the stability of the feedback loop under specific conditions. Characteristic a
rgF(ω) is shown on a logarithmic frequency scale in FIG. For the feedback loop in Figure 3, the phase transition of the feedback loop is /
100 (the same curve C) At the frequency -aJo• passing through f, the loop gain is correctly /. Gain A. The following formula holds true for . Ae = l/l F (m. > 1 - of the frequency (the same curve B), and when the phase of the loop filter is zero, the following formula holds true: Ae-1/ReF (ω.) Here, R, F (ω.) represents the real part of F(ω) in the case of frequency ω-ω. The reason why the phase change according to curve B is optimal is shown below. 1. In cc5) tuo, If argF(a+) is greater than zero (I/), the loop filter has differential characteristics and this means that IF(ω) at the baseband frequency
1 and IF(ω)1 at frequency ω can be reduced. 2 ω〈ω. If the margin m is made larger than necessary for the stability G, the difference between 1F(ω)1 at low frequencies and IF(ω)1 at high frequencies decreases. Reducing the differences mentioned in Reasons 1 and 2 above will reduce the signal-to-noise ratio at the baseband frequency and therefore lead to suboptimal results. The phase characteristics in FIG. 6 are τ-T/2. The change in argF(ω) of the optimal loop filter when m=o is shown. This characteristic curve is compared with the curve iE (corresponding to curves B & B in Fig. 1) when τ = '1'' and m≠0. It can be derived from the viewpoint that it is necessary to make the phase shift of the frequency ω- at which both the phase shift of and the phase shift of l due to the delay τ larger than troo. The upper limit of the frequency range that refills the transition to 1.rOO depends on having the lowest value, ω − E / τ or ω = 1
Located at r/T. The phase of the loop filter when τ〉T varies according to curve B (Fig. 1), and the phase of the loop filter when τ〈T changes according to a curve, e.g. curve D when τ - T/2 (Fig. 6). In this case, when the phase value is other than zero, bending occurs as the frequency increases. Since a minimum phase loop filter that changes according to the curve is used, as shown by the dotted line, -ff of argF (ω)
The IF(ω)1 in the region of =1r/T varying above /T is significantly different from the IF(ω)1 at low frequencies. Therefore, due to the feedback loop, =1r
Significantly attenuates signals at frequencies around lT, =11. The requirement that the sampling frequency S = 2VT be at least equal to twice the overall highest dominant signal frequency can be fully satisfied. We also confirmed that the signal to quantization noise ratio of the modulator in the frequency band from zero to 8 is expressed by the following formula: S/N-(3/r2/8ω.T2) (f ((3o
・(11Bl 1+old a+)l-2da+:j1kotm
&Z H(ffl) = F (al) eXI)
(-jT)/IF(W.) I and ω. is the lowest value of the frequency ω-7r/τ and =T/T, and T is the sampling period. Figure 7 shows the bit rate fb and the possible frequency band −f
The S/N characteristic calculated as a function of the ratio of B=ωB721'r to the upper limit value is shown. Curve parameters a τ= T/2, m = Ob y=
T/2, m=o, 11rCτ-T72. The loop filter is an ideal integrator d τ='r, m=. e r=T, m=o, 1rr f T'= 1.25T, ffl=0.
When the 17ir pit speed is 1 tlOMb/s and the video bandwidth is 5 MH2 (fb/fB = 28), the curve f shows an S/N ratio of 3 aB, which is a sinusoidal signal without noise weighting. expressed relative to the signal. This is 5trd when considering video signal and noise weighting.
It is equivalent to p. The ideal amplitude characteristic of the loop filter in this case (curve f in Fig. 7) is shown on a logarithmic frequency scale by curve G in Fig. r. Plot the Bode plot of the actual approximation of the ideal characteristics.
Indicated by lJH. FIG. 9 shows an example of the loop filter 7 that provides the phase characteristic H. This filter 7 consists of three RC networks //,
/2 and /13 and two separation amplifiers /2 and /3 are arranged in cascade. The transfer function of this filter has the following poles and zeros. Pole Zero 1.0MHz '(2X) 15MH2(2X
) 6.25MHz 70MH2t Brief description of the drawings Figure 1 is a block circuit diagram showing the configuration of a delta modulator, Figure 2 is a block circuit diagram showing the configuration of a delta-sigma modulator, and Figure 3 is a block circuit diagram showing the configuration of a delta-sigma modulator. The delta-sigma modulation model number shown is a block diagram of both circuits showing an example of the case where the present invention is applied.
FIG. 1 is a characteristic diagram showing the phase characteristics related to the feedback network shown in FIG. 3, FIG. S is a renal diagram showing the phase characteristics and amplitude characteristics of the minimum phase loop filter on a logarithmic scale, and FIG. A characteristic diagram showing the phase characteristics of the optimal loop filter when the delay amount (τ) values are different. Figure 7 shows the signal-to-noise ratio characteristics when the delay amount (τ) and phase margin (m) are different. Figure 1 is a characteristic diagram showing the ideal amplitude characteristics of a loop filter in a specific case and its approximate value. Figure 9 is an example of a loop filter. FIG. /...Input terminal, 2...Difference generator, 3...2 level generator, l...Clock pulse control sampler, 5
...Output terminal, 6...Feedback path, 7...Loop filter, g...Delay element, 9...Amplifier, /θ...
・Noise source, /l, /2. /3...RC circuit network, /F, /S...separation amplifier. NPA Phillips Patent Applicant, Yaira, 6A, Air 1. Ke.

Claims (1)

[Claims] L A signal to be encoded comprising a cascade notation formed by a difference generator, a loop filter, a two-level positioner, a clock pulse controlled sampler, and a feedback path from the output of this sampler to the input of the difference generator. In a delta modulator in which the sequence of difference generators and the loop filter are each independent, with a feedback loop to the other input that supplies On the frequency axis 8 (where τ denotes the frequency-independent delay of the sampler and T denotes the sampling period), the phase shift is upper bound by the lowest frequency of the frequencies W −KIT and W g ff/f, In the margin (ffl), rt /
) A delta modulator O characterized in that it is a minimum phase network having a phase characteristic such that it is refilled with 00, and the phase of the loop filter is constant above the cutoff frequency.