JPS5857068B2 - Switching power supply control method - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a control system for a switching power supply using an inverter or a converter.

Conventionally, various types of switching power supplies have been used, such as a one-stone flyback type, a one-stone forward converter type, a chopper type, a push-pull type, and a bridge type. This is done by changing the on period, the off period, or both of the switching elements.

By the way, switching power supplies often require electrical isolation between the input power supply side and the output side, and for this reason, conventional switching power supply control methods tend to have complicated circuit configurations. .

In view of the above points, the present invention provides a control method for a switching power supply that allows output control and, if necessary, stabilization control, to be performed with a simple circuit configuration while the input terminal and output side of the switching power supply are insulated. This is what we are trying to provide.

DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of a control method for a switching power supply according to the present invention will be described below with reference to the drawings.

FIG. 1 shows a first embodiment of the present invention, which is applied to a single-stone flyback type switching power supply.

In this figure, a transistor 2 as a switching element is connected in series to one end of the primary winding IIA of the output transformer 1, and the other end of the primary winding 1A is connected to the positive power supply terminal P of the DC power supply. The emitters are connected to the negative power supply terminals Nl.

A rectifier circuit including a diode 3 and a capacitor 4 is connected to the secondary winding 1B of the transformer 1, and an output terminal A is connected thereto.

A load 5 is connected between B and B.

On the other hand, the control circuit 10 includes an oscillator 11 that generates a rectangular wave of a constant frequency, a comparator 12, and a logic circuit 13.
It has a pulse width modulator 14 consisting of a pulse width modulator 14, and a drive circuit 15 for driving the transistor 2, and
The current transformer 20 has a number of windings.

The first winding 20A of the current transformer 20 is inserted into the collector side of the transistor 2, and the output voltage ecT2 of the second winding 20B is applied to one input terminal of the comparator 12.

The other input terminal of the comparator 12 has a reference voltage VRB.
F is added.

A variable resistor 2 is connected to the third winding 20C of the current transformer 20.
1 is loaded.

In the above configuration, the drive circuit 15 of the control circuit 10
The magnetic energy stored in the transformer 1 during the ON period of the transistor 2 is released from the secondary winding 1B through the diode 3 to the capacitor 4 and the load 5 during the OFF period of the transistor 2.

In this case, the collector voltage waveform of transistor 2 is as shown in Figure 2A, the collector current waveform is as shown in Figure 2B, and the collector current IC is Inductance, equivalent primary current of residual energy of transformer 1, t: time after transistor 2 is turned on).

For simplification, consider the case where all the magnetic energy of the transformer 1 is supplied to the capacitor 4 and the load during the off-period of the transistor 2. Since ■o-〇, the magnetic energy of the transformer 1 is supplied to the capacitor 4 and the load during the off-period of the transistor 2. The stored energy J is (where Icp is the maximum value of the collector current), and is equal to the energy supplied to the secondary side if efficiency is ignored.

In a steady state, all of the secondary side energy is supplied to the load, so the output power Pout is given by the frequency f.

From this equation (3), the collector current IC of transistor 2 is
By detecting the magnitude of the transistor 2 and controlling the on-period of the transistor 2, it is possible to supply constant power to the load without being affected by input voltage fluctuations, etc., and the collector current IC
It can be seen that the output power can be variably controlled by varying the detection setting value.

Control circuit 10? Here, the collector voltage iIc is detected by the current transformer 20, and each winding of the current transformer 20 generates a power having a square waveform as shown in FIG. 2C.

Further, the detection set value is adjusted by a variable resistor 21.

That is, the input impedance of the comparator 12 is R
c, and if the excitation current is ignored, the first
The winding voltage eOTs is (however, n2=N2/Nt t n3=N3/N1.R
v: resistance value of the variable resistor 21).

Further, the second winding voltage ec'r2 is as follows.

Here, since it becomes the input input of the comparator 12, it is possible to vary the voltage eOT□ by changing the variable resistor 21.

The second winding voltage ecT2 generated as described above is applied to one input terminal of the comparator 12, and the reference voltage ecT2 is applied to one input terminal of the comparator 12.
Compared to EF.

The output of the comparator 12 is the voltage ecT2, which is the reference voltage VR.
When it exceeds EF, it changes and controls the drive circuit 15 via the logic circuit 13 to turn off the transistor 2.

Therefore, by changing the resistance value RV of the variable resistor 21 (thereby, the second winding voltage ecT2 can be varied, and as a result,
Since the detection set point of the collector current IC of the transistor 2 can be varied and the on-period of the transistor 2 can be varied, the output power can be variably adjusted.

According to the first embodiment, the output power can be easily adjusted using the variable resistor 21 which is insulated from both the primary side circuit and the load side circuit from the current transformer 20c. Since the variable resistor 21 is insulated from the frequency and only a high frequency small signal voltage is applied, manual adjustment of the variable resistor 21 is extremely safe.

Further, it can be used as a power supply for a microwave oven whose output can be adjusted by making small changes such as adding a series inductor between the output terminals A and B and the magnetron with the magnetron as a load, and is suitable for a constant power supply power supply.

Furthermore, the variable resistor 21 is a thermistor, a photoconductor, an FE
A similar operation is possible even if the drain/source resistance of T is replaced, and a configuration in which a plurality of third windings 20C are provided and a variable resistor is connected to each is also possible. In addition to the fact that variable resistors, thermistors, etc. are insulated from the primary side and load side, it can be widely applied as a power source for devices that require various automatic adjustment functions.

Further, the load of the third winding 20C does not need to be a resistor whose resistance value changes directly, such as the variable resistor 21, but can be obtained as a variable impedance load that can change the equivalent impedance viewed from the winding side. That's fine.

Note that the first winding 20A is inserted into the emitter side of the transistor 2, and the voltage generated in the first winding 20A is transferred to the second winding 20A.
By setting the value corresponding to the winding voltage ecT2, the first winding 20A and the second winding 20B can be used in common.

FIG. 3 shows a second embodiment in which the present invention is applied to a flyback type switching power supply using a blocking oscillator, in which the above-mentioned equivalent impedance is changed.

In this figure, the collector winding 30 of the output transformer 30
A blocking oscillator is constructed by an A1 base winding 30B and a transistor 31, and this blocking oscillator receives a DC voltage obtained by rectifying the AC input supplied to the AC input terminals C and Dl with a rectifier 32 and smoothing it with a capacitor 33. Added.

A rectifier circuit including a diode 3 and a capacitor 4 is connected to the secondary winding 30C of the transformer 30, so that a DC output is output between output terminals A and B.

On the other hand, the control circuit 10A includes a current transformer 20 having three windings, a transistor 40 provided in the base circuit of the transistor 31 of the blocking oscillator, and a transformer 3.
0 secondary side circuit (and a transistor 41 provided therein).

The first winding 20A of the current transformer 20 is inserted on the collector side of the transistor 31, and the output voltage e of the second winding 20B is maintained.

T2 is applied between resistor 42 and the base and emitter of transistor 40.

Further, a resistor 44 is connected to the third winding 20Cl via a diode 43 as a load.

The resistor 44 is connected to the output terminal A via the transistor 41.
, B, and the base of the transistor 41 is connected between the resistor 45 and the constant voltage diode 4 connected between the output terminals A and B.
It is connected to the connection point with 6.

On the other hand, a resistor 47 and a capacitor 48 are connected between the base of the transistor 31 and the base winding 30B of the transformer 30.
The collector of the transistor 40 is connected to the base of the transistor 31, and the emitter of the transistor 40 is connected to the connection point between the base winding 30B and the parallel diode 49 and capacitor 50.

Further, a diode 51 is connected between the base and emitter of the transistor 31, and a DC bias is applied to the base via a resistor 52.

In addition, a spike killer 53 is connected in parallel to the collector winding 30A.
is connected.

In the above configuration (here, the collector current waveform of the transistor 31 of the blocking oscillator is as shown in FIG. 2nd winding voltage ecT2; 3rd winding voltage ec
The same relationship as in the above case occurs with T3.

Here, the equivalent load impedance Z3 of the third winding 20C
C. If the forward voltage of the diode 43 is ignored, the collector current of the transistor 31 is 10. This is shown by the relationship between the reference current c3 of the transistor 41 and the direction of .

That is, when the resistance value of the resistor 44 is R4, IO
If 2・R4≧e OTs, the diode 43 is biased in the reverse direction, so Z3-oo, and Ic3・R4<
At the end of ecT3, diode 43 is conductive (however, Z
D: forward resistance of the diode 43; cT3: current of the third winding 20C).

It can be seen that the load impedance Z2 of the second winding 20B is large and is a function of the collector current IC17IO2 of the transistors 31 and 41, ignoring ZD and ZD for the sake of simplicity.

In addition, the second winding 20B (second winding voltage ec
T2 is as follows, and this is also related to the collector currents Ic1 and Ic3 of the transistors 31 and 41.

Therefore, the timing at which the transistor 40 driven by the second winding 20B turns on changes in accordance with the collector current ICs of the transistor 41.

When the transistor 40 changes its on state, the transistor 31 is quickly turned off due to the negative voltage of the capacitor 50, so the timing at which the transistor 40 is turned on changes (therefore, the on period of the transistor 31 also changes).

Here, since the error amplifier is configured by the transistor 41 and the voltage regulator diode 46, the output voltage increases and the voltage (
When this happens, the transistor 41 is quickly forward biased, and the collector current I03 of this transistor 41 increases rapidly.

Therefore, the second winding voltage eOT2 becomes a dog, the time when the transistor 40 turns on is brought forward, the on period of the transistor 31 is shortened, and the output voltage is reduced.

The output voltage is kept constant by such a negative feedback effect that controls the output voltage.

In the above case, if the off period of the transistor 31 is determined by the time during which all the magnetic energy of the transformer 30 is released from the secondary side as a typical example in the blocking oscillator, then the operating primary voltage EC of the blocking oscillator is Blower output voltage E.

is the winding ratio n and the duty ratio KON=TON/(T
ON+TOFF )) Determined by this.

That is, Eo=n・(T□N/Topp)EC,・−
...-(io) (However, TON: ON period, TOFF
: off period), and the output energy J for one cycle is (where c, P: maximum collector current value of the transistor 31).

The output power P is , and it can be seen that the output can be controlled by variable control of the on-period, such as load fluctuations and input voltage fluctuations, although the oscillation frequency fluctuates.

According to the embodiment described above, output stabilization control can be easily performed with the current transformer 20 insulating the primary side circuit and the load side circuit.

Moreover, the second winding voltage eG of the current transformer 20! Equation (9) representing T2 includes a voltage term induced by the collector current a I 01 of the transistor 31 independently of the collector current I 03 of the transistor 41, so it is difficult to avoid the occurrence of an overload, an output short circuit, or the rise of a large capacitance load, etc. Even if the collector current Ic3 of the transistor 41 is zero, the circuit constants can be adjusted appropriately.
This setting (this prevents an overcurrent that prevents the collector current IOt of the transistor 31 from increasing and, in combination with the magnetic saturation of the transformer 30, destroying the transistor 31 or causing an abnormally large current to flow to the load and the switching power supply circuit. It is possible to perform a protective effect.

Note that by controlling the base DC bias of the transistor 31 of the blocking oscillator, the transistor 31
It is also possible to control the on-period or off-period of the.

Further, the second winding voltage ecT2 can be used not as a timing signal for directly turning off the base drive, but as a feedback signal using the amplitude value of the voltage.

Influences such as the spike removal circuit, the distributed capacitance of the transformer 30, and the recovery time of the rectifier diode. In the case of a push-pull converter, etc., where the change in collector current during the on period is small, l is
It is possible to modify the circuit by correcting or shaping the waveform of the second winding voltage ecT2 and then using it as a timing detection signal to control the on/off period of the switching element or as a control signal using the amplitude value. it is obvious.

In this case, since the control signal is synchronized with the operating frequency, it is often easy to avoid the effects of induced noise.

FIG. 4 shows a third embodiment of the present invention, which is applied to a push-pull type switching power supply.

In this figure, switching transistors 61 and 62 are push-pull connected to the primary winding 160A of the output transformer 60, and hole quadrature transformers 64 are connected between the bases and emitters of these transistors 61 and 62 via resistors 63, respectively. , 65 are inserted.

Primary winding 64A16 of their input transformer 64.65
Drive transistors 66 and 67 are provided on the 5A side.

Between the primary winding of the output transformer 60, the center point tap of %l60A and the emitters of the transistors 61 and 62, and between the connection point of the primary windings 64A and 65A of the input transformer 64 and 65 and the emitter of the transistors 66 and 67. A DC voltage obtained by rectifying the AC input supplied to AC input terminals C and D by a rectifier 32 and smoothing it by a capacitor 33 is supplied.

The secondary winding 60B of the output transformer 60 (this is the diode 6
8. .

69, a double-wave rectifier circuit is connected from the reactor 70 and the capacitor 71 so that a DC output is output between the output terminals A and B.

On the other hand, the control circuit 10B generates a constant frequency rectangular wave 2; A oscillator 8nμ; a pulse width modulator 81 including a logic circuit and a waveform shaping circuit; an operational amplifier 82; and a current transformer having three windings. It has a container 20.

The oscillator 80 and pulse width modulator 81 (this is the rectifier 3
The voltage across a constant voltage diode 84 connected to the output side of 2 via a resistor 83 is supplied as a power source.

A primary current 11. is supplied, and the output voltage e2 of the second winding 20B is supplied to the pulse width modulator 8.
1 logic circuit.

A resistor 87 is connected to the third winding 20C via a diode 86.
and operational amplifier 82 are connected.

A value obtained by dividing the output voltage by a resistor 88, a resistor 89, and a variable resistor 90 is applied to one input terminal of the operational amplifier 82, and a value obtained by dividing the output voltage by a resistor 88, a resistor 89, and a variable resistor 90 is applied to the other input terminal. A reference voltage determined from the voltage and the forward voltage l of the diode 92 is supplied, and an error amplifier is configured by these.

In the above configuration, the oscillator 80 generates a rectangular wave voltage e with a constant frequency as shown in FIG. The line 20A (one blow type (til) as shown in FIG. 5B synchronized with the voltage e1 is applied).

The second winding voltage e2 is equal to the primary current 11 and the third winding 20C.
The drive voltage e35e4 applied to the bases of the transistors 66 and 67 has a rising timing determined by the rise or fall of the voltage e1, respectively, and has a waveform as shown in FIG. 5C. When the voltage e2 matches the set value, the waveform falls as shown in the fifth diagram E.

Here, the voltage e2 (1) is the same as the voltage eOT2 in the second embodiment described above.

As the output voltage appearing between B increases, the on-period of the transistors 66, 67 and eventually the transistors 61, 62 becomes shorter, so that the output voltage can be controlled stably.

According to the third embodiment, the control feedback signal (that is, the voltage e
There is an advantage that the waveform of the one-blow type flow i1 of the current transformer 20 can be set so as to obtain 2).

In each of the above embodiments, a transistor is used as an example of a switching element, but a GC8, FET, etc. may be used as a switching element.

As described above, according to the present invention, there is provided a control method for a switching power supply that allows output control and, if necessary, stabilization control, to be performed with a simple circuit configuration while the input side and output side of the switching power supply are insulated. get.

[Brief explanation of the drawing]

FIG. 1 is a circuit diagram showing a first embodiment of a control method for a switching power supply according to the present invention, FIG. 2 is a waveform diagram for explaining its operation, and FIG. 3 is a circuit diagram showing a second embodiment. FIG. 4 is a circuit diagram showing the third embodiment, and FIG. 5 is a waveform diagram for explaining its operation. L30,60... Output Lance, 2,31
. 40.41, 61, 62, 66.67... Transistor, 3, 43, 51, 68, 69, 86°92.
...Diode, 4, 33, 48, 50°71.
...Capacitor, 10. IOA, IOB...
... Control circuit, 11, 80 ... Oscillator, 14,
81...Pulse width modulator, 15...Drive circuit, 20...Current transformer, 20A...
...First winding, 20B...Second winding, 20
C...Third winding, 21,90...Variable resistor, 42,44,45,63,85°87, 8B
, 89...Resistor, 46,84,91...
... Constant voltage diode.

Claims (1)

[Claims] An output transformer 1, 30 having a primary winding IA, 30A and a secondary winding 1B30C, and a switching element 2, 31 that switches the input DC voltage supplied to the primary winding side. In the switching power supply, the current of the switching element is transferred to the first winding 2 of the current transformer 20.
0A, and the switching element is turned off when the detected voltage across the second winding 20B of the current transformer 20 reaches a set value, and the detected voltage is applied to the current transformer 20. A control method for a switching power supply, characterized in that the on-period of the switching element is controlled by changing the load impedance on the third winding 20C side. 2. The load impedance on the third winding 20C side of the current transformer 20 is adjusted to the output voltage on the output transformer secondary winding side or A control method for a switching power supply according to claim 1, characterized by an output current. 31st windings IA, 30A, 60A and secondary windings IB, 3
Output Lance 1°30.60 with 0C, 60B,
In a switching power supply equipped with a switching element 2゜31.61, 62 that switches the input DC voltage supplied to the primary winding side, the current flowing through the switching element at the same frequency as the operating frequency of the switching power supply. A signal current having a synchronized period and an amplitude that increases or decreases with time within the period is passed through the first winding 20A of the current transformer 20, and the current transformer 20
The switching element is turned off when the detected voltage across the second winding 120B reaches a set value, and the detected voltage is applied to the load impedance on the third winding 20C side of the current transformer 20. 1. A control method for a switching power supply, characterized in that the on-period of the switching element is controlled by making it variable. 4. The load impedance on the third winding 20C side of the current transformer 20 is increased when the output voltage or output current on the output transformer secondary winding side reaches a predetermined value. A control system for a switching power supply according to claim 3, characterized in that: