JPS5821512B2 - Kouriyudendo Kinoseigiyosouchi - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a motor control device that operates an electric motor using a Cyrisk power conversion device, and particularly to a motor control device that reduces the installed capacity of the Cyrisk power conversion device and compensates for the armature reaction of an electric motor. The present invention relates to a control device for an AC motor that can obtain sufficient output.

FIG. 1 is a circuit diagram of an example of a conventional motor control device for operating a synchronous motor.

1 is a cycloconvert that receives AC voltage from a commercial AC power supply at its input and outputs variable frequency three-phase AC at its output, and is connected in antiparallel with a CYRISC pure bridge circuit UP,
It consists of three sets: UN, VP, V, and WP, WN.

Reference numeral 2 denotes a synchronous motor driven by the cycloconverter 1, which has three-phase armature windings U, V, and W and a field winding F.

3 is a field control circuit for supplying field current to the field winding F and controlling its magnitude; 4 is a speed generator for detecting the rotational speed of the synchronous motor 2; 5 is a speed command circuit; 6 is a speed deviation amplifier that matches and amplifies the speed command signal sent from the speed command circuit 5 and the speed feedback signal which is the output signal of the speed generator 4, and its output becomes a torque command signal.

7 has a phase according to the rotational angular position of the rotating shaft of the synchronous motor 2, and 3 has a constant amplitude value and has a phase difference of 120 degrees from each other.
A position detector that outputs two sine wave signals (three-phase signals).
In this example, a structure is used in which the rotor has a permanent magnet and the stator has three Hall generators attached facing the rotor.

8 is one of the output signals of the position detector 7 and the speed deviation amplifier 6
Multiplier 9 outputs a sinusoidal current command signal for controlling the output current (U phase) of cycloconverter 1 by multiplying the output signals of 10 is a current deviation amplification device that matches and amplifies the current command signal and the output signal of the current detector 9; 11 is a current deviation amplification device; 12 is an automatic pulse phase shifter for controlling the firing phase of the thyrisk circuit UP and UN according to the output signal of the thyrisk circuit 10; 12 is an automatic pulse phase shifter for controlling the firing phase of the thyrisk circuit UP and UN; This is a gate amplifier that alternately supplies gate signals to the UN.

Note that although only the control circuits for the thyrisk circuits UP and UN are shown in the figure, other thyrisk circuits VP, vN, and W are shown.
The same control circuits as part numbers 8 to 12 are also available for P and WN.

The description of them will be omitted.

Next, the operation of this circuit will be explained.

The phase detector 7 includes a rotor equipped with a permanent magnet connected to the rotating shaft of the synchronous motor 2, and a Hall generator that detects the magnetic field created by the permanent magnet and generates a signal according to the magnitude of the magnetic field. The permanent magnet 1 having a stator provided with the same number of poles as the number of poles of the synchronous motor 2 alternately has N poles and S poles.
They are arranged on the rotor to form poles, and the Hall generators are mounted on the stator 120 electrical degrees apart from each other.

As a result, the phase detector 7 outputs a sinusoidal signal S U "" S with a constant amplitude as shown in FIG. 2a.
W is detected.

This signal is a signal related to the relative position (angle) of the field and armature of the synchronous motor 2, and is a signal related to the relative position (angle) of the armature and the field of the synchronous motor 2, and
has the illustrated phase relationship with the armature voltage eU"'-eW.

In the multiplier 8, the output signals of the position detector 7 and the speed deviation amplification device 6 are multiplied together. Drawer 6
A current command signal proportional to the output signal of is extracted.

Next, the ignition of the thyrisk circuits UP and UN is controlled according to the operations of part numbers 9 to 12, and the output current thereof is controlled to a value according to the current command signal as shown in FIG. 2.

That is, the magnitude of the output current is controlled to a value proportional to the output signal of the speed deviation amplifier 6, which is a signal corresponding to the speed deviation, and the phase of the current is controlled to be in the same phase as the signal of the position detector 7. .

This relationship does not change at all even if the rotational speed of the synchronous motor 2 changes, and when the rotational speed is zero, the magnitude of each armature current iU to iw is determined by the magnitude of the speed deviation and the field at that time. The value depends on the relative positional relationship (angle) between the magnet and armature.

As a result of such control, when the rotational speed is near zero, the following problems occur.

That is, when the rotation speed is not zero, UP-W
Since each N sirisk circuit conducts current for a period of π, the average value ia of its output current is, where Im is the amplitude value of the armature current. However, when the rotation speed is zero, the output current A case arises in which the flow is equal to Im and continues, and in that case, i a = I m ・・・・・・・・・・・・・・・
・・・・・・・・・・・・・・・・・・・・・ (2
).

That is, compared to the normal case described above, a current π times as large as that flows through the Cyrisk circuit, which may lead to overheating of the Cyrisk.

This phenomenon is hereinafter referred to as a current concentration phenomenon, but in which of the SI risk circuits UP to UN the current concentration occurs is determined by the relative positional relationship between the armature and the field in that case. , it is necessary to take into account current concentration for all thyrisk circuits, and the capacity of all thyrisk circuits is required to be π times the capacity determined in the normal case.

Therefore, the cycloconverter 1 has become a considerably uneconomical device, which is a major disadvantage.

Another disadvantage is that as the load increases (increase in armature current), the power factor decreases and the armature terminal voltage increases. (Equipment) There is a disadvantage of increased capacity.

The reason for this is that, as shown in FIG. 2, when the output signal phase of the position detector 7 is set so that the armature current iU and the no-load induced voltage eou are in phase,
As shown in the vector diagram in Fig. 3, under load, armature reaction XsIM causes armature terminal voltage V to advance by a amount compared to armature current ■M, and the magnitude of V increases due to the no-load induced This is because the voltage becomes higher than the voltage Eo, resulting in a decrease in the power factor and an increase in the voltage.

The purpose of the present invention is to eliminate the disadvantages of conventional devices, prevent current concentration phenomenon and power factor drop during load, and reduce the installed capacity of thyristor power converters (cycloconverters) and electric motors. An object of the present invention is to provide a control device for an electric motor.

The present invention is characterized in that the motor has a structure equipped with multiphase field windings, and the frequency of the field current flowing through the field windings is controlled according to the output signal of the oscillator. , a signal that controls the magnitude and phase of the field current according to the magnitude of the armature current of the motor, and has a phase that corresponds to the relative positional relationship between the armature of the motor and the field, and the oscillator. The phase of the armature current is controlled in accordance with the signal obtained by performing a predetermined calculation based on the output signal of.

FIG. 4 shows a circuit configuration diagram of a motor control device showing an embodiment of the present invention.

Part numbers 1,4-6°8-12 are the same as those with the same numbers shown in FIG. 1, and therefore their explanation will be omitted.

13 are three-phase armature windings U, V, and W, and two field windings F1.13 that generate mutually orthogonal magnetomotive forces. The electric motor 14 equipped with F2 is a position detector that outputs two sine wave signals (two-phase signals) having a phase corresponding to the rotational angular position of the rotating shaft of the electric motor 13 and having a phase difference of 90 degrees from each other. The construction is similar to that described above (part number 7).

15 are two sine wave signals S1.15 having a phase difference of 90 degrees from each other. An oscillator outputting S2, 16 is a field winding F1. F
A constant excitation command circuit outputs a signal (hereinafter referred to as a constant excitation command signal) for commanding the magnitude of the constant excitation component of the field current IFtylF2 flowing through the constant excitation command signal 17, and the output of the oscillator 15. Multiplier for multiplying and combining signal S1, 18
19 is a multiplier that multiplies the output signal of the speed deviation amplifier 6 and the output signal S2 of the oscillator 15, and 19 is an alternating current of a Cyrisk pure bridge circuit 23.24 (hereinafter simply referred to as Cyrisk circuit 23.24) for field control, which will be described later. A current detector for detecting the input current; 20 a current deviation amplification device that matches and amplifies the respective output signals of the multipliers 17 and 18 and the current detector 19; 21 a current deviation amplification device according to the output signal of the current deviation amplification device 20; , an automatic pulse phase shifter for controlling the firing phase of the thyrisk circuits 23 and 24; 22 is a gate amplifier that supplies a gate signal to the thyrisk circuit 23 or 24 depending on the positive or negative direction of the field current iF1; 23°24 is the field winding F
This is a Cyrisk circuit that supplies field current jFx to 1.

25 is the constant excitation command signal and the output signal S of the oscillator 15;
A multiplier that multiplies 2 by 2, a phase inverter 26 that outputs a signal whose phase is inverted with respect to the output signal S1 of the oscillator 15, and a phase inverter 27 that outputs the output signal of the speed deviation amplifier 6 and the phase inverter 26.
28 is a current detector for detecting the AC input current of a field control thyristor pure bridge circuit 32.33 (hereinafter simply referred to as thyrisk circuit 32.33), which will be described later. A current deviation amplifier 30 matches and amplifies each output signal of the multiplier 25° 27 and the current detector 28, and a current deviation amplifier 30 controls the firing phase of the silisk circuit 32° 33 in accordance with the output signal of the current deviation amplifier 29. 31 is a gate amplifier that supplies a gate signal to the thyrisk circuit 32 or 33 depending on the direction of the field current 1p2, and 32.33 is a field winding F2.
This is a Sirisk circuit that supplies field current jF2 to.

A current phase 34 multiplies the output signals of the position detector 14 and the oscillator 15 and outputs a three-phase sine wave signal (hereinafter referred to as a current phase command signal) having a frequency equal to the sum of the frequencies of both signals. This is a command circuit.

FIG. 5 shows a detailed circuit diagram of the current phase command circuit 34.

35 to 38 are multipliers for multiplying and combining the output signals of the position detector 14 and the oscillator 15; 39 is a subtractor for taking out the difference between the output signals of the multipliers 35 and 36; and 40 is a multiplier for the output signals of the multipliers 37 and 38. Adder that takes out the sum, 41 to 43 are subtracters 3
9 and the adder 40 at a predetermined ratio as described later, and outputs a three-phase sine wave signal.

First, the operating principle of the present invention, in particular the compensation of armature reaction, will be explained using the vector diagram shown in FIG.

In the figure, IFA is the field magnetomotive force generated by the two field windings F1 and F2, and 1M is the magnetomotive force generated by the armature winding.

The former induces a no-load induced voltage Eo, and the latter causes a synchronous reactance drop The impact can be compensated.

In other words, this means that the magnitude and phase of the field magnetomotive force are determined according to the relationship of the vector IFA obtained by combining the vector IFO (corresponding to the terminal voltage V), which is orthogonal to IM and has a constant magnitude, and the vector ■M. This is achieved by controlling the

The following will specifically explain the operation of the circuit shown in FIGS. 4 and 5.

The oscillator 15 includes a V-F converter that oscillates at a predetermined frequency;
A ring counter that receives the output signal pulses and sequentially outputs output signals from a large number of output terminals, a D-A converter that generates two sinusoidal signals having a 90 degree phase difference according to the output signals of the ring counter, and It consists of a filter circuit used to make the output signal of the DA converter more similar to a sine wave.

The output signal S1. of the oscillator 15 thus obtained. S
2 can be expressed as in the following equation.

Note that the amplitude value of the signal has no important meaning in this case, so its description will be omitted.

Below, other signals will be similarly omitted unless necessary.

51=CO3(ωSt+θ)・・・・・・・・・・・・
・・・・・・・・・(3) S225IN(ωSt+θ)
(4) Here, ωS: Signal angular frequency θ: The signal phase multipliers 17 and 25 at time t=0 are In order to match the signal and the constant excitation command signal, Sl.

A signal P1.S2, which is in phase with S2 and has a magnitude proportional to the constant excitation command signal, is expressed by the following equation. Output P2.

p1= Ep cos (ωst+θ)・・・・・・
・・・・・・・・・・・・ (5) P2=EPSIN(ω
st 10)・・・・・・・・・・・・(6) Here, E
P: Amplitude value of the signal On the other hand, the multipliers 18 and 27 multiply the signal S2 and the output signal of the phase inversion circuit 26 by the output signal of the speed deviation amplifier 6, respectively, so that they are in the same phase as the former signal. and a signal Q expressed by the following equation having a magnitude proportional to the latter signal
1. Output Q2.

. EQ-8IN to Q (ω8 to 10θ)...
(7) Q2 = -EQ-cos (ωst + θ)...
・・・・・・・・・・・・(8) These signals "1 = F
2, Ql, and Q2 are field currents iF1. It is a signal that determines the magnitude and phase of iF2, and according to the operations of part numbers 19 to 24, jF+ is controlled so that it is in phase and has a similar relationship to the signal given by the sum of signals P1 and Ql, and similarly According to the operations of part numbers 28 to 33, iF2 is controlled so that it is in phase and has a similar relationship to the signal given by the sum of signals P2 and Q2.

The operations of part numbers 19 to 24 and 28 to 33 are the same as in the case of a conventional stationary Leonardo, which can control current in the forward and reverse directions, and therefore the explanation will be omitted.

From the above, field current iF1. iF2 is a current that changes sinusoidally, and is expressed by the following equation.

1p1=K (PI + Ql) = 1F-CO3(ωst+θ)+IC-8IN(ωst
10) ・・・・・・・・・・・・・・・・・・・・・
(9)) ip22K(P2+Q2) =Ip・5IN(ωsi+θ)−■c−cO3(ωs
1+θ)・・・・・・・・・・・・・・・・・・
...00) Here, K: proportionality constant ■F: signal P1. Amplitude value of field current based on P2 component ■c: Signal Q1. Amplitude value of field current based on Q2 component At this time, looking at the voltage induced in the armature winding, field current iF1. iF2 generates magnetic flux that interlinks with the armature winding, and by changing the number of magnetic flux linkages,
The voltage shown in the following formula (no-load induced voltage) is applied to the armature winding U phase.
eou induces.

■. As for the W-phase voltage, its phase differs by only 120 degrees from eou, so its description will be omitted.

eou=-(ωs ten ωr)-M-Ip-8IN((ωS
1ωr) to 1θ+ψ1)+(ωS+ωr) −M−I
c - CO3 ((ωS 1ωr) t+θ+ψ1)...
・・・・・・・・・・・・・・・・・・・・・(1υ Here, M: Maximum value of mutual inductance between field winding F, , F2 and armature winding ω, = 2πPM2.P: Number of pole pairs, N3: Number of revolutions per second ψ1: Angle (electrical angle) between field winding F1 and armature winding U phase at time t=0 As shown by the formula, generated by field magnetomotive force The armature voltage eu has an angular frequency of ωS+ω, and even if the rotational speed is zero (ω, 20), it becomes a voltage having the lowest angular frequency of ωS.

This is greatly different from the conventional example (the one in FIG. 1) in which when the rotational speed of the synchronous motor is zero, the frequency of the armature voltage is also zero.

On the other hand, the armature current is controlled as follows.

First, the position detector 14 detects two-phase signals H1 and H2 having phases corresponding to the rotational angular position of the rotor (field) as shown in the following equation.

H1=CO5(ωrt+φ2)・・・・・・・・・・・・
・・・・・・・・・・・・ 02) H2=SUN(ωrt×φ2)・・・・・・・・・・・・・・・03) Here,
φ2: Signal phase current phase command circuit 3 at time 1=0
4 is the output signal S1.4 of the oscillator 15. S2 and the signal H1,
Input H2 and perform the following calculation.

That is, the multipliers 35 and 36 respectively add the signal S1
By multiplying and combining H1, S2 and H2, and extracting the difference between the two obtained signals using a subtracter 39, a signal L1 expressed by the following equation is obtained.

L1=S1・H, -82・H2 1005 ((ωS+ωr) is 10θ+φ2) ...04
) and on the other hand, the multipliers 37 and 38 respectively add the signal S
2 and Hl and Sl and H2 are multiplied together, and the sum of the two obtained signals is taken out by an adder 40, thereby obtaining a signal L2 shown in the following equation.

L2-82・H, 10S1・)! 2 25IN ((ωs+ωr)t+θ+ψ2)...09 Next, these signals L1. L2 is phase number conversion circuit 41 to 43
is added to, and the circuit performs the operation shown in the following equation,
The following three signals (current phase command signal) DU, D■, Dw
get.

As shown in the equations, these signals DU-Dw are sinusoidal signals having the same frequency as the voltage eo described above and having a phase difference of 120 degrees from each other.

Next, a multiplier 8 multiplies the output signal of the speed deviation amplifier 6 and the signal DU to generate an armature current pattern signal that determines the magnitude and phase of the armature current iU.

The current iU is according to the operation of part number 9-12,1.
It is controlled to have the same phase and similar relationship to the armature current pattern signal.

In addition, part numbers 9 to 12.1 (Sirisk circuit UP
.

The operation of UN) can be controlled with respect to the current in the forward and reverse directions.
Since this is the same as in the case of a conventional stationary Leonard, the explanation will be omitted.

The current iU is expressed by the following formula. iU=-Im 5IN((
ωs 1ωr) to 10 + ψ2)・・・・・・・・
(L→Here, Im: Amplitude value of current) Note that the other phase currents iv and iw are only 120 degrees different in phase from iU, so the description above will be omitted.

As a result, the electric motor 13 generates a torque τ expressed by the following equation.

Here, the value of (ψ2 - ψ1) is the value of the position detector 1 described above.
4, by adjusting the relative positional relationship between the Hall generator and the permanent magnet, and adjusting the output signal phase of the position detector 14 corresponding to the rotation angle of the rotor (field).

The torque τ' when set to ψ22ψ1 is given by the following equation.

This equation is similar to the equation for a conventional synchronous motor as shown in FIG. It shows the scales that pull out the torque.

The equation also shows that torque τ' is a value determined only by the magnitude of the armature current and field current Im·IF, and is not affected by the output signal frequency or rotational speed of the oscillator 15.

Next, if we look at the armature terminal voltage, the armature current iu
The flow of −iw causes an armature reaction, so
The terminal voltage is lower than the no-load induced voltage eo by the synchronous reactance drop.

In the end, the terminal voltage eu is eu = e o u−XsImCO5((ωs+ω
r) t + θ + ψ 1) = (ω8 + ωj) ・M・IF
SIN((GJS +(t)r>t+θ+ψ1)+((
ωS tenωr) M・■.

X2 ・Im jcO5(((IJS +cor)
t-% 0+ψ1)・・・・・・・・・・・・・・・
・・・・・・・・・・・・・・・・・・ (221Here,
Xs is the synchronous reactance.

By the way, the amplitude value IC of the field current and the amplitude value Im of the armature current are values proportional to the torque command signal which is the output signal of the speed deviation amplification device 6, and the ratio of the two can be expressed as Ic/Im2Xs /(ωs ten (1) l-)M ++, +
+, (23) (this adjustment can be done by adjusting the input resistance of the current bias A amplifier 10), the terminal voltage eu becomes eu = - (ωS + ωr) ・M・IPS IN (
ωB + ωr) (10θ+ψ1)・・・・・・・・・
・・・・・・・・・・・・・・・・・・).

The formula shows that the terminal voltage will not be affected by the armature current, that is, its magnitude will be determined by the magnitude of the IF commanded by the constant excitation command signal, and will not vary depending on the armature current. and its phase is expressed by equation (11
As shown by the relationship between (ψ2=91) and equation (2), it is always in phase with the armature current (power factor is 1.0).

This means that it is possible to prevent a drop in power factor, a rise in terminal voltage, and the accompanying problems that occurred in the conventional example due to an increase in load. It is clear that: (1) The magnitude of the torque can be arbitrarily controlled by controlling the magnitude Im of the armature current.

Therefore, as shown in FIG. 4, if the armature current is controlled in accordance with the output signal of the speed deviation amplifier 6, the rotational speed of the motor can be controlled in accordance with the speed command.

(2) The frequencies of the armature current and armature voltage have a frequency equal to the output signal frequency of the oscillator 15 ωS even if the rotational speed is zero (○ in ω), and therefore become zero. There isn't.

Therefore, the frequency of the oscillator can be changed to cycloconverter 1
Considering the thermal time constant of the thyristors that make up the
By setting H2), it is possible to prevent current concentration as in the conventional system.

(3) Armature terminal voltage is not affected by armature current.

That is, the magnitude of the voltage and the phase difference (power factor) with the armature current are always kept constant regardless of the magnitude of the armature current.

Therefore, a decrease in power factor and an increase in terminal voltage that occur in the conventional example due to an increase in load are prevented.

Note that the capacity of the field control thyristor circuit is required to be approximately 5% to 10 times the capacity of the cycloconverter 1, but due to (2) and (3) above, the thyristor circuit capacity of the cycloconverter 1 is lower than that of the conventional one. Since this can be greatly reduced compared to , a great effect can be obtained in that the capacity of the SIRISK device can be greatly reduced overall.

According to the present invention, an electric motor that can significantly reduce the installed capacity of a Cyrisk power converter by preventing the current concentration phenomenon observed in conventional systems and power factor fluctuations due to changes in load size. A control device can be provided.

In the embodiment described above, the output signal frequency of the oscillator 15 is
The case where the value is a set constant value is shown.

However, if the rotational speed of the motor is no longer near zero, even if the frequency of the oscillator 15 is zero.

Since the frequency of the armature current is 0S+(')r,
2π No longer causes current concentration.

Rather, in such a case, in order to reduce the power flowing into the field control thyristor circuit, the oscillator frequency ωS
is preferably near zero.

This means that as the rotational speed increases from zero, the oscillator 1
This can be carried out by controlling the frequency of No. 5 to approach zero stepwise or continuously.

To do this, for example, the output signal of the speed generator 4 or the speed command signal is input, and if the value of the signal is below a certain level, the signal at a certain set level is changed to a certain level. In the above case, a function generating circuit that generates a zero level signal may be provided, and the signal of this circuit may be applied to the input of the V-F converter constituting the oscillator 15.

Also, instead of using the output signal of the speed deviation amplifier 6 as the input signal of the multiplier 18.27, a signal proportional to the input current or output current of the cycloconverter 1 is detected and used. Good too.

Furthermore, even if the relationship shown in equation 23 is not completely satisfied,
It is clear that even if the field current Ic and the armature current are in a proportional relationship, it is possible to suppress fluctuations in the power factor and armature voltage compared to the conventional example.

FIG. 7 is a circuit configuration diagram showing another embodiment of the present invention.

Part numbers 5, 6, 8, 15, 16°34 are the same as the parts with the same numbers shown in FIG. 4, so their explanation will be omitted.

In addition, 1.4.9 to 14.21 to 2 of the parts shown in Figure 4
4. Descriptions of the same parts as 30 to 33 are omitted in FIG.

44 and 45 are function generating circuits that receive the constant excitation command signal and the output signal of the speed deviation amplifier 6 as inputs, and output signals having a functional relationship as described later, and 46 to 49 are the oscillator 15 and the function generating circuit. A multiplier 5051 multiplies the output signals of the circuit 45 and 5051 an adder and a subtracter 525 add the output signals of the multipliers 46 and 47, 48 and 49, respectively.
3 is a multiplier that multiplies the output signals of the function generation circuit 44 and the adder 50, and also the output signals of the subtracter 51;
4.55 is a current deviation amplification device which compares and amplifies the output signals of the multiplier 52 and the current detector 19, as well as 53 and 28, respectively.

The operation of this circuit will be explained below.

The function generating circuit 44 inputs the constant excitation command signal BP and the output signal EQ of the speed deviation amplifier 6, and generates a signal ET expressed by the following equation.
Output.

Note that this calculation can be executed by using a plurality of function generation circuits whose input/output relationship is a quadratic curve relationship.

On the other hand, the function generating circuit 45 similarly receives EP and EQ and outputs signals SINδ and cosδ expressed by the following equations.

Next, multipliers 46, 47 and adder 50, and multiplier 48
．． 49 and subtracter 51 perform the calculations shown in the following equations to obtain signals S; , S/, respectively.

3' -S-cos δ+S2・SINδ 1=cos(ωSt+θ-δ)・・・・・・・・・・・・
...(2al, S'2=S2・cosδ−81・
SINδ=SIN(ωst+θ−δ) ・・・・・・・・・
(A) Next, multipliers 52 and 53 output the output signal ET of the function generating circuit 44 and the S1. S
'2 signals are multiplied together and the signal IPt t shown in the following equation is obtained.
IF5.

obtain.

By the way, the value of the signal IPt e IF5 is the sum of the signals P1 and Ql in the above embodiment, and the sum of the signals P2 and Q
2, and according to this signal, the field current IF
These field currents are controlled in exactly the same way as in the previous embodiment except that t t IF2 is controlled.

Therefore, it is clear that the same effects as in the above embodiment can be obtained.

FIG. 8 is a circuit diagram showing another embodiment of the present invention.

The present invention is applied to a so-called DC thyristor motor that uses a thyristor power converter consisting of a first converter 56a that converts AC voltage to DC and a second converter 56b that converts the DC to AC. Shows the case where it is applied.

Even in this case, when a synchronous motor like the conventional example is used, a current concentration phenomenon occurs in the second converter 56b when the rotational speed is zero.

Furthermore, the power factor and armature voltage of the motor fluctuate in response to changes in load.

Therefore, by applying the present invention, effects similar to those of the above embodiment can be expected.

The configuration of this embodiment will be explained below.

Part numbers 4 to 6 and 13 to 34 are the same as those with the same numbers shown in FIG. 4, so their explanation will be omitted.

56 is a Cyrisk power converter consisting of the first converter 56a and the second converter 56b as described above, and 57 is a current detector for detecting a signal proportional to the AC input current of the first converter 56a. , 58 is a current deviation amplifier that amplifies the output signals of the speed deviation amplifier 6 and the current detector 57, and 59 is the firing phase of the first converter 56a according to the output signal of the current deviation amplifier 58. and ignition control of the converter; 60 is a current phase command circuit 34;
a gate amplifier 6 which generates a gate signal for a second converter 56b having a phase relationship as described later from the output signal of the converter 6;
1.62 is a circuit consisting of a capacitor and a resistor provided to facilitate commutation of the second converter 56b.

Next, the operation of this embodiment will be explained.

The input current of the first converter 56a and the armature current of the electric motor 13 which is in a proportional relationship thereto are determined by part numbers 57 to 59.
According to the operation of step 56a, it is controlled to a value commensurate with the output signal of the speed deviation amplifier 6.

This operation is similar to the operation of a well-known thyristor motor device, so a detailed explanation will not be provided.

On the other hand, the gate amplifier 60 receives the aforementioned current phase command signal DU-DW and outputs a gate signal (b in the figure) of the second converter 56b having a phase relationship with it as shown in FIG.

Since the second converter's sirisk is energized in accordance with this gate signal, the armature current of the motor 13 eventually flows in the same phase as the current phase command signal.

As described above, in this embodiment, the magnitude and phase (frequency) of the field current and armature current are controlled in the same way as in the previous embodiment, so basically the same operation as in the previous embodiment is performed. Therefore, it is clear that the same effects as described above can be obtained.

Furthermore, the present invention can be applied to a chopper circuit that converts a constant voltage DC into a variable voltage DC instead of the first converter 56a in the embodiment shown in FIG. 8, and the same effect can be obtained. The gains are clear.

Further, even if the present invention is applied to a so-called AC thyristor motor using a current type cycloconverter as a thyristor power converter, the same effects as described above can be obtained.

Note that the circuit and operation when the present invention is applied are similar to those of the DC thyristor motor described above, and the automatic pulse phase shifter shown in FIG. The only difference is that the gate signal is generated from a signal that is the AND of the output signal of 59 and the output signal of the gate amplifier 60 (in this case, a signal at the logic circuit level), and the operation is completely similar. Therefore, detailed explanation will be omitted.

Furthermore, the motor 13 is not limited to having a three-phase armature winding and a two-phase field winding as in the embodiment, but may also have polyphase windings each having a different number of phases. , there is basically no change and the same effect can be obtained.

[Brief explanation of drawings]

FIG. 1 is a configuration diagram of a conventional device, FIGS. 2 and 3 are waveform diagrams for explaining the operation shown in FIG. 1, FIG. 4 is a configuration diagram showing an embodiment of the present invention, and FIG. FIG. 4 is a detailed configuration diagram of the circuit components shown in FIG. 6. FIG. 6 is a vector diagram explaining the basic principle of the present invention. FIGS. FIG. 3 is a waveform diagram for explaining the operation. Explanation of symbols, 1...Cycloconverter, 4.
... speed generator, 5 ... speed command circuit,
8...Multiplier, 11...Moving pulse phase shifter, 13...Synchronous motor, 14...Position detector, 15...・Oscillator, 16...
Constant excitation component command circuit, 21... Automatic pulse phase shifter, 34... Current phase command circuit.

Claims (1)

[Claims] 1. A Cyrisk power converter that performs AC frequency conversion,
An AC motor having a multiphase field winding driven by the power converter, an oscillator that generates two or more signals with different phases, and a relative positional relationship between the armature of the motor and the field. a position detector that generates a position signal according to the position signal; a speed control circuit that outputs a torque command signal according to the deviation between the speed command value and the actual speed value of the AC motor; a current phase command circuit that generates a phase control signal for the armature current based on the signal; and a field command circuit that receives the output signal of the oscillator and outputs a field current command signal whose magnitude and phase change depending on the magnitude of the torque command signal. an armature current control circuit that controls the thickness of the armature current according to the torque command signal and controls the phase based on the phase control signal; and the field winding according to the field current command signal. A control device for an AC motor that includes a control circuit for controlling field current supplied to a line.