JPS5813159A - Knocking control device - Google Patents

Abstract

PURPOSE:To provide a knocking control device for properly controlling knocking by retarding an amount of knocking signal in proportion to the total pulse width of the number of knocking pulses. CONSTITUTION:A knocking control device consists of a knocking sensor 100 to detect a knocking signal, a knocking control unit 101 generating control signal output for controlling ignition timing of a ignition coil 135 according to knocking signal output from the knocking sensor, a pick-up coil 105 for detecting spark timing of the ignition coil 135 and a contactless igniter 103. The contactless igniter 103 ignites the ignition coil with the outputs of the pick-up coil 105 and knocking control unit 101 while sending feedback signal to the knocking control unit 101 which takes the detecting signal of the knocking sensor 100 and the output signal of the contactless igniter 103 to control the unit 103 and carry out contrably angle advancing or retarding in response to knocking.

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a knock control device, and more particularly to a knock control device that does not generate knock-like noise and affect engine characteristics.

Knock that occurs in an engine causes an unpleasant knocking sound that reduces running performance, and also causes a reduction in engine output due to the generation of reverse torque or damage to the engine due to overheating. This knock has a close relationship with the ignition timing, and it is known that adjusting the timing of the engine can maximize engine output. Therefore, as a result of reducing the ignition advance angle to avoid the occurrence of knock, it will also reduce the engine output. Optimal ignition timing is required to maintain efficiency.

In conventional knock control devices, a method is used in which ignition noise caused by the ignition is masked from among the signals from the knock sensor, a knock signal is detected, and the engine is retarded by a certain amount based on the knock signal. In other words, it retards a certain amount (for example, 0.4 degrees) due to the knock that occurs during the ignition timing, and determines whether or not to retard depending on whether or not there is a knock signal. A system was adopted in which the engine was retarded by the following steps (so-called one-start retard).

However, with the conventional method, when knocking occurs due to coupling, the retard amount is small (one step), so the control response to the knock is poor, and the running performance is extremely poor. had. To solve this problem, a method of increasing the amount of retard per step has been considered, but this has the disadvantage that the amount of retard is large with only one knock signal, and the torque is not sufficient.

An object of the present invention is to provide a knock control device that can appropriately control knocking.

The present invention attempts to appropriately control knocking by starting the number of knocking pulses of a knocking signal in proportion to the total pulse width.
゛Hereinafter, embodiments of the present invention will be described.

FIG. 1 shows the overall configuration of a knock control device showing one embodiment of the present invention.

In the figure, the knock control device of the present invention includes a knock sensor 100 for detecting a knock signal, a knock sensor 10
The ignition coil 135 is activated by the knock signal output from
A knock control device 101 outputs a control signal to control the ignition timing of the ignition coil 135, a pickup coil 105 detects the spark timing of the ignition coil 135, and the output from the pickup coil 105 and the knock control device 101 causes the ignition coil to ignite. It also includes a non-contact ignition device 103 for sending a feedback signal to the knock control device 101.

The knock control device 101 takes in the detection signal of the knock sensor 100 and the output signal of the non-contact ignition device 103, and controls the non-contact ignition device 103 in response to knocking to perform advance or retard control.

The knock control device 101 includes a failsafe device 102 that detects a failure of the knock sensor 100 and sends a signal to forcibly retard the ignition timing, an AC coupling circuit 112 that removes the DC component, and a device that synchronizes with the spark timing. an ignition noise cut circuit 113 having a gate for cutting ignition noise; a bandpass filter (BPF) 114 for bandpassing the knock signal; BPF1;
Automatic gain control circuit (AGC circuit) 1 that controls the gain of its own amplifier in proportion to the input signal ratio by the output of 14.
60, AGC output (a mask circuit 150 for masking a predetermined evening timing section; pack ground level (BGL) detection for obtaining the average value of the knock signal by inputting from the AGC circuit 160 via the mask circuit 150; The circuit 119 amplifies the output of the BGL detection circuit 119 and outputs the AG signal.
Amplifier 16 for feeding back to C circuit 160
1. An amplifier 151 for amplifying the output of the mask circuit 150.
A comparator 118 that compares the output voltage of the BGL detection circuit 119 with the output signal of the amplifier 151 and generates a retard signal proportional to knocking, and a mask that masks the output of the comparator 118 at a predetermined timing and outputs it. A circuit 153, an integrator 154 that integrates the output of the mask circuit 153, a retard signal setting circuit 126 for setting a retard signal proportional to knocking based on the output of the integrator 154, and a signal from the non-contact ignition device 103. The monostable circuit 128 generates a signal with a constant pulse width in synchronization with the interruption of the ignition coil 135 (that is, in synchronization with the base current of the power transistor 134), while the output pulse of the monostable circuit 128 is output. An advance angle signal setting circuit 127 that outputs a signal for advancing the ignition timing of the ignition coil 135 at a constant voltage/cycle rate, a retard signal set in the retard signal setting circuit 126, and an advance angle signal setting circuit 127. an integrating circuit 125 that generates a DC voltage proportional to an advance signal set to , and sends out an output for advancing or forcibly retarding the ignition timing of the ignition coil 1350 using the output from the failsafe device 102 ; It consists of a reference voltage generation circuit 120. Furthermore, this embodiment includes a rotation speed detection circuit 156 for setting the mask timing of the mask circuits 150 and 153.

The reference voltage generation circuit 120 includes an operational amplifier OP3, a capacitor C5, resistors R, 15, R, 11, R, 12, and R14.
Consists of. By appropriately selecting resistors R14 and R15, the output RV of the operational amplifier OP3 is fixed to the reference voltage. This reference voltage RV is, for example, 3V, and is used for various reference voltages.

The fail-safe device 102 has a seven-work safety function that detects various failures or abnormal mode numbers of the knock sensor 100 and performs retard control according to the mode.

This device 102 includes a constant current circuit 107, at least a sensor short detection circuit 108, and a sensor open detection circuit 1.
The sensor abnormality detection circuit 102A has a sensor abnormality detection circuit 102A having a sensor abnormality detection circuit 102A, and the output of each circuit is supplied to 12 color integrating circuits, thereby achieving a fail-safe function. The BGL detection circuit 119 is composed of a half-wave rectifier 116, an integrating circuit 117, and an amplifier 117A, and receives the output signal of the A G, C" knee path 160, which is transmitted through the mask circuit 150. Rectification and integration are performed to obtain an averaged signal of the entire signal. This averaged signal becomes a BGL signal. Comparison circuit 118 has at least two comparators 118A and 118B. Comparator 11
8A compares the BGL output with the output of the amplifier 151, and the comparator 118B compares the output of the amplifier 151 with a reference voltage.

The non-contact ignition device 103 includes an amplifier 131 that shapes the waveform of the output signal of the pickup coil 105, a retard circuit 132 that controls the ignition timing according to the output voltage of the knock control circuit 101, and a high voltage generated on the secondary side of the ignition coil 1350. It consists of a power transistor 134.

Next, the operation of the above configuration will be explained based on the waveforms shown in FIG. FIG. 2(1) shows an ignition timing waveform, and this waveform signal is actually a pace signal of a power transistor 134 of a non-contact ignition device 103, which will be described later. At H level, the power transistor 134 is turned on (ON), and at L level, the power transistor 134 is turned off (OFF). Sparks in the ignition coil 135 are generated during the switching process from ON to OFF. The signal S2 in FIG. 2 (2) is based on the above.

Monostable circuit 12 which inputs a signal and generates a pulse signal of a constant width (tl) when triggered when the signal changes from ON to OFF.
8 constant width pulse output signal.

Figure 2 (3) shows the upgraded knock sensor 100.
shows the output of The signal detected by the knock sensor 100 is a signal that swings positive and negative with the DC zero (0) level as a reference. The detection signal includes a knock signal.

On the other hand, in order to enable failsafe, the detection circuit 108,
109 has been established.

In the failure mode of these detection circuits, the mode cannot be detected if the DC zero level signal remains.

As a countermeasure against this, a constant current circuit 107 is provided. A constant current from the constant current circuit 107 is superimposed on the output of the sensor 100, thereby providing a constant positive bias. This constant current circuit 107 plays an important role. That is, the output of the knock sensor 100 changes between ±5 mV and ±600 mV. The range of change is 120 times. When attempting to attenuate the knock sensor output using general bias means using resistive voltage division for a 120 times change width, for example, if the knock sensor output is close to ±5 mV and the knock sensor output by the bias means is If the attenuation rate is reduced to 1/20, the output after bias will drop to 0.25 mV. This obviously leads to a decrease in detection accuracy, which in turn makes it impossible to detect the failure mode. By providing the constant current circuit 107, the present invention improves knock detection accuracy and facilitates failure mode detection. FIG. 2(3) shows the sensor output biased by the constant current circuit 107.

AC coupling circuit 112 is provided to remove DC components. Furthermore, it may have a role of causing resonance in the knocking frequency region. By removing the DC bias in the AC coupling circuit 112, it makes a significant contribution to improving the noise cutting performance of the ignition noise cutting circuit 113 in the next stage.

This is because ignition noise cannot be cut unless the ignition noise is cut after removing the DC bias. Ignition noise gas':・''1t circuit 113 is monostable circuit 1
It is controlled by a pulse output as shown in FIG. 2 (2) of 28. The noise cut circuit 113 is a type of AND gate, and inputs the output pulse of the monostable circuit 128. Therefore, only in the monostable circuit, the sensor output obtained via the AC coupling circuit 112 becomes the output of the ignition noise cut circuit 113 as is #1. This output is shown in Figure 2 (
4). Ignition noise is cut depending on the weather. The BPF 114 emphasizes the knock signal (reduces other signals) and outputs it, and has a characteristic of being slightly attenuated at a higher frequency than the knock signal of knocking. AGC circuit 160 connects BG via amplifier 161.
Upon receiving the feedback signal from the L circuit 119, it changes its own gain in inverse proportion to the feedback signal, ie, the BGL output. The mask circuit 150 masks the output of the AGC circuit 16'0 at a predetermined timing. This mask uses the pulse signal S2 in Fig. 2 (2).
done by. The output of this mask circuit 150 is
At 1 digit, the BGL detection circuit 119 performs BGL detection.

The comparison circuit 118 compares the BGL output (voltage) of the BGL detection circuit 119 with the output of the amplifier 151 using the i118A, and compares the output of the amplifier 151 with the reference voltage using the comparator 11.
Do it in 8B. The state of both signals at this time is shown in FIG. 2 (5). The comparator 118A of the comparison circuit 118 shapes and outputs only the output of the amplifier 151, which is larger than the BGL output. Further, the comparator 118B removes the output of the amplifier 151 at a level higher than the reference voltage applied to the negative terminal (this voltage is a reference value for limiting the abnormal amplitude of the knock signal). - This clamps abnormally high voltages in the knock signal. The output of the comparator 118 is input to a mask circuit 153 and a retard signal output circuit 126 and converted to a DC level, and the DC level signal becomes the knock signal to be obtained.

Here, the mask circuit 153 has the role of removing a sound similar to a knock signal generated from the intake cylinder during high-speed rotation.

This mask signal 810 is generated by the constant time width signal generation circuit 157.
is formed. Furthermore, the mask is also performed by 82. The output S6 of the comparator 118 is in the form of a single pulse, and the function of averaging this single pulse is provided by the retard signal output circuit 12.
6 has.

FIG. 2(6) shows the knock signal detected by the comparator 118. FIG. 2 (7) is an enlarged view of a portion thereof.

Integrating circuit 125 receives the output signal of retard signal output circuit 126 and performs predetermined integration. Therefore, an integral value corresponding to the number of knock pulses is outputted from the integrating circuit 125. The retard angle control of the retard circuit 132 is performed using this integral output.

FIG. 1 shows a specific circuit example of the configuration from the knock sensor 100 to the BPF 114. In the figure, a knock sensor 100 is an inductive sensor using a magnetostrictive element, and isometrically constitutes a series circuit of an inductance L and a resistor R2. Generally, the value of inductance L is I
H, the value of the resistor R2 is approximately 840Ω.

The constant current circuit 107 includes a resistor R1 and a Zener diode Z.
D1 and transistor T1. Power for the circuit 107 is supplied from a power supply section 120.

The power supply unit 120 includes resistors R11, R12, R14, and R1.
5, a capacitor C5, and an operational amplifier OP3. The operational amplifier OP3 plays the role of a buffer, and the resistors R14 and R1,
5 and is applied as a partial pressure.

The output of operational amplifier OP3 is set to 3(v) and PNP
The collector current of transistor T1 is approximately 1.7 mA
is set to Therefore, the DC bias voltage at point a is
The length is approximately 1.6m, and the knocksen is connected to the 1.6(V) voltage.
The 100 output signals will overlap.

The reason for adopting the constant current circuit 107 will be described below.

The output impedance Z of the knock sensor 109 is determined by the values of the resistance R2 and the inductance L. Therefore, the value of impedance Z at the knocking frequency (approximately 7 KHz) is as follows: Z=2xfL+R2=2XX7000X1+840=45(KΩ). Therefore, if the input impedance of the knock sensor connection portion of the knock control device is low, the knock sensor output signal will be significantly attenuated. Knock sensor output is ±5 (mV)
It varies between a 120-fold range of ~±600 (mV). A constant current circuit 107 is provided to increase input impedance. In particular, in the structure shown in the figure, since the DC bias circuit has a constant current configuration, the impedance of the DC bias circuit has a value close to infinity.

On the other hand, if the input impedance of the knock control device is increased, disturbance noise is likely to be superimposed. An unusual type of disturbance noise is ignition noise (Ig noise) that occurs in synchronization with the ignition timing. -Next, ignition noise will be explained.

Base control of the power transistor is performed by pulses as shown in 2nd M). When the pulse is at H level, the power transistor is turned on (ON) and when it is at L level, it is turned off (OFF). During this process of switching from ON to OFF, or when the ignition coil turns OFF, the ignition coil, 1 of 2
The secondary voltage rises rapidly, and primary noise is generated. Furthermore, this rise in secondary voltage breaks down the insulation of the air layer between the plugs, causing ignition. Secondary noise occurs during this ignition. The second-order noise includes noise due to a capacitive discharge current flowing at the initial stage of ignition, and noise due to an induced discharge current flowing at a subsequent stage. Among the second-order noises, the former noise is a major noise source. When the input impedance is increased, primary noise and secondary noise (the former noise) are superimposed on the knock sensor output as disturbance noise that adversely affects knock signal identification.

It is necessary to remove such disturbance noise. This disturbance noise continues for a time of about 50 to 60 μ(8). Therefore, during this period, it is only necessary to "mass the knock sensor output." To achieve this purpose, the AC coupling circuit 112
, and a noise cut circuit 113 are provided. However, the actual mask section is set to a time width that is sufficiently larger than the noise duration time, for example, about 0.8 m sec.

AC coupling circuit 112 consists of capacitor C1 and resistor R3. The noise cut circuit 113 includes resistors R4, R5, and R6.
, R8, R223, capacitor C2, transistor T2
．． It consists of an operational amplifier OPI. The AC coupling circuit 112 is provided as a means for extracting the knocking signal from the knock sensor output signal in a good manner, and by passing the knocking signal through this circuit 112, the DC bias voltage riding on the knock sensor output signal is removed. If you try to extract only the knock signal from the knock sensor output obtained by superimposing the DC bias component, and if you try to apply the above-mentioned noise mask, the processing becomes extremely complex. Although the idea of cutting off direct current is simple, it is an extremely practical technical means for correctly classifying knock signals.

The noise cut circuit 113 mainly includes a transistor T2.
Ig noise is cut by the function of . Transistor T2 is turned on and off by output S2 of monostable circuit 128. The monostable circuit 128 is triggered by the fall of the base signal of the power transistor shown in FIG. 2(1), and generates a pulse having the width of the mask section. Figure 2 (2
) is the output S2 of this monostable circuit 128, and the time width t
1 is the mask section width. - Transistor T only in the period t,' where the output S2 of this monostable circuit 128 becomes "1"
Turn on 2. As a result, in this period t1, the knock sensor output is short-circuited to ground, and the input to the operational amplifier OPI is eliminated, producing a masking effect for masking engineering noise. As the load impedance of the sensor 100, resistors R3, R4, R5, and capacitors C1 and C2 can be considered, but by setting the resistor R5 to a high resistance, for example, IMΩ, the resistors R4, R5, and the capacitor C2 can be The load can be ignored. Therefore, the load on the sensor 100 can be considered to be only the capacitor CI and the resistor R3.

Therefore, in addition to the role of DC removal, the AC coupling circuit 112 can also serve as a resonance/circuit means for emphasizing and extracting the knock signal from the knock sensor output. This will be explained below. ,
,・:2,.

As mentioned above, by setting the impedance resistor R5 as a load when looking at the knock control device from the knock sensor 100, the -sa C2 can be ignored as a load, so only the AC coupling circuit 112 can be used. Become.

Therefore, the resonant circuit is formed by a circuit formed by the sensor 100 and the AC coupling circuit 113, which includes the six currents L, the resistor R1, the capacitor C1, and the resistor R3.

The resonant circuit has a frequency of 3 to a knock signal ($J7KHz).
It can function as a bypass filter that damps engine vibrations of ~5KHz (approximately 5dB). The gain G at this time is as follows.

A□ Figure 4 shows the characteristics of the resonant circuit obtained in this way.
・It has been done. Figure 4 shows the characteristics of the Innox filter when the capacitance of capacitor C1 is 100PF.
This is the characteristic when the size is set to 00OOPF, and shows the characteristic of a low-pass filter. Figure 4 (B) shows the capacitor CI
This is the characteristic when the size is set to 470PF, and shows the characteristic of a bandpass filter. The resonant circuit actually formed has characteristics similar to that of a bandpass filter shown in FIG.

In addition, the value of capacitor C1 is 470PF, and the value of resistor R3 is 20
When set to Ω, the resonant frequency was 7 KHz and the physical properties of a bandpass filter were exhibited.

BPP 114 consists of resistors R7, R9, RIO, capacitors C3, C4, C22, and operational amplifier OP2. The non-inverting terminal of the operational amplifier OP2 is connected to the output RV [3V reference voltage] of the operational amplifier OP3 of the power supply unit 120 that functions as a buffer. Therefore, it is known that the output of BPF1, 14 is DC-biased to the reference voltage. This BPF 114 has a filter function to increase the level difference between the knock signal and the non-knock signal.

This facilitates identification by level.

The output S1 from operational amplifier OP2 of BPFI 14 is input to resistor R16 of their AGC circuit 160.

Next, the operation of the knock sensor failure detection circuit 102A will be explained. The sensor short detection circuit 109 consists of a comparator CO2. The sensor open detection circuit 108 includes a resistor R5.
5. It consists of a comparator C01. ``Furthermore, the potential at point a of comparator CO2 is set to approximately 0.2 (V), and the potential at point b of comparator COI is
It is assumed that the point potential is set to approximately 2.8 (V).

Now, if the knock sensor 100 is short-circuited, the potential at point a becomes approximately 0 (V). Therefore, the potential at point a becomes lower than the potential at point a, and the output of comparator CO2 becomes approximately 0 (V). When the knock sensor 100 is normal and does not short-circuit, the potential at point a is 5. Since the voltage is also set large, the output of the comparator CO2 is approximately at the point a potential (
In other words, it means greater than 0■). On the other hand, when the knock sensor 100 is open, the voltage at point a becomes approximately 3.2 CvI, which is also higher than the voltage at point b. Therefore, the output of comparator COI becomes 0 (V). In other words, when the knock sensor is disconnected or short-circuited,
The g-point voltage becomes 0 (V). These results become a signal s3 from one transistor T80, and the output to the integrator 125 becomes 0 (V). Thus, the comparators CO2, COI
can output short and open faults.

BPFI 14 (D output s) is input to the AGC circuit 160. The output of the AGC circuit 160 is input to the mask circuit 150.
It is divided into two systems through. The first system is a system consisting of an amplifier 151 that amplifies the knock signal and inputs it to one input terminal of the comparator circuit 11/81. The second system includes a half-wave rectifier circuit 116 and a maximum clamp circuit 116.
A, a BGL circuit 119 consisting of an integrating circuit 117 and an amplifier 117A.

The output of the amplifier 117A is input to the other input terminal of the comparison circuit 118. The output of the amplifier 6117A is the amplifier 161.
The negative feedback knock sensor output swings from ±5 (mV) to 600 (mV) to the AGC circuit 160 via the negative feedback knock sensor. If this output is simply amplified (e.g. 100 times),
It becomes ±0.5 (g) to ±60 (V). However, in a car, the maximum battery voltage (approximately 12 (V), 60
(V) cannot have a value.

Therefore, in the past, either a smaller gain was used to prevent saturation, or a process was performed with the expectation that saturation would occur. The former has the disadvantage that it is less sensitive to small-scale human forces, and the latter has a disadvantage that it is less sensitive to large-amplitude human forces. 9 In the configuration of this embodiment, the AGC circuit 160 is provided, and furthermore, this AGC circuit 160 is connected to the BPFI
It is characterized by being provided on the output side of 14. With this configuration, the level difference between the knock signal and the non-knock signal becomes large in the BPFI 14, and since this sharp level difference is input to the 400 circuit 160, the S/N
A good output ratio can be obtained.

As is clear from the input/output characteristics shown in FIG. 6, the AGC circuit 160 has low level input (±VL) and high level input (
±VH) case! Except for this, the output v0 can be controlled to be substantially constant.

A specific example of a circuit from the AGC circuit 160 to the comparison circuit 118 is shown in FIG. The AGC circuit 160 includes resistors R16, R
17, R19, R18, R192, R102, and operational amplifier OF2 FET T17. Half-wave rectifier, path 116, capacitor C6, resistors R20, R,21
, R, 22, diode D, , D, , operational amplifier O
Consists of P4.

The clamp circuit 116A includes resistors R25, )'L26, R
27, R29, operational amplifier OP6, comparator c.

5. It consists of a capacitor C8 and a diode DIO. Integrator 117 consists of resistor R29 and capacitor c9.

The amplifier 117A has resistors R30% R31, R32, R3.
3, R, 34, R35, and an operational amplifier OP7.

The amplifier 161 includes an operational amplifier OP8, resistors R, 93, and R
37, R38, R36, R40, R39, capacitor C
Consists of IO.

The operation of the circuit shown in FIG. 7 will be explained. BPF114
The filtered output s1 is input to the AGC circuit 160 (DOP3) via the resistor R16.

Ru. An FET (T17) whose gain is controlled via an amplifier 161 is provided at the negative end of OF2. As a result, the gain of the AGC circuit 160 is
It is changed according to the output of OF2 of 17A. The output of the AGC circuit 160 is masked at a predetermined timing by the mask circuit 150, and is input to the half-wave rectifier 116 via C6 and R20. Half-wave rectification of only the positive direction component is performed by the action of the diodes DI and D2, and input to the maximum clamp circuit 116A.In this clamp circuit 116A, the predetermined maximum value (via the positive terminal (e.g., 5'V).

The output of this clamp circuit 116A is integrated and smoothed by an integrating circuit 117 formed of a resistor R29 and a capacitor C9, and further amplified by an amplifier 117A and output. Next, it will be explained in detail.

AGC circuit! 60 input voltage (i.e., BPF114G output voltage) is VI, the output voltage is Vo, and F'ET(T
Since the value of the resistor R19 is set to a high resistance, the amplification factor of the patented operational amplifier OP3, which is characterized by an output impedance of 17), is a value determined by ZF and the resistor R,19. That is, the gain G is as follows. The important point in equation (3') is that the impedance ZF
is the gate-source voltage V o s of FET (T15)
It changes depending on the situation. For example, s Vos is 0 (V
) to −2M, Zr increases and the gain G decreases due to the output feedback of the amplifier 161. If it gets bigger, the opposite is true. As a result, the AGC circuit 160
The BGL output voltage output from the amplifier 117A is AG
It remains constant regardless of fluctuations in the input voltage to the C circuit 160, and the 8N ratio becomes nearly constant.

In the range of Vcsw±200 (mV), AGC is fully charged.
・ Work for ′ minutes. -- Incidentally, the resistor R19 is for maintenance purposes in the event of a disconnection failure of the FET (T15), and is set to a high resistance. Furthermore, Zy is set to a value of about 2 Ω from 200 Ω.

The mask circuit 150 consists of a transistor T15 and a resistor R97. Transistor T15 is rendered conductive by signal S4 or S2 applied to its base.

Due to conduction of transistor T15, the output of AGC circuit 160 drops to ground potential and is masked.

The resistor R95 of the amplifier 151 performs 360 (mV) DC correction on the output of the operational amplifier OP9.

Furthermore, the resistor R35 and capacitor CIO have a ripple prevention function for BGL.Furthermore, throughout the entire i, the operational amplifier (OP) is Hitachi's HA 17902, comparator (
CO) uses Hitachi HA17901 and amplifier 161
When there is no BGL input, the Φ terminal of the operational amplifier OP8 is connected to the output terminal (3■) of the operational amplifier OP3, so the terminal input is 3■.
At this time, the output v2 of the operational amplifier OP8 has a DC correction function that sets it to 4 (V) as shown in FIG. Furthermore, the operational amplifier OP8 is an inverting amplifier, and as the input (1) increases, the output V decreases. This output V2 is the FET of the AGC circuit 160.
Since it is the gate input of (T15), when BGL increases, ■ increases, ■2 lowers the gain G, and A
GC works. Here, the AGC circuit 160 has Vx=3(V
Starting from 1, ZF remains constant between 4CV) and 3(V:).

Furthermore, the knock signal detection sensitivity is adjusted by the transistor TO1 resistors R215 and R216. In other words, when the rotation speed exceeds a predetermined number (for example, a o o o rpm) detected by a rotation speed detection circuit 156, which will be described later, the transistor TO turns on, and the output voltage from the operational amplifier OP9 reaches the voltage across the resistors R216 and R215. Go down in proportion. This is the same as lowering the bias voltage of the knock signal, and the sensitivity of the detection level deteriorates. That is, knock detection sensitivity at high speeds is lowered to improve running performance.

In this way, the SN ratio for knock detection at high speed and low speed is changed.

The operational amplifier OP6 of the maximum clamp circuit 116A is a buffer provided for impedance exchange, and when the non-inverting terminal voltage V, attempts to exceed 5μ, the comparator CO5
conducts, current is discharged through the diode DIO1 resistor R28, and as a result, the terminal voltage of the non-inverting terminal of the operational amplifier OP6 is clamped to a maximum of 5 (V). As a result, the inverted output voltage (background voltage level) of the comparator CO5 does not change depending on the intensity of knocking, and there is no possibility that BGL will rise at the time of knocking, making it difficult to detect knocking.

The comparison circuit 118 includes comparators 118A and 118B.

It consists of resistors R, 43, R, 59, and diode D3.

Comparator CO4 of comparison element 118A receives a signal including the knock signal which is the output of amplifier 151, and resistors R, 41 and R42.
A comparison is made with the divided reference voltage.

The reference voltage is set to 4 (g), for example. On the other hand, comparator CO3 compares the BGL output and the output of amplifier 151. As a result, from the output of the comparison circuit 118, the amplifier 15 which is higher than the reference voltage and higher than the BGL output
1 output S6 outputs.

The f#J% diode D3 has a role of transmitting this output, and the resistor 1't59 has a role of a discharging resistor between it and a capacitor C13 of a retard amount setting circuit 126, which will be described later.

The output S6 of the comparison circuit 118 is sent to the next stage mask circuit 153.
Enter.

Figure 9 shows the physical circuit configuration.

The lead angle signal setting circuit 127 is a fixed lead angle setting circuit 127A.
and a variable advance angle setting circuit 127B.

The fixed lead angle setting circuit 127A includes resistors R56, R68, and R7.
0, R, 98, R, 99, R100, transistor Tl
Consists of l. The variable advance angle setting circuit 127B includes a resistor R65,
It consists of R69 and transistor T8.

The advance angle output signal of the fixed advance angle setting circuit is determined by the power supply voltage of the advance angle reservoir at the time of starting. Variable lead angle setting circuit 127B
The advance angle output signal is determined by the output S9 of the monostable circuit 128. The output S9 is a signal with a pulse width proportional to the rotation speed.1. Therefore, the advance angle output signal of the circuit 127B becomes an advance angle signal proportional to the rotation speed. Lead angle signal setting circuit 1
The advance angle output signal of 27 becomes the integrator 1400 Å force.

The output of the comparator 118 may be directly applied to the integration circuit 125, but it is applied to the integration circuit 12 via the retard signal output circuit 126.
5 may be applied. Furthermore, a retard signal setting circuit 126
By providing a mask circuit 153 between the comparator 118 and the comparator 118, an accurate retard signal can be formed. This example is disclosed in this example.

The mask circuit 153 includes a resistor R57, TL58, and a transistor T7. The retard signal output circuit 126 smoothes and integrates the output pulse of the comparator 118 that has escaped the mask, and performs the corresponding retard control.

- The angle signal setting circuit 126 includes a capacitor C13, a resistor R
63, R64゛, R66, R67, transistor T9,
Consists of TIO. Transistor T7 receives signal S2 or 81
The output S6 of the comparator 118 at this time flows to the ground via the transistor T7 and is output to the ground. When transistor T7 is off, capacitor C1
3, the signal S6 is accumulated and drives the transistor TlO via the resistor R63. Transistor TIO is also driven by signal S3 via resistor R64.

Signal S2 is the output of monostable circuit 128. Signal S3 is the output of abnormality detection circuit 102A. Transistor T9
The power supply voltage applied to the pace of the power supply circuit (described later)
Receive supplies from. When starting the engine, the battery voltage drops below a predetermined minimum allowable voltage. The same applies when the battery capacity becomes low. When this abnormal voltage drop occurs, the voltage is high, and when the voltage is normal, it is low. When the voltage is high, transistor T9 turns on,
The transistor remains off regardless of the signal applied through resistors R63 and R64. On the other hand, when the voltage is low, the transistor T9 is turned off, and as a result, the transistor TIO is turned on and off depending on the voltage value via the resistors R63 and R64.

The integration circuit 125 includes an integrator 140, a maximum voltage clamp circuit 141, and a minimum voltage clamp circuit 142. The integrator 140 includes an operational amplifier 0P15, a capacitor C14,
C15 and resistor R200. The maximum voltage clamp circuit 141 includes an operational amplifier 0P16, resistors R71, R74, and R
75, R76, R7'? , R78, diode D7, and transistor T20. Minimum voltage clamp circuit 14
2 consists of an operational amplifier 0P17, resistors R60, R61-, and a diode D6.

Next, the circuit operation of the integrating circuit 125 will be explained.

Now, the transistor T
IO is turned on in synchronization with the knock signal. Therefore, as shown in FIG. 2 (7), the pulse width t of the knock signal. (about 4
Between 0 and 70μ), the transistor TIO is conductive;
Current i1 flows from operational amplifier 0P15 to capacitor C14,
C15, resistor R, 67,) flows to ground via transistor TIO. Further, the output voltage of the operational amplifier 0P15 at this time is 3 (v).

Therefore, the voltage increase rate per pulse (voltage rise/1 pulse) Δvl of the operational amplifier 0P15 at this time is as follows.

However, the capacitance C is the series capacitance value of the capacitors C14 and C15. As is clear from equation (5), the output voltage of the operational amplifier 0P15 increases in proportion to the number of knocking pulses.

On the other hand, in every cycle, the inverted output 8. of the monostable circuit 128.
9 is applied from transistor T6 to the pace of transistor T8, and for a constant mask time 11, transistor T8
Turn off. Therefore, during this period, a current i flows from the power source 1 to the operational amplifier 0P15 via the resistors R98 and R100 and the capacitors C14 and C15. - The Zener voltage of Zener diode Z-4 is 6M. Furthermore, the e terminal of the operational amplifier 0P15 is at 13 volts. Therefore, every time one pulse is input from the monostable circuit 128 to the operational amplifier 0P15, the operational amplifier 0P
The output voltage of No. 15 will fall according to the following voltage fall rate (falling voltage value/cycle) Δ■1.

Therefore, Δv, = −t, ......
... The voltage drop rate ΔV'' in (7) is set to approximately 1150 of the voltage rise rate Δ■, taking into consideration the power performance such as engine torque and horsepower.The output of the integrator is The value is clamped by the clamp voltage of the maximum clamp circuit 141, and its minimum value is clamped by the clamp voltage of the minimum clamp circuit 1420'.

The integrating circuit 125 is configured to have a specific lead angle characteristic (lead angle value) by turning on the transistor Tll due to voltage increase when the engine is started. This lead angle characteristic is determined by the integrator circuit 125 and the retard circuit 132.
performs actual advance (retard) angle control. This retard circuit 132 is described, for example, in the following document (U, S, Pate
nt application, ser.

No, 80202 t by Noboru Su
giura, filedoctober 1
, 1979 and assigned to
theasignee of this app
lication % ■gnition timing
control system for 1n
internal &ombu-%1ion engine
e // ) The one shown is used.

Here, the operation of the retard circuit 132 will be explained.

In general, one ignition timing characteristic is relative and determined according to a certain operating mode determined by the distributor and ignition system being used. Additionally, a maximum retardation characteristic is given during knocking, and this characteristic is used during knocking. Figure 9 shows the lead angle retardation characteristics, and the actual minimum retard angle (i.e., minimum clamp voltage) characteristic in a certain driving mode, and point # is the maximum retard angle (i.e., maximum clamp voltage) characteristic during knocking. ) shows the characteristics. At low speed, e.g. 200
Below the rpm, the maximum advance angle characteristic determined by the ignition timing characteristic is adopted.The reason why the control characteristic is adopted is to reliably achieve starting at the time of start-up. That is, at the time of starting, if the ignition timing is delayed, the engine generates reverse rotation torque, and the load on the starter becomes extremely large. As a result, the driving current of the starter becomes abnormally large, making it impossible for the starter to start the engine, resulting in a so-called starting failure.

In order to eliminate such startup failures, for example, 200
Below rpm, the maximum advance angle characteristic is determined by the ignition timing characteristic.

FIG. 11 shows the characteristics of the retard circuit 132 that should achieve the above characteristics. As shown in the figure, the output of the integrating circuit 125,
That is, it has a retard characteristic that is a constant angle slope characteristic with respect to the output voltage of the integrator 140. Therefore, the advance angle is a constant angle every cycle. That is, the ignition timing is configured to advance by a constant angle every cycle while being retarded in accordance with the number of knocking pulses.

Next, the integration circuit 1 that controls the retard circuit 132
The following describes the operation of No. 25, especially the countermeasures against starting by using a special starting angle. This start-up countermeasure is related to the power supply circuit, so
First, the power supply circuit shown in FIG. 12 will be explained.

This power supply circuit includes a start detection power supply device 50. It consists of an actual power supply device 51. Power supply device 50. Resistor R86, R8
7, R88, R,89, Zener diode ZD3, capacitor c19, and diode D9. Power supply device 5
1 are resistors 9o, 91, capacitors C20, C21, Zener diode ZD4. Consists of ZD5. The battery power supply is connected to the BW terminal, and voltages higher than the Zener die constant voltage (6.2V) are cut off, so that B=6.
2v is output. - Voltages A and D reflect starting detection. Therefore, the battery voltage decreases during startup. When the amount of decrease exceeds the reference value, transistor T1
4, the off-position voltages A and D have the same value. The same operation occurs when the power capacity of the battery decreases. If the battery power supply voltage is normal, transistor T! 11:4 is on, the D voltage is approximately at ground potential, and the point voltage is equivalent to a drop voltage due to resistor R86. The resistor R86 is set to a relatively high resistance (2iKΩ). This D voltage is the pace of transistor T9, transistor T, +11
is applied to the pace of the engine, and sets a predetermined advance angle characteristic at the time of starting.

Further, T is a terminal connected to the engine cooling water temperature sensor inch. In order to turn off the transistor T14 when the temperature detected by the engine coolant temperature sensor is below a certain value (for example, 70°C), the resistor R22 is
1. A diode D12 and a Zener diode ZD4 are provided.

Integrating circuit 12 that controls next retard circuit 132
Let's talk about the operation in step 5, especially the start-up countermeasure that involves special start-up angle. The Zener diode ZD3 has about 6 (V) of Zener! When the power supply voltage (V) is low, that is, when starting the engine with the starter on, resistors R88 and R89
The midpoint voltage of can no longer turn on the Zener diode ZD3. Therefore, 1::'5° transistor TI4 is turned off, and transistors T9 and TI
O turns on. At this time, transistor TIO is turned off. Further, when the transistor Tll is turned on, a current flows in the same direction as the voltage aLt through the resistor R70, and the output of the operational amplifier 0P15 decreases to the same voltage as the point voltage and is clamped. The voltage at this point corresponds to the minimum clamp voltage of 1.5 (V) shown in FIG. This clamped output sets the maximum retardation characteristic at the time of starting as shown by the dotted line in FIG. As a result, the retard circuit 132 is controlled and set to the maximum retard characteristic.

Next, the operation of the integrating circuit 125 when the sensor 100 fails will be explained. The outputs from the sensor short detector 108 and the sensor open detector 109 shown in FIG. 3 turn off the transistor T7 shown in FIG. 8, and turn on the transistors TIO and T20. When the transistor TIO is turned on, the current 11 continues to flow through the capacitors C14 and C15 in the same way as in the retard operation described above, so the output voltage of the operational amplifier 0P15 is clamped to the voltage of 6V (maximum voltage), which is the same as the h-point voltage. Ru. Furthermore, as a result of transistor T20 being turned on, the voltage at point h is controlled to a fail-safe voltage of 5V, which is lower than the normal voltage. As a result, appropriate retardation characteristics can be obtained even in abnormal situations. Incidentally, when the transistor T20 is omitted, the fail-safe clamp voltage is 6■ as shown in FIG.

FIG. 12 shows an operating waveform diagram when an abnormal voltage is superimposed on point a in FIG. 3. This abnormality detection is performed by the detection circuits 108 and 109.
This is done by operating in combination. Figure 13 (
1) is the pace signal of the power transistor, and if the knock sensor 100 becomes an abnormal signal for some reason, the first
As shown in Figure 2 (2), a voltage higher than the voltage at point b is continuously generated, and therefore the output of the comparator continuously decreases to 0 (v) as shown in Figure 12 (3). Integrating circuit 125
After the operation as shown in FIG. 12 (4), the output of is clamped to a fail-safe clamp voltage (5.4 V in the figure). Therefore, it is possible to sufficiently deal with abnormal voltage9. 13 shows a timing signal generation circuit that generates various timing signals. The monostable circuit 128 includes a resistor R,4
4, R, 45, R46, R47, R48, R50, R5
1, I't52, R53, R54, capacitor C1l,
C12, transistors T3, T4, T5 . =T6, and diodes D4 and D5. The rotation speed detection circuit 156 includes resistors R79, R80, R81, R83, R84, R96,
R103, R201, R202, R2O3, operational amplifier 0P-10, comparator C06, capacitor 018,
It consists of a diode D8 and a transistor T13.

The operation of monostable circuit 128 will be explained. A pace signal of the power transistor shown in FIG. 2(1) is applied to the terminal P. This base signal H turns on the transistor T3 and turns off the transistor T4. Transistor T4
is turned off, the capacitor C12 is connected to the power supply B→resistance R4.
8→R50→C12→A path to the pace of transistor T5 is formed. On the other hand, when the base signal is L, transistor T3 is turned off and transistor T4 is turned on, and the power supply B→
Resistor R51 → Capacitor C12 → Resistor R,50-'D5
→ Transistor T4 → R49 → Ground path is formed. These two paths are charging/discharging circuits to the capacitor C12, and the collector terminal of the transistor T5 is connected to the
) A pulse S2 is generated that is synchronized with the spark timing having a time width t as shown in FIG. Further, since the transistor T6 is in an anti-phase relationship with the transistor T5, it outputs a pulse S9 that is in an anti-phase with the signal 482. This signal S2 is applied to the pace of the transistor T2 of the ignition noise tatting circuit 113 to become an ignition noise cut signal, and is also applied to the mask circuit 1.
It is applied to the pace of transistor T7 of No. 53 and plays the role of cutting ignition noise. Signal 9 is applied to transistor T8 of advance angle signal output circuit 127, and is used for advance angle control. Furthermore, the signal S2 serves as an input signal to the rotation detection circuit 156 and an input signal to the constant time generation circuit 157.

The operation of the rotation speed detection circuit 156 will be explained. Transistor T13! The transistor T3 is turned on due to two conditions being satisfied when the signal S2 is H and the transistor T3 is off. As a result, it turns on with the pulse width t shown in FIG. Since the period of this pulse is proportional to the rotational speed, the transistor T13 is eventually driven according to the rotational speed. The positive terminal of the operational amplifier 0P10 has a pyrovoltage (approximately 1.
7 V) is applied. When the transistor T13 is turned on, a path from the output side of the operational amplifier 0P10 to the capacitor C18→l't80→T13→ground is created, and the capacitor C18 is charged. Transis) Tl
3 is off, the charge on capacitor 018 is transferred to resistor R81.
flows to The operational amplifier 0P10 generates an output corresponding to the deviation of the voltage applied to the positive terminal and the negative terminal, and is applied to the negative terminal of the comparator CO6.

Resistors R84 and R8 are connected to the positive terminal of comparator CO6.
A constant voltage (3, OV) divided into three voltages is applied. The negative terminal of this comparator CO6 has 1.7V.
A voltage corresponding to the rotation speed and above is applied, and a constant voltage of 3
■It is compared. Comparator C06O when 3■ or more
The output becomes LOW, and becomes l (high) when it is 3V or less.The reference voltage of 3V is a voltage corresponding to high-speed rotation.

Specifically, the rotation speed corresponding to this voltage 3■ is 3000
It is set to rpm. Therefore, a o o o rp
Only when it is less than m, the output of comparator C'06 becomes H. The output of comparator CO6 is resistor R201, R20
The voltage is divided by 2 and applied as a signal S7 to the pace of the transistor T20, and the transistor T20 of the clamp circuit 141 is turned on when the rotation is 9 times over 3000 rpm. When the transistor T20 is turned on, the voltage applied to the negative terminal of the operational amplifier 0P16 becomes lower than when the transistor T20 is turned off. However, diode D8 and resistor R103 are designed to have hysteresis characteristics, so it takes time for this circuit to respond to 3000 rpm, and during this time the rotational speed may rise slightly, so the output is adjusted in anticipation of this increase. I'm trying to get it.

The retard control of this embodiment configured in this way will be described next.

As shown in FIG. 14(1), ignition is performed by a signal from a power transistor that provides an ignition signal.

Therefore, an ignition mask signal as shown in FIG. 14(2) is required. This ignition causes the engine to operate, but knocking is detected due to engine vibration.The output from the knock sensor 100 is
The signal is amplified and inputted to the input terminal (a) of the combination IC03 as a signal shown in (a) of FIG. 14 (3), and on the other hand,
A signal as shown in b in FIG. 14(3) outputted from the BGL detection circuit 119 is input to the (c) input terminal of the comparator CO3 and compared. This comparator CO
The output signal of 3 is connected to capacitor C1 via diode D3.
3 is charged and integrated. This integrated voltage (Figure 14 (4)
))U,)Transistor T100 performs on/off operation depending on whether it is higher or lower than the threshold label of VBE of transistor TIO (°Fig. 14(5)). As a result, as shown in FIG. 14 (6), the retard amount of the transistor T1
Therefore, according to this embodiment, since proportional retardation is performed by the on-off operation of the transistor, the actual running characteristics are not affected even in the presence of noise, and the retard speed is quickly increased in the event of knocking. can be retarded.

As explained above, according to the present invention, it is possible to appropriately control knocking.

[Brief explanation of the drawing]

Fig. 1 is an overall configuration diagram of the present invention, Fig. 2 (1) to (8) are time charts, Fig. 3, Fig. 6, Fig. 8, Fig. 11,
Fig. 13 is a concrete circuit diagram, Fig. 4 (G, Fig. 5,
Figures 7, 9, and 10 are illustrations of each characteristic, and Figure 12 (
1) to (4) and 141t(1) to (6) are time charts. 100... Knock sensor, 101... Knock reservoir leg device.

Claims (1)

[Claims] 1. A knock sensor and a first device for inspecting ignition noise during ignition from the sensor output detected by the knock sensor.
means for comparing the knock signal from which the ignition noise has been removed with a predetermined first signal level and outputting a pulse signal; and a second means for integrating the output signal from the second means. and a fourth means for comparing the integral signal outputted from the third means with the second signal level and outputting a rectangular wave.
means, a fifth means for outputting an ignition timing retard amount proportional to the output pulse width from the fourth means, and a sixth means for retarding the ignition timing based on the output from the fifth means. A knock control device consisting of. 2. The knock control device according to claim 1, wherein the rectangular wave of the fourth means is generated by a transistor that is turned on and off by an integral signal output from the third means. . 3. A knock sensor and a first device that removes ignition noise during ignition from the sensor output detected by the knock sensor.
means, a second means for comparing the knock signal from which the ignition noise has been removed with a predetermined first signal level and outputting a pulse signal, and a third means for integrating the output signal from the second means. means, fourth means for comparing the integral signal output from the third means with a second signal level and outputting a rectangular wave, and ignition proportional to the output pulse width from the fourth means. In a knock control device comprising a fifth means for outputting a timing retard amount and a sixth means for retarding the ignition timing based on the output from the fifth means, when the engine cooling water temperature is below a predetermined value, A knock control device characterized by having a means for preventing retard.