JPH11285289A - Non-bearing rotating machine - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a non-bearing rotating machine that is capable of stable floating position control, even in an induction machine using a cage-type rotor that is simple in structure and can be manufactured easily, especially at a low-frequency region where control is difficult. SOLUTION: A rotating machine is provided with a control circuit for adjusting a control magnetic field, so that original magnetic flux distribution instruction value can be obtained by integrating a back electromotive voltage induced in the terminal voltage of a coil winding that is wound around a stator S, detecting the distribution of magnetic flux in the gap of a rotor R and the stator S, and correcting the deformation of the distribution of magnetic flux caused by the induction current in the rotor R. The control circuit is provided with an operation circuit 43 with a transfer function, where the analog integration output of the back electromotive voltage is input and a corrected integral value is output for performing a correction so that the frequency characteristic of the integral output of the back electromotive voltage can be detected down to a low-frequency region. The transfer function is expressed by G(s)=(s+2πfc )/(s+2πfc '), where fc is the lower limit of a frequency for detecting the distribution of magnetic flux before correction, and fc ' is the lower limit of a frequency for detecting the distribution of magnetic flux after correction.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a bearingless rotating machine having both an electric motor function for rotatingly driving a rotating body and a magnetic bearing function for controlling magnetic levitation of the rotating body, and more particularly to a current path for a rotor. The present invention relates to a bearingless rotating machine capable of performing stable levitation control even when an induction rotor having a secondary conductor is used.

[0002]

2. Description of the Related Art Conventionally, a cylindrical rotor is incorporated in a cylindrical stator, and an exciting winding circuit is arranged on the stator to form two types of rotating magnetic fields having different numbers of poles. Various types of bearingless rotating machines have been proposed which apply a force and at the same time apply a position control force for floating and holding at a predetermined radial position.

[0003] In this method, a stator is provided with a winding for rotation driving and a winding for position control, and a three-phase alternating current is supplied to each of the stators. It is formed in a gap between the rotors, and radially distributes magnetic attraction to the cylindrical rotor.

In such a bearingless rotating machine, an m-pole rotating magnetic field and an n-pole rotating magnetic field are generated by passing a current through the winding of the stator. After that, the rotating magnetic field of m poles is used as the driving magnetic field,
The n-pole rotating magnetic field is called a position control magnetic field. The driving magnetic field is used to apply a rotational driving force to the rotor like a normal electric motor. By superimposing the position control magnetic field on the drive magnetic field, it becomes possible to distribute the radial force to the rotor.
For this reason, the floating position of the rotor in the radial direction can be freely adjusted similarly to the magnetic bearing. The m-pole and the n-pole have a relationship of n = m ± 2, so that the above-mentioned flying position control becomes possible.

Accordingly, the rotor is magnetically attracted,
In addition to functioning as an electric motor that applies a rotating force to the rotor, the floating position of the motor can be controlled to function as a magnetic bearing capable of supporting the stator in a non-contact floating manner. This eliminates the need for an electromagnet yoke and windings that constitute a magnetic bearing, which was conventionally required for holding the rotating shaft of the electric motor, shortening the shaft length of the rotating machine and reducing the limitation of high-speed rotation from shaft vibration. can do. Further, the size and weight of the rotating machine can be reduced. In addition, the operation equivalent to a magnetic bearing can be performed by the synergistic action of the magnetic field distribution generated by the current of the position control winding and the current of the drive winding. Control power is generated, and significant energy savings are possible.

[0006] One of the methods of applying an induced current to a secondary conductor of the rotor by a rotating magnetic field generated by the stator to impart a rotational driving force is an induction rotor. The induction rotor has various structures, and a typical one is a cage rotor. In this method, a number of low-resistance metal conductor rods (secondary conductors) are concentrically arranged parallel to the rotation axis as current paths on the rotor, and each end of each metal conductor rod is connected to a low-resistance metal conductor ring (end ring). In this structure, a current path is provided in the rotor. In such a rotor, by cutting off the rotating magnetic flux formed by the stator winding, an induced voltage is generated in the secondary conductor of the rotor, and an induced current flows. Lorentz force is generated by the interaction between the magnetic flux generated by the stator winding and interlinking with the secondary conductor and the induced current flowing through the metal conductor rod of the rotor, and a rotational driving force is generated in the induction rotor. I do.

[0007]

However, in a bearingless rotating machine, a driving magnetic field and a position control magnetic field are mixed and generated by a stator winding current (primary current). When a cage type rotor is used, a current induced by both magnetic fields flows through the rotor current path (secondary conductor). Since a rotating magnetic field of m poles applies a rotational driving force to the rotor, in principle, it does not function as an induction motor unless an induced current flows. On the other hand, when an induced current caused by the n-pole position control magnetic field flows through the rotor current path, a magnetic field generated by the rotor current as a disturbance occurs in addition to the magnetic field generated by the stator winding. Is not determined only by the magnetic field formed by the winding current of the stator, and stable floating control of the rotor cannot be performed.

When an induction rotor is used, the generated control force applied to the rotor depends on the control magnetic flux distribution formed in the air gap between the stator and the rotor, regardless of the stator winding current distribution. Depends on. Therefore, if the magnetic flux distribution in the air gap is directly detected and controlled so that the detected value follows the magnetic flux distribution command value calculated from the rotor displacement, stable levitation control is achieved.

As a method for detecting magnetic flux, there is a method using a semiconductor element such as a Hall element or a magnetoresistive element.
By arranging a plurality of these semiconductor elements in the gap between the rotor and the stator or by embedding them in the magnetic material of the stator, it becomes possible to detect the magnetic flux distribution and use it for control. However, when using such semiconductor components,
(1) If the temperature condition of the magnetic flux measurement site is out of the normal operation range of the detecting element, accurate magnetic flux detection becomes impossible. In an extreme temperature environment, the worst element may be destroyed.
(2) In order to secure a space for installing the detection element, the stator magnetic material must be cut. Due to this processing, the magnetic flux distribution changes from the original state, and accurate measurement becomes impossible. (3) It is necessary to provide electrical wiring for the detecting element in the minute space of the stator, and reliability and mechanical There is a problem that it is not practical from the viewpoint of strength, and it is not realistic.

As a method of detecting the magnetic flux distribution in the air gap between the rotor and the stator, in addition to the method described above, a method of measuring and calculating the terminal voltage of a winding existing in a magnetic flux path in the stator to extract the amount of magnetic flux. There is. This uses the fact that the amount of change in flux linkage (differential amount) and the terminal voltage are proportional to each other, based on the principle of Faraday electromagnetic induction. The number of turns of the winding used for detection is n, the amount of interlinkage magnetic flux is Φ, and the winding area is S
, The winding terminal voltage V _SC is

(Equation 1) Substituting the magnetic flux density B = Φ / S into this, s = d /
When dt is used, a relationship of V _SC = s · nSB is obtained. The transfer function from B to V _SC is represented by G _SC
Notation.

When the detected magnetic flux density signal is represented by V _Out , in order to obtain a relation in which V _Out is proportional to B,

(Equation 2) Is required. The integral operation of the above equation means that a large gain is required at an extremely low frequency, and is not feasible. As a practical solution, the cutoff frequency f
A low-pass filter (LPF) having _C and a DC gain A is used as an incomplete integrator, and an operation similar to the above equation is performed. The transfer function G _int of this incomplete integral is

(Equation 3) _Therefore , the transfer characteristic between the detected magnetic flux density signal V _Out and the actual magnetic flux density B is

(Equation 4) Becomes This shows that in the magnetic flux detection method using the winding terminal voltage, the detection characteristic is a high-pass filter (HPF) having a gain of 2πf _C nSA and a cutoff frequency f _C. This means that it is impossible to detect a magnetic flux that fluctuates in a long cycle.

To extend the lower limit of the detectable frequency range,
It is necessary to reduce the cutoff frequency f _C of the low-pass filter (LPF) described above. (1) The low-pass filter (LPF)
Is produced, the time constant of the circuit element increases with a decrease in f _C , and the circuit operates abnormally. (2) Low-pass filter (LP)
When F) is realized, if the fluctuation of the detected magnetic flux is long, the amplitude of V _SC is very small. When this value is converted into a digital value, a quantization error increases, and accurate calculation cannot be expected.

For this reason, in the conventional method, the low frequency fluctuation of the magnetic flux cannot be detected, and it has been difficult to control in the low frequency region.

The present invention has been made in view of the above-described circumstances, and is stable even in an induction machine using a cage type rotor which is simple in structure and easy to manufacture, especially in a low frequency region where control is difficult. It is an object of the present invention to provide a bearingless rotating machine capable of performing the above-described floating position control.

[0015]

According to the present invention, there is provided a bearingless rotating machine wherein n is m ± 2 in synchronism with the rotating magnetic field of m poles.
The control magnetic field of the pole is superimposed to apply a rotational force to the rotor, and at the same time, the control magnetic field of the n-pole is increased / decreased from the displacement of the rotor detected by the displacement detection means of the rotor, and the rotor is magnetically levitated. In the bearingless rotating machine, by detecting the back electromotive force induced in the terminal voltage of the winding wound on the stator, the magnetic flux distribution in the air gap between the rotor and the stator is detected.
A control circuit that adjusts the control magnetic field so as to obtain an original magnetic flux distribution command value by correcting deformation of the magnetic flux distribution caused by the induced current of the rotor, and the control circuit includes:
In order to correct the frequency characteristic of the analog integrated output of the back electromotive voltage so that it can be detected up to a low frequency region, a transfer function G (s) that inputs the integrated output of the back electromotive voltage and outputs the corrected integrated value as an output. = (S + 2πf _C ) / (s + 2πf _C ′) f _C : Lower limit of frequency at which magnetic flux distribution before correction can be detected f _C ′: Lower limit of frequency at which magnetic flux distribution after correction can be detected And

According to the present invention described above, the actual magnetic flux distribution in the air gap can be detected by integrating the back electromotive voltage generated in the terminal voltage of the winding. Since the detected magnetic flux distribution in the air gap is deformed by the induced current of the rotor, the current distribution of the stator winding is corrected so as to be the original magnetic flux distribution, so that the rotor floating position control can be performed. It is possible to follow an appropriate magnetic flux distribution. Thus, even if a cage type rotor or the like is used, an appropriate magnetic flux distribution for maintaining the floating of the rotor can be formed, so that stable floating position control can be performed together with the rotational driving.

However, since this integration operation means that a large gain is required at an extremely low frequency and cannot be realized, a low-pass filter (LPF) having a cutoff frequency f _C is used as an incomplete integrator. ing. For this reason, there is a problem that the low-frequency fluctuation of the magnetic flux cannot be detected as described above. For this purpose, means for correcting a low-pass filter (LPF) by an analog circuit by a digital calculator is used. That is, the low-pass filter (LPF) having the cutoff frequency f _C is configured by an analog circuit, and its output Vout is input to a digital arithmetic unit. The transfer characteristic of the digital arithmetic unit is represented by Gcomp (s), and the output is represented by Vout ′. It is described below. The purpose of this configuration is to calculate the magnetic flux

(Equation 5) Is to substantially reduce f _C.

The magnetic flux detection characteristic is obtained by correction using a digital arithmetic unit.

(Equation 6) To obtain, Gcomp (s) is

(Equation 7) Should be fine. This is calculated by a digital calculator. As a result, f _C can be reduced by f _C ′, and the control region in the low frequency range can be expanded.

When Gcomp (s) is realized by an analog circuit, the time constant of electronic components in the circuit increases, and the problem cannot be solved.

The transfer function G (s) is subjected to arithmetic processing using a digital arithmetic unit. As a result, the calculation of G (s) = (s + 2πf _C ) / (s + 2πf _C ′) becomes possible, and the control range in the low frequency region can be expanded.

Further, it is preferable to use a search coil as a winding wound around the stator for detecting the magnetic flux density in the air gap between the stator and the rotor. Also, as the winding wound around the stator that detects the magnetic flux density in the gap between the stator and the rotor, the winding itself for generating a magnetic field in the gap between the rotor and the stator should be used. You may.

[0022]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a diagram showing a configuration of a control system of a conventionally used general bearingless rotating machine, which is a premise for describing an embodiment of the present invention. The rotor R is driven to rotate by a two-pole rotating magnetic field formed by a two-pole driving winding provided on the stator S, and a floating position is set by superimposing a four-pole rotating magnetic field formed by a four-pole position control winding. Controlled. Around the rotor R, a rotational speed detector 10 for detecting the rotational speed of the rotor R, and a gap sensor 11 for detecting the floating position of the rotor R in the x direction and the y direction.
x and 11y are respectively arranged.

In the speed control system (bipolar rotating magnetic field), a speed command value ω ^* is given in advance, and this is compared with the actual rotation speed ωm detected by the rotation speed detector 10 by the comparator 20. Then, this deviation is input to the PI (D) controller 21, and the torque component current I is adjusted so that the deviation becomes zero.
t ^* is output. On the other hand, an exciting component current Io ^* corresponding to the exciting current is given in advance. Then, the rotation coordinate-fixed coordinate conversion calculator 22 inputs the current I
From t ^* and Io ^* , two-phase currents Ia ^* and Ib ^{* in} a fixed coordinate system are obtained by a matrix operation shown in FIG.

Then, the two-phase current Ia ^* ,
The Ib ^* is converted into three-phase currents Iu ₂ ^* , Iv ₂ ^* ,
Iw ₂ ^* , and power-amplified by the power amplifier 24 to a predetermined current value, and supplied to the two-pole winding of the stator S. As a result, a two-pole rotating magnetic field that drives the rotor R to rotate at the speed command value ω ^* is formed.

On the other hand, the control of the position control system (quadrupole rotating magnetic field) is roughly as follows. First, the gap sensor 1
The floating position of the rotor R is detected by 1x, 11y, and is compared with a preset floating position command value x ^* , y ^* by the comparator 25. Then, the respective deviations Δx, Δy are input to the PI (D) controller 26, and position control force command values Fx ^* , Fy ^* for making the deviation zero are calculated.
Then, in the controller 27, the position control force command values Fx ^* , F
From y ^* , two-phase control current command values Iα ^* and Iβ ^*, which are converted from a rotating coordinate system to a fixed coordinate system by a matrix operation shown in the figure for rotation angle ωt, are output. And a two-phase to three-phase conversion circuit 2
At 8, the current is converted into a three-phase current command value Iu ₄ ^* , Iv ₄ ^* , Iw ₄ ^* , and a predetermined current is supplied to the four-pole stator winding by the power amplifier. A quadrupole floating position control magnetic field is formed in the gap between the stator and the rotor, and is superimposed on the bipolar driving magnetic field, thereby controlling the floating position of the rotor R.

However, the two-phase current command values Iα ^* and Iβ ^* obtained by the operation of the controller 27 are determined without considering the induced current (secondary current) flowing in the current path of the rotor.
Therefore, when an induced current flows through the rotor by a cage rotor or the like, the magnetic flux distribution based on the two-phase current command values Iα ^* and Iβ ^* based on the levitation position detected by the magnetic flux sensor and the actual stator-rotor A difference occurs in the magnetic flux distribution in the air gap.
As described above, the floating position control magnetic field distribution is deformed by the induced current generated in the rotor current path, and the normal floating position control force cannot be applied.

FIG. 2 is a diagram showing a section perpendicular to the rotation axis of a stator to which a search coil is added. As described above, there are various problems when a semiconductor sensor is used for detecting magnetic flux. For this reason, in this embodiment, a search coil is arranged as shown in FIG.
The magnetic flux distribution is determined. 24 on the stator side
In this figure, a four-pole winding indicated by a large circle in the figure is provided outside the slot (S _{L1 to} S _L24 ).
Two-pole windings, indicated by small circles, are arranged. This two-pole winding is a winding for forming a two-pole driving magnetic field,
The four-pole winding is a winding for forming a four-pole position control magnetic field. Twelve search coils (S _{c1 to} S _c12 ) are wound around the teeth between the slots of the stator at equal intervals in the circumferential direction.

FIG. 3 is a block diagram of a control system of a bearingless rotating machine of a system that integrates the output of a search coil. The speed control system for controlling the two-pole drive winding current is omitted because it is completely the same as in FIG. When focusing only on the position control system, FIG.
Is intended to calculate four-pole current command values Iu ₄ ^* , Iv ₄ ^* , Iw ₄ ^* and to apply current to the windings according to the command values. On the other hand, in the configuration of FIG. 3 of an embodiment of the present invention, 4-pole magnetic flux distribution command value Biarufa ^*, calculates the Bbeta ^*, is the circuit configured to form a magnetic flux distribution on the command value as expected.

That is, as shown in FIG. 2, search coils (S _{c1 to} S _c12 ) are provided on the stator teeth in order to measure the magnetic flux distribution in the air gap between the rotor and the stator of the motor.
Further, an integrator 31 by an analog circuit for converting the terminal voltage into a magnetic flux density, a dipole magnetic flux distribution calculator 32 for obtaining a dipole magnetic flux distribution vector and a quadrupole magnetic flux distribution vector from the calculated magnetic flux density, a quadrupole magnetic flux It has a distribution calculator 33 and the like. Furthermore, the command values Bα ^* , Bβ of the position control magnetic flux distribution
^{* From} the detected values Ba and Bb of the two-pole magnetic flux distribution calculator 32 and the command values Fx ^* and Fy ^{* of the} generated control force.
A position control magnetic flux distribution command value calculator 34 for obtaining α ^* and Bβ ^* is provided. Further, there is provided a calculator 35 for adding the calculated quadrupole magnetic flux distribution detection values Bα, Bβ and the low frequency component of the quadrupole winding current.

By integrating each terminal voltage of the search coils (S _{c1 to} S _c12 ) by the integrator 31,
The magnetic flux density of the stator teeth around which the search coils (S _{c1 to} S _c12 ) are wound can be obtained. For convenience, ₁ flux density obtained in each part of the search coil _{B, B 2, ···, B} 12
Expressed by

The detected value of the bipolar magnetic flux density distribution vector (Ba,
Bb) is calculated by the bipolar magnetic flux distribution calculator 32 using the detected magnetic flux densities B _{1 to} B ₁₂ and using the equation (8).

(Equation 8)

The quadrupole magnetic flux density distribution vector detection values (Bα, Bβ) are the magnetic flux densities B _{1 to} B ₁ similarly detected.
₁₂ and is calculated by the quadrupole magnetic flux distribution calculator 33 using equation (9).

(Equation 9)

The obtained two-pole magnetic flux density distribution vector detection values (Ba, Bb) are, as shown in FIG.
x ^*, the command value of the quadrupole magnetic flux distribution in the calculator 34 with ^{Fy * (Bα *, Bβ *} ) used in the calculation of the. 4-pole magnetic flux density distribution vector detection value (Bα, Bβ) is computed command value (Bα ^*, Bβ ^*) is subtracted by the comparator 35 from obtaining a deviation (ΔBα, ΔBβ). The obtained two-pole magnetic flux distribution vector detection values (Ba, Bb) are used as control force command values Fx ^* , Fy ^*.
At the same time, a command value (Bα ^* ,
Bβ ^* ). The quadrupole magnetic flux distribution vector detection values (Bα, Bβ) are calculated command values (Bα ^* , Bα
β ^* ) is subtracted by the comparator 35, and the deviation (ΔBα, Δ
Bβ).

The deviation signal (ΔBα, ΔBβ) between the command value and the detection value of the quadrupole magnetic flux distribution obtained in this way is converted by the two-phase to three-phase converter 39 so as to be adapted to the three-phase winding of the stator. Then, the command values ΔBu ₄ ^* , ΔBv ₄ ^* , ΔBw _{4 of} the magnetic flux density distribution
^{Get *} . The sign of this signal is discriminated by the hysteresis comparator 40 and used as an on / off control signal for each power element of the three-phase inverter. That is, ΔBα and ΔBβ have signs +
In this case, the sign of the supply current of the inverter is negative, that is, the current is reduced so that the deviation is adjusted to zero. Thus, the magnetic flux density distribution in the air gap between the stator and the rotor follows the command without delay, and as a result, the expected position control magnetic flux distribution can be generated.

Since the magnetic flux detection method using the search coil uses an incomplete integration circuit, it cannot detect the DC component of the detected magnetic flux. For this reason, in the present embodiment, the low frequency components of the four-pole winding currents Iu ₄ , Iv ₄ , and Iw ₄ are separated and fed back as an alternative. That is, the four-pole winding current I
u ₄ , Iv ₄ , and Iw ₄ are detected by CT, phase-converted by a three-phase to two-phase converter, processed by a low-pass filter (LPF), and fed back. As a result, a low-frequency component having low magnetic flux detection sensitivity can be supplemented, and a required magnetic flux distribution can be formed in the gap between the rotor and the stator.

However, in the above-described method, low-frequency magnetic flux fluctuations cannot be captured, so that there is an uncontrollable region. For example, when the rotation driving magnetic field rotates at a low speed, the magnetic flux distribution in each part of the search coil is slow, so that the magnetic flux distribution in this state cannot be detected. This means that a satisfactory floating state cannot be achieved at a low rotational speed of the rotor. As is clear from the above description, it is essential to increase the lower limit of the region where the magnetic flux distribution can be detected.

FIG. 4 is a diagram showing a control configuration of a bearingless rotating machine according to an embodiment of the present invention which solves this point.
The speed control system for controlling the two-pole drive winding current is omitted because it is exactly the same as that in FIG. When focusing only on the position control system, the difference between the control system shown in FIG. 3 and the control system shown in FIG. 4 is as follows: (1) The search coil output voltage is input to an integrator constructed by an analog circuit; And (2) the calculation result of the detected magnetic flux distribution is taken into the digital calculator 43, and the calculation for expanding the detection area is performed.

When the cutoff frequency f _C of the analog integrator 31 used in the configuration of the low-pass filter (LPF) is used, the calculation in the digital calculator 43 is expressed by using the s parameter, G (s) = ( A transfer function of s + 2πf _C ) / (s + 2πf _C ′) is realized. Here, f _C is the lower limit of the detection frequency range without the digital calculator, and f _C ′ is the lower limit of the detection frequency range when the correction by the digital calculator is performed.

Conventionally, the detection frequency range of the magnetic flux distribution has been restricted by the cutoff frequency f _C. On the other hand, in the present invention, the analog integrator 31 and the digital arithmetic unit 43
Can lower the detection frequency lower limit of the magnetic flux distribution to f _C ′. The digital arithmetic unit (integrator) 43 is arranged at the subsequent stage of the analog integrator 31 for the following reason.

(1) The terminal voltage of the search coils (S _{c1 to} S _c12 ) mainly includes the switching frequency component of the inverter (power amplifier), and this frequency component is much higher than the sampling frequency of the digital calculator 43. . For this reason, the search coil terminal voltage is directly
However, accurate calculation of magnetic flux is impossible. Since the analog integrator has a low-pass filter (LPF) configuration, the analog integrator has an effect of removing voltage fluctuation of a switching frequency component. As a result, the problem of the inverter noise is eliminated by using the output of the analog integrator 31 as the input of the digital calculator 43. (2) The resolution of the signal handled by the digital arithmetic unit 43 is A /
It depends on the number of bits in the D converter and the D / A converter.
For this reason, when the voltage level of the input signal or the output signal is extremely small, the quantization error becomes large and accurate calculation becomes impossible. If the integrator is constituted only by the digital computing unit, the frequency region that can be detected becomes extremely narrow due to this problem. The terminal voltage of the search coil is first processed by the analog integrator 31 so that the digital arithmetic unit 43
At which the quantization error hardly occurs. (3) Generally, analog circuits are less expensive than digital arithmetic units. During operation of the non-bearing rotary machine, cause some trouble, when the output voltage terminal voltage of the search coil (S _c1 ~S _c12) is impossible Usually, need only inexpensive analog circuit from being broken. As a result, emergency damage can be minimized.

The purpose of converting the magnetic flux density signals (B _{1 to} B ₁₂ ) into magnetic flux distribution signals Ba, Bb, Bα, and Bβ and then processing the signals with the digital calculator 43 is to reduce the calculation processing. That is, in the configuration of the control circuit shown in FIG. 4, four digital arithmetic units 43 may be prepared, but the magnetic flux density signal (B ₁
In order to correct .about.B ₁₂ ) with a digital calculator, 12 digital calculators must be prepared.

As is clear from the above description, the present invention has realized the expansion of the detection frequency range in the magnetic flux distribution detection method using the search coil. This makes it possible to detect and control a magnetic flux distribution that fluctuates in a long cycle, which was previously impossible. As a result, it has become possible to remarkably widen the operation operation area in which the bearing-free rotating machine can stably float.

In the above description, a magnetic flux detection method using a search coil has been used for convenience. However, the present invention can be similarly applied to a magnetic flux detection method using a driving winding wound around a stator and a position control winding. Further, in the configuration of the winding shown in FIG. 2 described above, the winding divided into the driving winding forming the driving magnetic field distribution and the position control winding forming the position controlling magnetic field distribution is used.
Any winding may be used as long as it can form a desired magnetic field distribution.

Although the winding wound on the stator is premised on the winding of the three-phase midpoint connection, if the above-described magnetic flux distribution can be generated, the winding distribution does not matter. Any number of poles can be applied as long as the rotation drive magnetic field of m poles and the position control magnetic field of n poles have a relationship of m = n ± 2.

[0045]

According to the present invention as described above,
Both the magnetic levitation and the rotational drive, which are the objects of the bearingless rotating machine, can be achieved by using a rotor that deforms the original position control magnetic flux distribution of an ordinary and widely used induction motor. This eliminates the necessity of using a rotor having a complicated current path structure, and makes it possible to use an inexpensive and robust, for example, a commonly used cage type rotor as a rotor of a bearingless rotating machine, and to reduce a low frequency. The operating range can be extended to the area.

[Brief description of the drawings]

FIG. 1 is a block diagram showing a general configuration of a control system of a bearingless rotating machine.

FIG. 2 is an explanatory diagram showing a stator winding structure and an arrangement of search coils of a bearingless rotating machine.

FIG. 3 is a block diagram of a bearingless rotating machine that integrates a search coil output.

FIG. 4 is a block diagram illustrating a configuration of a control system of the bearingless rotating machine according to the embodiment of the present invention.

[Explanation of symbols]

 Reference Signs List 31 integrator 32 two-pole magnetic flux distribution calculator 33 four-pole magnetic flux distribution calculator 34 position control magnetic flux distribution command value calculator 35 comparator 39 two-phase three-phase converter 40 sign determiner 42 inverter (power amplifier) 43 digital calculator R rotor S stator

Claims (4)

[Claims] 1. The method according to claim 1, further comprising the steps of:
2, the n-pole control magnetic field is superimposed to apply a rotational force to the rotor, and at the same time, the n-pole control magnetic field is increased / decreased from the rotor displacement detected by the rotor displacement detection means, and the rotation is increased. In a bearingless rotating machine that magnetically levitates the rotor, the magnetic flux distribution in the air gap between the rotor and the stator is detected by integrating the back electromotive force induced in the terminal voltage of the winding wound on the stator. And a control circuit that adjusts the control magnetic field so as to obtain an original magnetic flux distribution command value by correcting deformation of the magnetic flux distribution caused by the induced current of the rotor. In order to correct the frequency characteristic of the integrated output of the back electromotive voltage so that it can be detected up to the low frequency region,
Transfer function G (s) = (s + 2πf _C ) / (s + 2πf _C ′) f _{C where} the analog integrated output of the back electromotive voltage is input and the corrected integrated value is output. Lower limit f _C ′: A bearingless rotating machine comprising an arithmetic circuit having a lower limit of the frequency at which the corrected magnetic flux distribution can be detected. 2. The bearingless rotary machine according to claim 1, wherein the transfer function G (s) is subjected to arithmetic processing using a digital arithmetic unit. 3. The bearingless rotating machine according to claim 1, wherein a search coil is used as the winding wound around the stator. 4. The method according to claim 1, wherein a winding for generating a magnetic field in a gap between the rotor and the stator is used as the winding wound around the stator. Bearingless rotating machine.