JPH10271064A - Optical signal demodulator for remote controller - Google Patents

Abstract

PROBLEM TO BE SOLVED: To prevent malfunction due to an external disturbance light such as a fluorescent light in a receiver for an infrared ray remote commander. SOLUTION: An-optical code signal subject to photoelectric conversion by a photo diode PD is given to a preamplifier A1, an amplifier A2 and a BPF, where a carrier frequency component is extracted, and its level is discriminated by a detection circuit 12 with a threshold voltage Vth set by a pulse width discrimination circuit 13 and the discrimination result is outputted from an output terminal 14 as a code signal pulse. A peak level being a voltage division level is extracted via voltage division resistors R11, R12, and a peak hold circuit 21 in the pulse width discrimination circuit 13 and the peak level is given to a detection circuit 22, where an output of the BPF is detected and a pulse component of the code signal is extracted. The pulse width is converted into a voltage by an integration circuit 23 and when the voltage is discriminated to be at the outside of a prescribed range at comparators A11, A12, a threshold level change circuit 25 increases the threshold voltage Vth.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an optical signal demodulation device of a remote control device suitably implemented as a so-called infrared remote control receiving device.

[0002]

2. Description of the Related Art FIG. 10 is a block diagram showing the electrical configuration of a typical prior art infrared remote control receiver 1. The receiving device 1 generally includes a photodiode p
d, a preamplifier a1, an amplifier a2, a bandpass filter bpf, a detection circuit 2, and an integration circuit 3.

[0003] From a transmitting device not shown, 30 to 70 k
A carrier frequency is set to Hz, and an infrared optical code signal whose carrier is modulated corresponding to the presence or absence of a pulse of a code signal representing control information is transmitted.

[0004] The optical code signal is a photodiode p
The light is received at d and converted into a current corresponding to the optical code signal. The output current of the photodiode pd is current-voltage converted by the feedback resistor r1 of the preamplifier a1, and is supplied to the amplifier a2 via the AC coupling capacitor c1. In the amplifier a2, the input signal is the resistance r on the input side.
After being amplified by r3 / r2 times by 2 and the resistor r3 on the feedback side, it is output to the band-pass filter bpf. The band-pass filter bpf extracts the 30-70 kHz component, which is the carrier frequency component, and outputs it to the detection circuit 2 and the integration circuit 3 in common.

[0005] The detection circuit 2 includes the band pass filter b
The output voltage v1 from the pf is integrated with an integrating circuit 3 as described later.
, The carrier signal component is removed by level discrimination with a predetermined threshold voltage vth, and the code signal is demodulated by level discrimination with a predetermined discrimination level vref2 and output to the output terminal 4.

The integrator 3 is a circuit for generating the threshold voltage vth based on the output voltage v1, and includes comparators a3, a4, constant current sources 5, 6, and a capacitor c.
2, c3 and a buffer b. The comparator a3 is a current output type differential amplifier. The output voltage v1 is supplied to an inverting input terminal of the comparator a3, and the threshold voltage vth is supplied to a non-inverting input terminal of the comparator a3. The current i1 is sucked from the integrating capacitor c2,
The suction is stopped when v1 ≦ vth. Also, a constant current i2 is constantly supplied to the capacitor c2 from a power source having a predetermined constant voltage Vcc via a constant current source 5. The capacitor c2 is charged and discharged with the currents i2 and i1, and its terminal voltage v2 is given to the inverting input terminal of the comparator a4. The comparator a4 is a current output type differential amplifier. When the terminal voltage v2 is lower than a predetermined reference voltage vref1, the comparator a4 outputs a predetermined constant current i3 to a smoothing capacitor c3. When the voltage is equal to or higher than the voltage vref1, the output is stopped. A constant current source 6 is also provided in parallel with the capacitor c3, and the constant current source 6 constantly discharges a predetermined constant current i4 from the capacitor c3.
A voltage exactly equal to the terminal voltage v3 of the capacitor c3 charged and discharged by the currents i3 and i4 is output by the buffer b as the threshold voltage vth.

The detection circuit 2 comprises comparators a5 and a6,
It comprises a constant current source 7 and an integrating capacitor c4. The comparator a5 is a current output type differential amplifier similar to the comparator a3. The inverting input terminal receives the output voltage v1. The non-inverting input terminal receives the threshold voltage vth. When v1> vth, a predetermined constant current i5 is drawn from the capacitor c4, and v1 ≦ vt
When it is h, the suction of the current is stopped. A predetermined constant current i6 is constantly supplied to the capacitor c4 from the power source of the constant voltage Vcc via the constant current source 7. The terminal voltage v4 of the capacitor c4 charged and discharged by the currents i6 and i5 is input to the inverting input terminal of the comparator a6. A predetermined reference voltage vref2 is input to a non-inverting input terminal of the comparator a6.
Converts the output voltage v5 to the output terminal 4 into v4 <vre
When f2, the high level is set, and v4 ≧ vref2
, It is set to low level.

FIG. 11 is a waveform chart for explaining the operation of the receiving apparatus 1 configured as described above. The output voltage v1 from the band-pass filter bpf is shown in FIG.
As shown in (a), the component of the carrier frequency is output only during the period in which the pulse of the code signal exists, and the output voltage v
1 is discriminated by the comparators a3 and a5 based on the threshold voltage vth, respectively.
3 and a5 sink the currents i1 and i5, respectively. Therefore, during this period, the terminal voltages v2 and v4 of the capacitors c2 and c4 become the voltages shown in FIGS.
It goes low as shown in FIG. On the contrary,
In the period where v1 ≦ vth, the terminal voltages v2 and v4 of the capacitors c2 and c4 are equal to those shown in FIGS.
As shown by (c), the current rises due to the charging of the currents i2 and i6. However, i1≫i2,
By setting i5≫i6, these terminal voltages v
2 and v4 are less than the reference voltages vref1 and vref2, respectively, during the pulse period, and the carrier frequency component is removed. The smoothed value of the discrimination result of the terminal voltage v2 at the reference voltage vref1 becomes the threshold voltage vth as described above, and the reference voltage vth of the terminal voltage v4 is obtained.
The result of discrimination by ref2 is an output voltage v5 corresponding to the code signal as shown in FIG.

[0009]

In the receiving apparatus 1 configured as described above, whether to output a demodulated code signal is determined by whether or not v1> vth. The threshold voltage vth is determined by the charging current i3 to the capacitor c3 from the comparator a4 and the discharging current i4 by the constant current source 6.

On the other hand, an optical code signal used in an infrared remote controller of a home electric appliance or the like is a pulse train having a carrier wave of 30 to 70 kHz, and a pulse width w1 thereof is, for example, 200 to 800 μsec. The duty ratio, which is the ratio of the above, is selected to be about 10 to 40%. Therefore, if the ratio of the currents i3 and i4 becomes large, for example, about 10: 1, 30%
When an optical code signal having a duty ratio of? Is input, the threshold voltage vth output from the integration circuit 3 to the carrier detection comparator a5 becomes substantially equal to the output voltage v1, and the carrier can be detected. Will be gone. For this reason, the ratio of the currents i3 and i4 is selected to be about 2: 1 so that the optical code signal whose duty ratio is about 40% of the upper limit can be received.

However, the fluorescent lamp, which is a main noise source of the infrared remote controller, has a duty ratio of 40%.
%, The threshold voltage vth from the integrating circuit 3 does not reach the peak of the noise with respect to the noise from a fluorescent lamp having a low duty ratio, and the noise is discriminated. However, there is a problem of malfunction.

An object of the present invention is to provide an optical signal demodulation device of a remote control device which can prevent a malfunction with respect to a noise source having a low duty ratio.

[0013]

According to a first aspect of the present invention, there is provided an optical signal demodulation device for a remote control device, wherein a pulse having a specific frequency is used as a carrier wave, and the pulse is modulated in accordance with control information. An apparatus for receiving an optical code signal by a light receiving element, demodulating a photoelectric conversion output of the optical code signal obtained from the light receiving element by demodulation means, and demodulating a code signal corresponding to the control information, Determining whether the pulse width of the signal is within a predetermined range; permitting the output of the code signal within the predetermined range; and inhibiting the output of the code signal outside the predetermined range. It is characterized by comprising means.

According to the above arrangement, for example, for a so-called infrared remote controller, the pulse width of the demodulated code signal is 200 to 200 which is the pulse width of the infrared remote controller.
Only when the time is 800 μsec, the output of the code signal is permitted, input to the decoding circuit at the subsequent stage, demodulated into a control output corresponding to each control target, and output to each control target.

Therefore, even if disturbance light such as an inverter-type fluorescent lamp driven and lit at a frequency close to the carrier frequency of 30 to 70 kHz and a duty ratio larger than the duty ratio of the code signal is input to the light receiving element, The pulse width obtained by level-discriminating the disturbance light falls outside the predetermined range, the output of the code signal with respect to the input is prohibited, the optical noise separation characteristics can be improved, and malfunctions can be prevented.

Further, in the optical signal demodulating device of the remote control device according to the present invention, the pulse width judging means is configured to amplify the signal by the amplifier of the demodulating means and further extract the light by the band-pass filter. A peak hold circuit that receives a carrier frequency component of the photoelectric conversion output, integrates or peak-holds the divided voltage output of the band-pass filter, and creates a reference discrimination level corresponding to a received light level; and an output of the band-pass filter. A detection circuit that detects the code signal by removing a carrier component, and an integrator that integrates the code signal from the detection circuit and generates a voltage corresponding to the pulse width. , The output of the integrator to the first of the shorter pulse widths.
2 corresponding to the second threshold and the longer second threshold, respectively.
First and second comparators for comparing two voltages, and in response to outputs of the first and second comparators, wherein the pulse width is greater than or equal to the first threshold and less than or equal to the second threshold. A threshold changing means for lowering a threshold for demodulating the output of the band-pass filter in the demodulation means into a pulse signal when the output is within the range.

According to the above configuration, when discriminating the pulse width, a voltage corresponding to the pulse width of the code signal is generated by the integrator, and the voltage is subjected to the wind comparison by the first and second comparators. A determination is made as to whether the pulse width is equal to or greater than the first threshold and equal to or less than the second threshold. The threshold value for demodulation is reduced only when it is within the predetermined range to enable demodulation of the code signal, and when it is out of the predetermined range, it is increased to improve noise immunity. Thus, the permission / inhibition control of the output of the code signal by determining the pulse width as described in claim 1 can be specifically realized.

Further, in the optical signal demodulation device of the remote control device according to the third aspect of the present invention, the threshold value changing means detects a leading edge of the code signal by differentiating the code signal from the detection circuit. A third comparator, and reset to a low level by an output from the third comparator;
A first set-reset flip-flop which is set to a high level by an output from the comparator of the second comparator, and which is set to a high level by an output from the third comparator and which is set to a low level by an output from the second comparator. A second set-reset flip-flop that is reset to a level, and a charge of a hold capacitor that holds the threshold when outputs from the first and second set-reset flip-flops are both at a high level. And discharging means for lowering the threshold value by discharging the voltage.

According to the above configuration, in the configuration of the second aspect, a voltage corresponding to the pulse width is generated by the integrator, and the voltage is window-compared by the first and second comparators. When ascending and descending,
By using the first and second set-reset flip-flops as described above, only when the voltage rises, the two threshold voltages will be in the range of two threshold voltages. It is determined that the voltage is within the range of the voltage. Thereby, the determination accuracy can be further improved.

Further, in the optical signal demodulator of the remote control device according to the present invention, the third comparator includes a differential amplifier having a predetermined offset voltage therein, and a pair of the differential amplifier. It is characterized by having a pair of bias resistors for biasing the input terminals to the same voltage, and an AC coupling capacitor for differentiating and inputting the code signal from the detection circuit to one input terminal.

According to the above configuration, in the steady state, the comparison result of either the high level or the low level is output, and when the pulse corresponding to the start of the pulse of the code signal is input from the AC coupling capacitor. The output can be quickly switched to either the high level or the low level. Thus, a highly sensitive comparator can be realized.

Furthermore, the optical signal demodulation device of the remote control device according to the present invention is characterized in that the ratio of the charge current to the discharge current of the hold capacitor is 10: 1 or more.

According to the above configuration, the terminal voltage of the hold capacitor is substantially equal to the peak output of the band-pass filter, so that malfunction due to disturbance light having a low duty ratio can be almost completely prevented.

Further, in the optical signal demodulation device for a remote control device according to the present invention, the predetermined range is 200 to 800 μsec which is a pulse width of a code signal used in the infrared remote control device. And

According to the above configuration, the pulse train of the code signal in the infrared remote controller of the home electric appliance or the like can be demodulated separately from the noise signal of the fluorescent lamp or the like.

[0026]

DESCRIPTION OF THE PREFERRED EMBODIMENTS One embodiment of the present invention will be described.
The following is a description based on FIGS. 1 to 9.

FIG. 1 is a block diagram showing an electric configuration of a receiving device 11 of an infrared remote controller according to one embodiment of the present invention. The receiving apparatus 11 is generally provided with a photodiode PD, a preamplifier A1, an amplifier A2, a bandpass filter BPF, a detection circuit 12, and a pulse width determination circuit 13.

From a transmitting device not shown, 30 to 70 k
A carrier frequency is set to Hz, and an infrared optical code signal whose carrier is modulated corresponding to the presence or absence of a pulse of a code signal representing control information is transmitted.

The optical code signal is generated by a photodiode P
D is received and converted into a current corresponding to the optical code signal. The output current of the photodiode PD is current-voltage converted by the feedback resistor R1 of the preamplifier A1, and is supplied to the amplifier A2 via the AC coupling capacitor C1. In the amplifier A2, the input signal is the resistance R on the input side.
After being amplified by R3 / R2 times by the resistor 2 and the feedback side resistor R3, it is output to the band-pass filter BPF. The band-pass filter BPF extracts the 30-70 kHz component, which is the carrier frequency component, and outputs it to the detection circuit 12 and the pulse width determination circuit 13 in common.

The detection circuit 12 discriminates the level of the output voltage V1 from the band-pass filter BPF with a predetermined threshold voltage Vth obtained from the pulse width determination circuit 13 as described later to remove a carrier component, and further removes the carrier component. And demodulates the code signal by level discrimination at the discrimination level Vref1 and outputs it to the output terminal 14. Therefore, the detection circuit 12 includes the comparators A3 and A4, the constant current source 15, the integrating capacitor C2, and the offset voltage source 16.

The comparator A3 is a current output type differential amplifier. The output voltage V1 from the band-pass filter BPF is input to an inverting input terminal, and the non-inverting input terminal is connected to the pulse width judgment circuit 13. The offset voltage ΔV generated by the offset voltage source 16 is added to the threshold voltage V10 created in
The threshold voltage Vth created by adding th is input. When V1> Vth, the comparator A3 sinks a predetermined constant current I1 from the capacitor C2, and
When 1 ≦ Vth, there is no response,
Stop the suction. The capacitor C2 also has
A predetermined constant current I2 is constantly supplied from a power supply having a predetermined constant voltage Vcc via a constant current source 15.

The terminal voltage V2 of the capacitor C2 corresponding to the charging and discharging of the currents I2 and I1 is input to the inverting input terminal of the comparator A4. A predetermined reference voltage Vref1 is input to a non-inverting input terminal of the comparator A4, and the comparator A4 outputs the output terminal 14 when V2 <Vref1.
Output signal V3 to a high level, and V2 ≧ Vref1
Becomes low level. The code signal thus demodulated is supplied to a decoding circuit (not shown) or the like, and a control output corresponding to each control target is demodulated.

The pulse width determination circuit 13 includes a detection circuit 1
It is determined whether the pulse width of the code signal to be demodulated in step 2 is within a predetermined range. If the pulse width is within the range, the threshold voltage V10 is lowered to allow demodulation of the code signal, and the code signal is out of the range. Sometimes it is a circuit for raising the threshold voltage V10 to inhibit demodulation of the code signal. The pulse width determination circuit 13 generally includes a voltage dividing resistor R11,
R12, peak hold circuit 21, and detection circuit 22
, An integrating circuit 23, a differentiating circuit 24, a comparator A11,
A12 and a threshold changing circuit 25.

The output voltage V1 from the bandpass filter BPF is divided into R12 / (R11 + R12) by the voltage dividing resistors R11 and R12 and input to the peak hold circuit 21, and the voltage V11 obtained by the voltage dividing is obtained. Is the comparator A
Thirteen non-inverting input terminals. The output from the comparator A13 is input to a holding capacitor C11 via a diode D11, and the terminal voltage of the capacitor C11 is input to the inverting input terminal of the comparator A13. Therefore, the comparator A13 includes the capacitor C11
When the terminal voltage is lower than the voltage V11, a high-level output is derived to charge the capacitor C11, and thus a peak value of the voltage V11 is created. The peak value is faithfully output as the hold voltage V12 via the buffer B11.

The hold voltage V12 is used as a detection circuit 2 for removing the carrier frequency component when determining the pulse width.
2 is input. The detection circuit 22 includes the detection circuit 1
The comparator A14, the constant current source 26, which are configured similarly to the comparator A3, the constant current source 15, and the capacitor C2 of the
And a capacitor C12. The comparator A14 is a current output type differential amplifier. The output voltage V1 is input to an inverting input terminal, the hold voltage V12 is input to a non-inverting input terminal, and V1> V12.
, The current I11 is drawn from the capacitor C12, and when V1 ≦ V12, the suction is stopped. A constant current I12 is constantly supplied to the capacitor C12 from a power source having a predetermined constant voltage Vcc via a constant current source 26. The terminal voltage V1 of the capacitor C12 generated by charging and discharging these currents I12 and I11.
3 is I11≫I12, and thus substantially corresponds to the pulse of the code signal from which the pulsation of the carrier frequency component has been removed. The terminal voltage V13 is input to the integrating circuit 23 and also to the differentiating circuit 24 via the buffer B12 that transmits the voltage faithfully.

When measuring the pulse width of the pulse of the code signal detected by the detection circuit 22, the integration circuit 23 is a circuit for converting the pulse width into a voltage.
It comprises a comparator A15, a capacitor C13 for integration, and a constant current source 27.

The comparator A15 is a current output type differential amplifier. The terminal voltage V13 is inputted to an inverting input terminal thereof, and a predetermined reference voltage Vre is inputted to a non-inverting input terminal thereof.
When f13 is input and V13 <Vref11, a predetermined constant current I21 flows out to the capacitor C13, and when V13 ≧ Vref11, the output of the constant current I21 is stopped. The capacitor C13
, The constant current I22 is constantly discharged by the constant current source 27.

The time when V13 <Vref11 is almost equal to the pulse width ts of the code signal. Therefore, the output voltage V14 from the integrating circuit 23 is as follows: V14 = {(I21-I22) × ts} / C13 The voltage is proportional to the pulse width ts. By setting I21: I22 = 2: 1, if the duty ratio of the code signal is 50% or less, there is no problem in receiving the code signal. Thus, the conversion of the pulse width ts into the output voltage V14 can be performed.

The output voltage V14 is supplied to comparators A11 and A1.
The reference voltages Vref21 and Vref22 are input to the non-inverting input terminals of the comparators A11 and A12, respectively. When V14 ≧ Vref21, the comparator A11 sets its output voltage V21 to low level, and V14 <Vr
When it is ef21, the output voltage V21 is set to a high level. When V14> Vref22, the comparator A12 sets its output voltage V22 to low level,
When 4 ≦ Vref22, the output voltage V22 is set to a high level. Note that Vref21 <Vref22
Has been chosen. The reference voltage Vref21 is a value corresponding to the shorter threshold value t1 of the pulse width ts, and is set such that Vref21 = {(I21−I22) × t1} / C13. Furthermore, the reference voltage Vref22 is equal to the longer threshold value t of the pulse width ts.
2, and is set so that Vref22 = {(I21−I22) × t2} / C13 holds.

Therefore, when V14 <Vref21, that is, ts <t1, the output voltages V21, 22
Become high level, and Vref21 ≦ V14 ≦ V
When ref22, that is, when t1 ≦ ts ≦ t2, the output voltage V21 becomes a low level, the output voltage V22 becomes a high level, and V14> Vref22.
, That is, when ts> t2, the output voltages V21 and V22 are both at the low level.

The output voltages V21 and V22 from the comparators A11 and A12 are held in set-reset flip-flops FF1 and FF2 in the threshold value changing circuit 25, respectively. These set-reset flip-flops FF1 and FF2 are configured by NAND circuits, the output voltage V21 is input to the set input terminal of the set-reset flip-flop FF1, and the output voltage V22 is set-reset flip-flop. F
It is input to the reset input terminal of F2. A pulse from the differentiating circuit 24 is input to a reset input terminal of the set-reset flip-flop FF1 and a set input terminal of the set-reset flip-flop FF2.

The differentiating circuit 24 is a circuit for receiving the output voltage V13 of the detecting circuit 22 via the buffer B12 and detecting its falling timing, that is, the beginning of the pulse of the code signal. C14
, Resistors R13, R14, R15, R16, and comparator A
16 are provided. The non-inverting input terminal of the comparator A16 is connected to a voltage Vcc by a voltage dividing resistor R13, R14 interposed between a power supply of a predetermined voltage Vcc and a ground level.
15 = Vcc × R14 / (R13 + R14), and the output from the buffer B12 is input via the capacitor C14. On the other hand, the inverting input terminal of the comparator A16 is connected to the resistors R15 and R16 in the same manner as the non-inverting input terminal.
6 / (R15 + R16).

The resistance values of the resistors R13 to R16 are selected to be equal to each other.
= V16 = Vcc / 2. The time constant τ1 of the differentiating circuit constituted by the capacitor C14 and the resistors R13 and R14 is given by: τ1 = {C14 × (R13 × R14)} / (R13 + R
14) and is chosen so that τ1≪ts.

Therefore, the output voltage V1 of the comparator A16
Numeral 7 generates a low-level pulse at the timing when the pulse of the code signal is input to the differentiating circuit 24, that is, at the beginning of the pulse, and is stable at a high level in a steady state.

The comparator A16 includes voltage dividing resistors R13 and R14.
And a high-sensitivity differential amplifier that quickly detects the fall of the pulse of the code signal to the inputs balanced by R15 and R16, and can be configured, for example, as shown in FIG. FIG. 2 is an electric circuit diagram showing a specific configuration of the comparator A16. The comparator A16 includes transistors Q1 to Q4 and a constant current source 40.

The transistors Q1 and Q2 form a differential pair, the emitters of which are commonly connected to the power supply of the voltage Vcc via a constant current source 40, and the base thereof has a voltage dividing resistor R1.
3, R14; the voltages V15, 16 respectively generated by R15, R16 are provided. The collector of transistor Q2 is connected to the collector of transistor Q4, and the collector of transistor Q1 is connected to the base and collector of transistor Q3 and the base of transistor Q4. The emitters of the transistors Q3 and Q4 are both grounded. The emitter area ratio between the transistor Q3 and the transistor Q4 is formed to be n: 1 (n> 1).

Therefore, when the code signal pulse is not input, V15 = V16, so that the collector currents I41 and I42 of the transistors Q1 and Q2 are equal, while the emitter area ratio is n: 1. , The transistor Q4 becomes high impedance,
The output voltage V17 becomes high level.

On the other hand, k is Boltzmann's constant,
When T is an absolute temperature and q is a charge amount of electrons, if V15 = V16− (kT / q) × ln (n), then I41: I42 = n: 1, and the transistor Q4 becomes active.

Further, when V15 <V16− (kT / q) × ln (n), the transistor Q4 becomes saturated, and the output voltage V17 becomes low.

Therefore, the threshold voltage of the comparator A16 is
(KT / q) ln (n), and when there is no signal, (kT / q) ln (n)
/ Q) A comparator which internally has an offset voltage ΔV of ln (n) and always outputs a high level. When the offset voltage is, for example, n = 2 and the ambient temperature is 25 ° C., the offset voltage is about 18 mV, which makes it possible to realize a very small and very sensitive comparator circuit.

The output voltage V17 of the comparator A16 is input to the reset input terminal of the set-reset flip-flop FF1 and the set input terminal of the set-reset flip-flop FF2 in the threshold value changing circuit 25 as described above. . Therefore, at the moment the pulse is input,
Output voltage V of set-reset flip-flop FF1
31 is reset to a low level, the output voltage V32 of the set-reset flip-flop FF2 is set to a high level, and holds these states. Then, ts ≧
At time t1, the set-reset flip-flop FF1
Is set to a high level, and ts> t
When it becomes 2, the output voltage V32 of the set-reset flip-flop FF2 is reset to a low level.

The threshold value changing circuit 25 calculates the pulse width t.
The threshold voltage V10 output to the detection circuit 12 according to whether or not s is within the predetermined range of the time t1 to t2.
Is a circuit for changing This threshold changing circuit 25
Are the set-reset flip-flops FF1, F
F2, a NAND gate G, a comparator A17 that performs a detection operation similar to that of the detection circuit 22, a constant current source 28 and a capacitor C14 for integration, a comparator A18 that performs an integration operation like the integration circuit 23, Capacitor C1 for integration
5 and constant current source 29, constant current source 30 and switch 31 for changing the discharge time constant of capacitor C15.
And a buffer B13.

In this threshold value changing circuit 25, the set
Output voltage V of reset flip-flops FF1 and FF2
31 and V32 are input to the NAND gate G.
Therefore, the output voltage V33 of the NAND gate G becomes
After the pulse is detected, the signal is at a high level until ts <t1, and at a low level when t1 ≦ ts ≦ t2,
When s> t2, the level is high. In this way, it can be determined that the pulse width falls within the predetermined times t1 to t2.

On the other hand, the comparator A17 is different from the comparator A14.
The output voltage V1 is input to the inverting input terminal, the threshold voltage V10 is input to the non-inverting input terminal, and when V1> V10, the capacitor C14 A predetermined constant current I31 is sucked out, and when V1 ≦ V10, the sucking is stopped. The capacitor C14 also has a predetermined voltage Vcc.
From the power supply of the constant current source 32 via the constant current source 28,
It is always supplied.

The terminal voltage V34 of the capacitor C14, which changes by charging and discharging the currents I32 and I31, is input to the inverting input terminal of the comparator A18. This comparator A1
Similarly, reference numeral 8 denotes a current output type differential amplifier. A predetermined reference voltage Vref31 is input to a non-inverting input terminal of the differential amplifier. When V34 <Vref31, the capacitor C1
5 is supplied with a constant current I33, and when V34 ≧ Vref31, the supply is stopped. The capacitor C15 is provided with a constant current source 29 and a series circuit composed of the constant current source 30 and the switch 31 in parallel.
9, the constant current I34 is constantly sucked out, and when the output voltage V33 of the NAND gate G goes low and the switch 31 is turned on, the constant current I35 is sucked out by the constant current source 30. The terminal voltage of the capacitor C15 is accurately transferred to the threshold voltage V via the buffer B13.
It is output as 10.

Therefore, when V34 <Vref31 and the pulse width is within the predetermined range, the capacitor C15 is charged with a relatively small current of I33-I34-I35, and the threshold voltage V10 is lowered. On the other hand, even if V34 <Vref31, when the pulse width is not within the predetermined range, the capacitor C15 is connected to I
The battery is charged at 33-I34, and the threshold voltage V10 increases.

For example, the charging current I33 is 200 nA, the discharging current I34 is 10 nA, and the discharging current I3 is
5 is 100 nA. Therefore, when the pulse width ts of the code signal is within the predetermined range, V34 <Vre
ON during the period of f31, that is, substantially equal to the pulse width
Since the switch 31 conducts during the duty period, the capacitor C15 is charged at I33-I34-I35 = 90 nA. On the other hand, during the period of V34 ≧ Vref31, that is, during the OFF duty period, the capacitor 15 becomes I34 + I35 Discharge is performed at 110 nA.

Therefore, in this case, 110/200 =
From 0.55, the duty ratio of the code signal pulse may include 0.55 (55) including 0.4 or less, which is the duty ratio of the code signal of the infrared remote controller used for home appliances and the like.
%), The threshold voltage V10 output from the threshold changing circuit 25 does not reach the peak of the output voltage V1 from the bandpass filter BPF, so that the code signal up to the duty ratio can be received. .

As a result, when a code signal having a pulse width within a predetermined range is received, the threshold voltage V10 decreases, and the detection circuit 12 responds to the code signal to demodulate the pulse train of the code signal. Will be When the pulse width of the code signal is not within the predetermined range, V34 <V
Since the switch 31 is also shut off during the period of ref31, for example, as described above, I33 = 200 nA, I3
When 4 = 10 nA, the charging current of the capacitor C15 becomes I33-I34 = 190 nA, and
During the period of 34 ≧ Vref31, discharge is performed by the discharge current I34. Therefore, for a code signal having a duty ratio of 10/200 = 0.05 (5.0%) or more, the threshold voltage V10 can be the peak value of the output voltage V1 of the band-pass filter BPF. .

Normally, the duty ratio of the noise signal of disturbance light such as fluorescent light and sunlight is 0.1 (10%) or more. Therefore, if I33-I34: I34 is 10: 1 or more, all the external light A malfunction can be prevented with respect to noise.

FIGS. 3 to 5 are waveform diagrams for explaining the operation of the receiving apparatus 1 configured as described above. FIG. 3 shows a case where ts <t1, and FIG. 4 shows a case where t1 ≦ ts ≦. FIG. 5 shows a case where ts> t2. When the carrier component is detected, the hold voltage V1 of the buffer B11, which is the peak hold value of the output voltage V1 from the band-pass filter BPF and the voltage V11 obtained by dividing the output voltage V1
2 is as shown in FIGS. 3 (a), 4 (a) and 5 (a). Therefore, when the hold voltage V12 of the buffer B11 which is the peak hold value, the output voltage V1
And the terminal voltage V of the capacitor C12, which is the output of the detection circuit 22 that extracts the pulse component of the code signal,
13 is as shown in FIGS. 3 (b), 4 (b) and 5 (b).

The output voltage V14 of the integrating circuit 23 rises from the timing of the start of the pulse, and ts <t1
, The output voltage V, as shown in FIG.
14 does not reach the reference voltage Vref21 of the comparator A11, and when t1 ≦ ts ≦ t2, as shown in FIG. 4C, the output voltage V14 is equal to the two comparators A11 and A1.
2 are within the reference voltages Vref21 to Vref22, and when ts> t2, as shown in FIG. 5C, the output voltage V14 is equal to the reference voltage Vr of the comparator A12.
It becomes larger than ef22.

Therefore, the output voltage V21 from the comparator A11 is always at a high level in the case of FIG. 3 (f) where ts <t1, and FIG. 4 (f) where t1 ≦ ts ≦ t2.
And in the case of FIG. 5F where ts> t2,
It becomes low level only during this period. Further, the output voltage V22 from the comparator A12 is always at the high level in the case of FIG. 3 (g) where ts <t1 and in the case of FIG. 4 (g) where t1 ≦ ts ≦ t2, and ts> FIG. 5 at t2
In the case of (g), it is at the low level only during that period.

On the other hand, from the differentiating circuit 24, for a constant input voltage V16, the voltage V15 falls and rises at the beginning and end of the pulse, respectively.
(D), FIG. 4 (d) and FIG. 5 (d). Thus, the output voltage from the comparator A16 is
As shown in FIG. 3 (e), FIG. 4 (e) and FIG. 5 (e), a fall occurs at the beginning of the pulse.

Therefore, the output voltage V31 from the set-reset flip-flop FF1 remains reset to a low level at the beginning of the pulse in the case of FIG. 3 (h) where ts <t1, and t1 ≦ ts ≤t
4 and FIG. 5 where ts> t2.
In the case of (h), after resetting to the low level at the start end, the level is set to the high level when V14 ≦ Vref21. The output voltage V32 from the set-reset flip-flop FF2 is ts <t
In the case of 1 and in the case of t1 ≦ ts ≦ t2, as shown in FIG. 3 (i) and FIG. 4 (i), the level is always high, and in the case of ts> t2, FIG. When V14> Vref22 as shown by, it is reset to the low level.

As a result, when ts <t1, the output voltage V33 from the NAND gate G rises to the high level at the beginning of the pulse as shown in FIG. In the case of t1 ≦ ts ≦ t2, as shown in FIG. 4 (j), the signal is raised to a high level at the start end, and then lowered to a low level when ts = t1, and In the case of> t2, as shown in FIG. 5 (j), the pulse rises to the high level at the beginning of the pulse, and once falls to the low level when the time t1 has elapsed, and then when the time t2 has elapsed To bring it back to high level.

Therefore, as shown in FIG.
Only when 1 ≦ ts ≦ t2, the terminal voltage of the capacitor C15 decreases, the threshold voltage Vth decreases, and the terminal voltage V2 from the comparator A3 falls to a low level as shown in FIG. A pulsation corresponding to the carrier frequency will occur. The pulsation component is level-discriminated by the comparator A4, and the output signal V3 output is as shown in FIG. On the other hand, when ts <t1 and ts> t2, the reference voltage Vth increases, and the output voltage V1 is set to the level as shown in FIGS. 3 (k) and 5 (k). 5 cannot be discriminated, and therefore the terminal voltage V2 from the comparator A3 is
It remains at the high level as shown in (l). As a result, the output signal V3 also remains at the low level as shown in FIG. 3 (m) and FIG. 5 (m).

As described above, only when t1 ≦ ts ≦ t2, the threshold voltage Vth decreases, and decoding of the code signal pulse is permitted. Otherwise, decoding of the pulse is prohibited. Will be.

FIGS. 6 to 9 are diagrams showing output waveforms of the band-pass filter BPF when the main disturbance light is input to the infrared remote controller. 6 shows the case of fluorescent lamp light that is lit by an impulse pulse, FIG. 7 shows the case of fluorescent lamp light that is lit by a commercial AC power supply, FIG. 8 shows the case of sunlight, and FIG. 9 shows the frequency of 40 to 70 kHz. Shows the case of inverter fluorescent light lit by. As shown in these figures, assuming that the level of about half of the peak value is the reference value, that is, the voltage divided value by the voltage dividing resistors R11 and R12, the pulse width in the example of FIGS. 200μse
It can be understood that the pulse width is larger than 800 μsec in the case of the continuous pulse in the example of FIG.

On the other hand, the pulse width of the code signal is
Except for a read pulse portion called a header pulse, all are within the range of 200 to 800 μsec. Therefore, by setting t1 = 200 μsec and t2 = 800 μsec, it is possible to judge these disturbance lights as noise and prohibit the output of the code signal, thereby preventing malfunction.

The header pulse has a pulse width of several msec. The pulse width determination circuit 13 shuts off the switch 31 and the threshold voltage V10 rises toward the peak value of the header pulse signal. As described above, for example, I33 = 200 nA, I34 = 10 n
If the peak value Vp of the header pulse is 100 mV by setting A, C15 = 0.1 μF, the time required to reach the peak value is (C15 × Vp) / (I33−I34) = 50 (mse
c) only

Therefore, the threshold voltage V10 does not reach its peak value during the period of the header pulse, and the header pulse can be received. In this way, malfunction of the infrared remote controller due to disturbance light can be prevented.

The present invention is not limited to the infrared remote controller,
The present invention can be suitably implemented when the code signal and the disturbance light can be separated by the pulse width.

[0074]

As described above, the optical signal demodulator of the remote control device according to the first aspect of the present invention is constructed such that a pulse of a specific frequency is used as a carrier wave and the pulse is modulated in accordance with control information. In the optical signal demodulation device such as a so-called infrared remote control receiving device that receives the optical code signal and demodulates the code signal corresponding to the control information, only when the pulse width of the code signal is within a predetermined range. ,
The output of the code signal is allowed.

Therefore, even if disturbing light having a frequency close to the carrier frequency and a duty ratio larger than the duty ratio of the code signal is input to the light receiving element, the pulse width obtained by level-discriminating the disturbing light is as described above. Since the output of the code signal is out of the predetermined range and the output of the code signal in response to the input is prohibited, it is possible to improve the optical noise separation characteristics and prevent a malfunction.

Further, in the optical signal demodulation device of the remote control device according to the second aspect of the present invention, as described above, the pulse width determination means creates a voltage corresponding to the pulse width of the code signal by using the integrator. The voltage is window-compared by the first and second comparators to determine whether or not the pulse width is within a predetermined range. The threshold for demodulating the output of the pass filter is lowered to enable demodulation of the code signal.

Therefore, the permission / inhibition control of the output of the code signal by determining the pulse width as described in claim 1 can be specifically realized.

Further, according to the optical signal demodulation device of the remote control device according to the third aspect of the present invention, when the voltage corresponding to the pulse width is window-compared by the first and second comparators, When the voltage rises and falls, a period that exists within the range of the two threshold voltages is generated. On the other hand, the first and second set-reset flip-flops are used to increase the voltage only when the voltage rises. It is determined that the values are within the range of two threshold voltages.

Therefore, the determination accuracy can be further improved.

Further, the optical signal demodulation device of the remote control device according to the fourth aspect of the present invention includes the third comparator for detecting the leading edge of the code signal, and the predetermined offset voltage therein. A differential amplifier having a pair of bias resistors for biasing a pair of input terminals of the differential amplifier to the same voltage, and an AC coupling capacitor for differentiating and inputting a code signal from the detection circuit to one input terminal. It has and is comprised.

Therefore, when a pulse corresponding to the beginning of the pulse of the code signal is input, the output can be switched quickly, and a highly sensitive comparator can be realized.

Further, in the optical signal demodulation device for a remote control device according to the fifth aspect of the present invention, as described above, the ratio between the charge current and the discharge current of the hold capacitor is set to 10: 1.
Above.

Therefore, the terminal voltage of the hold capacitor is substantially the peak output of the bandpass filter, and malfunction due to disturbance light having a low duty ratio can be almost completely prevented.

According to the optical signal demodulating device of the remote control device according to the sixth aspect of the present invention, as described above, the predetermined range is set to 200 to 800 μsec, which is the pulse width of the code signal used in the infrared remote control device. And

Therefore, the pulse train of the code signal in the infrared remote controller of the home electric appliance or the like can be demodulated separately from the noise signal of the fluorescent lamp or the like.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating an electrical configuration of a receiving device of an infrared remote controller according to an embodiment of the present invention.

FIG. 2 is an electric circuit diagram showing a specific configuration of a comparator of a differentiating circuit in the receiving device.

FIG. 3 is a waveform diagram for explaining an operation in a state where the pulse width of the code signal is smaller than a shorter threshold value.

FIG. 4 is a waveform diagram for explaining an operation in a state where the pulse width of the code signal is within a predetermined threshold.

FIG. 5 is a waveform diagram for explaining an operation in a state where the pulse width of the code signal is larger than a longer threshold value.

FIG. 6 is a diagram illustrating a bandpass filter output waveform of fluorescent lamp light that is turned on by an impulse pulse, which is an example of disturbance light.

FIG. 7 is a diagram showing a bandpass filter output waveform of fluorescent light lit by a commercial AC power supply, which is an example of disturbance light.

FIG. 8 is a diagram illustrating an output waveform of a bandpass filter of sunlight, which is an example of disturbance light.

FIG. 9 is a diagram illustrating a bandpass filter output waveform of inverter fluorescent light, which is an example of disturbance light.

FIG. 10 is a block diagram showing the electrical configuration of a typical prior art infrared remote control receiver.

11 is a waveform chart for explaining the operation of the receiving apparatus shown in FIG.

[Explanation of symbols]

 Reference Signs List 11 receiving apparatus (optical signal demodulating apparatus) 12 detecting circuit (demodulating means) 13 pulse width determining circuit 16 offset voltage source 21 peak hold circuit 22 detecting circuit 23 integrating circuit 24 differentiating circuit 25 threshold changing circuit A1 preamplifier (demodulating means) A2 Amplifier (demodulation means) A3, A4, A11 to A18 Comparator BPF Bandpass filter C1 AC coupling capacitor G NAND gate FF1, FF2 Set-reset flip-flop PD Photodiode R11, R12 Voltage dividing resistor

Claims (6)

[Claims] An optical code signal generated by modulating a pulse corresponding to control information with a pulse having a specific frequency as a carrier is received by a light receiving element. An apparatus for demodulating a photoelectric conversion output by a demodulating means and demodulating a code signal corresponding to the control information, wherein it is determined whether a pulse width of the code signal is within a predetermined range and the predetermined value is determined. An optical signal demodulation device for a remote control device, comprising: pulse width determination means for permitting output of the code signal within a range and prohibiting output of the code signal outside the predetermined range. 2. The band-pass filter according to claim 1, wherein the pulse width determination unit receives a carrier frequency component of a photoelectric conversion output of the light receiving element, which is amplified by an amplifier of the demodulation unit and further extracted by a band-pass filter. A peak hold circuit that integrates or peak-holds the divided voltage output of the above, and creates a reference discrimination level corresponding to the received light level, and discriminates the output of the band-pass filter at the reference discrimination level, removes a carrier component, and executes the code. A detection circuit for detecting a signal; an integrator for integrating a code signal from the detection circuit to generate a voltage corresponding to the pulse width; and an output of the integrator, the first of which has the shorter pulse width. And first and second comparators comparing two voltages respectively corresponding to a threshold value of the first and second longer threshold values, and outputs of the first and second comparators. And when the pulse width is within the range of not less than the first threshold and not more than the second threshold, the demodulating means demodulates the output of the band-pass filter into the pulse signal by level discrimination. 2. An optical signal demodulation device for a remote control device according to claim 1, further comprising a threshold changing means for lowering a threshold for performing the operation. 3. A threshold comparator, comprising: a third comparator for differentiating a code signal from the detection circuit to detect a leading edge of the code signal; and a low level output from the third comparator. A first set-reset flip-flop that is reset to a high level and is set to a high level by an output from the first comparator; and is set to a high level by an output from the third comparator; A second set-reset flip-flop that is reset to a low level by an output from the comparator, and when the outputs from the first and second set-reset flip-flops are both at a high level, 3. The optical signal demodulation device according to claim 2, further comprising: discharging means for discharging the held capacitor to lower the threshold value. Place. 4. The third comparator includes: a differential amplifier having a predetermined offset voltage therein; a pair of bias resistors for biasing a pair of input terminals of the differential amplifier to the same voltage; 4. The optical signal demodulator according to claim 3, further comprising an AC coupling capacitor for differentiating and inputting the code signal from the detection circuit at an input terminal. 5. The optical signal demodulator according to claim 3, wherein a ratio between a charge current and a discharge current of the hold capacitor is 10: 1 or more. 6. The predetermined range is a pulse width of a code signal used in an infrared remote control device.
The optical signal demodulation device for a remote control device according to claim 1, wherein the optical signal demodulation time is 00 μsec.