JPH10191270A - Television receiver - Google Patents

Abstract

PROBLEM TO BE SOLVED: To reproduce a transmission signal obtained by multiplexing a video signal with a digital signal by reducing hindrance between the signals by providing first and second reproduction systems, permitting the second reproduction system to synchronously detect the transmission signal, to demodulate the digital signal and to reproduce the outputted signal to a signal before a polarity is inverted. SOLUTION: The signal of a sound signal band which is multiplexed and transmitted by BPF 414 and which is digitally encoded is extracted from a signal converted into an intermediate frequency in a frequency conversion circuit 403. The extracted signal is synchronously detected by the carrier reproduced in a carrier reproduction circuit 415 and a signal which is orthogonal and is multiplexed/transmitted is detected/demodulated in a synchronism detection circuit 415. The demodulation signal is processed in a spectrum suppression processing signal reproduction circuit and it is returned to a signal before a spectrum suppression processing is executed. An error occurred in the middle of transmission is detected and corrected in a digital signal processing circuit 418. The digital signal after correction is returned to an analog signal in DAC 419 and it is outputted from an output terminal 420.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a television receiver, and more particularly to a television receiver for receiving a transmission signal in which a carrier of a video signal and a carrier of a digital signal are orthogonally multiplexed.

[0002]

2. Description of the Related Art Conventionally, a method of multiplexing a digitally encoded audio signal (hereinafter abbreviated as PCM audio) to a video signal and transmitting the video signal is described in Japanese Patent Laid-Open No. It is reported in the Technical Survey Report Part 1 "Satellite Broadcasting Receiver". The contents are shown below.
5.72 for NTSC video signals up to 4.2 MHz
MHz subcarriers are multiplexed at different frequencies. The sub-carrier is QPSK-modulated with PCM voice. However, this method cannot be performed while maintaining compatibility with the current terrestrial television broadcasting because the subcarrier frequency is outside the band of the current terrestrial television broadcasting.

A method of multiplexing another signal to the current terrestrial television broadcasting is described in Japanese Patent Publication No. 2 (published by the Japan Broadcasting Publishing Association in January 1983). It is described on pages 205 to 208. However, there is no description of a method for obtaining a transmission rate of about 1 Mbit / sec or more necessary for transmitting high-quality PCM voice.

[0004]

The above-mentioned prior art is about 1
No consideration is given to multiplexing a signal with a transmission rate of megabits / second to the current terrestrial television broadcast and transmitting it, and there is a problem that high-quality PCM audio cannot be multiplexed and transmitted.

An object of the present invention is to provide a television receiver capable of receiving a transmission signal transmitted while making a large-capacity digital signal compatible with a current television broadcast and reducing interference between the two signals. Is to do.

[0006]

To achieve the above object, a television receiver according to the present invention comprises: a first modulated wave obtained by amplitude-modulating a first carrier with a video signal; And a second carrier having a substantially quadrature relationship with
A television for receiving a vestigial sideband transmission signal in which a second modulated wave amplitude-modulated with a digital signal whose polarity is inverted every period corresponding to one horizontal scanning cycle of the video signal is multiplexed. A receiver for reproducing a video signal included in the received transmission signal; a second reproduction system for reproducing a digital signal included in the transmission signal; Is a demodulation means for synchronously detecting the transmission signal to demodulate the digital signal, and a reproduction process for reproducing the digital signal output from the demodulation means into a signal before the polarity is inverted. And a reproducing means for performing the reproduction.

[0007] The transmission signal may include a third modulated wave modulated by an analog audio signal.

[0008] The digital signal may be a PCM audio signal. Further, the polarity of the digital signal is preferably inverted for each period corresponding to one horizontal scanning period in a period corresponding to two horizontal scanning periods of the video signal.

In the case of reproducing such an inverted digital signal, prior to performing the actual reproduction processing,
It is preferable that the signal output from the demodulation unit is delayed for a period corresponding to one horizontal scanning period, and the delayed signal and the signal before the delay are subtracted from each other.

In the case of reproduction, a signal obtained by subtracting the same data having different polarities from each other is selected from the signals subjected to the subtraction processing, and the selected signal may be extended in the time axis.

[0011]

According to the present invention, a first modulated wave amplitude-modulated by a video signal and a second modulated wave modulated by a digital signal such as a PCM sound are orthogonally multiplexed and transmitted. Thus, a digital signal having a high transmission rate can be transmitted within the band of the current television broadcast. When the transmission signal is demodulated by a television receiver, interference between multiplexed signals can be reduced by performing synchronous detection.

In addition, if the polarity of the transmitted digital signal is inverted for the same data during a period corresponding to one horizontal scanning cycle during a period corresponding to two adjacent horizontal scanning periods, the detection method of the receiver is an envelope detection method. In the case of the system or the pseudo-synchronous detection system, the phase fluctuation of the chrominance subcarrier caused by the signal multiplexed in the orthogonal relationship becomes the opposite phase, and the hue fluctuation on the screen can be reduced.

When demodulating and reproducing such a digital signal whose polarity has been inverted, the demodulated digital signal is delayed by a period corresponding to one horizontal scanning period by the delay means, and the delayed signal and the undelayed signal are delayed. (Input signal of the delay means) is subtracted by the arithmetic processing means, whereby the amplitude of the signal data is increased by a factor of two, but the noise is increased only by a factor of two due to its randomness. The noise-to-noise ratio can be improved by about 3 dB. Further, if such a subtraction process is performed, the interference from the video signal leaked due to a ghost or the like is offset by the correlation of the video signal in each horizontal scanning period (vertical correlation on a television screen). Therefore, interference from a video signal can be reduced.

Further, if a signal in which identical data having different polarities are subtracted from each other is selected from the signal subjected to the subtraction processing and the selected signal is expanded on the time axis, the same data is previously stored in one horizontal line to reduce interference. It is possible to easily reproduce a digital signal processed so that the polarity is inverted every scanning cycle.

[0015]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is an example of a block diagram of a transmission signal generation device for television signal transmission embodying the present invention.

Reference numeral 101 denotes an audio signal input terminal, and reference numeral 102 denotes an FM.
Modulator 103, audio signal carrier generation circuit 104, video signal input terminal 104, matrix circuit 105, luminance signal processing circuit 106, color difference signal processing circuit 107, 108
Is an addition circuit, 109 is a video modulation circuit, 110 is a video signal carrier generation circuit, 111 is an input terminal of an audio signal to be digitally encoded and transmitted, 112 is an analog / digital conversion circuit (hereinafter abbreviated as ADC), 113 is a digital signal A processing circuit 114; a spectrum suppression processing circuit 115;
Is a low-pass filter, 116 is a phase shift circuit, 117 is a modulation circuit, 118 is an equalizer, 119 is an addition circuit, 120
Is a transmission VSB filter for residual sideband amplitude modulation, 121
Is an adder, and 122 is an antenna.

The carrier for the audio signal generated by the audio signal carrier generation circuit 103 is frequency-modulated in the FM modulator 102 by the audio signal applied to the audio signal input terminal 101. The RGB three primary color signals of the video signal applied to the video signal input terminal 104 are
At 5, the signal is divided into a luminance signal and a color difference signal. These signals are processed by a luminance signal processing circuit 106 and a color difference signal processing circuit 107, respectively, and then added by an addition circuit 108, and NT
Processed into a composite video signal such as the SC system. The video signal carrier generated by the video signal carrier generation circuit 110 is amplitude-modulated in the video modulation circuit 109 by the processed video signal. The modulated signal is limited to a television transmission bandwidth by the transmission VSB filter 120 for vestigial sideband amplitude modulation, and then added to the audio signal carrier modulated by the audio signal by the addition circuit 121,
The signal is transmitted from the antenna 122. The above description is that of the current apparatus for generating a terrestrial television broadcast transmission signal.

The parts related to the present invention will be described below.
The multiplexed audio signal applied to the input terminal 111 is converted to a digital code by the ADC 112. The converted signal is subjected to so-called digital signal processing in which a code for error detection and correction is added or interleaved so that an error occurring during transmission can be detected and corrected by a receiver on the reproduction side in a digital signal processing circuit 113. You. In the processed signal, low frequency components and the like among the signal components are suppressed by the spectrum suppression processing circuit 114. Unnecessary high-frequency components of the suppressed signal are deleted by the low-pass filter 115. The carrier for the video signal generated by the video signal carrier generation circuit 110 is changed in phase by 90 degrees by the phase shift circuit 116 and then the modulation circuit 11
At 7, the signal is modulated by a signal from which unnecessary high-frequency components output from the low-pass filter 115 have been removed. The amplitude characteristic of the modulated signal is corrected by the equalizer 118.
The corrected signal is added and combined with the modulated carrier for the video signal in the adder circuit 119 and transmitted from the antenna 122. The amplitude characteristic for correction of the equalizer 118 is a characteristic symmetrical with respect to the amplitude characteristic of the IF Nyquist filter provided in the video signal intermediate frequency stage of the television receiver with respect to the video carrier frequency. Equalizer 1
Reference numeral 18 is for correcting a change from the quadrature phase of the multiplex transmission wave by the IF Nyquist filter of the television receiver on the transmission side. Regarding the phase relationship of the video signal after passing through the IF Nyquist filter of the television receiver and before detection, the modulated wave of the signal multiplexed and transmitted with respect to the video signal carrier has a quadrature phase. If the addition circuit 119 adds a small number of multiplexed signals to the video signal, it is possible to reduce interference from the multiplexed signal on the video signal detected by the television receiver.

FIG. 2 is a spectrum diagram of the transmission signal and the like generated in FIG. 201 is the spectrum of the transmission signal of the video signal, 202 is the spectrum of the audio signal that is FM-modulated and transmitted, 203 is the spectrum of the multiplexed signal after the spectrum suppression processing and unnecessary high-frequency components have been removed, 204
Is the spectrum of the multiplexed signal after modulation, 205 is the spectrum of the multiplexed signal whose amplitude characteristics have been corrected by the equalizer 118, 206 is the Nyquist filter characteristic of the receiver, 207 is the general pseudo-synchronous detection type television receiver. It is an amplitude characteristic of a band-pass filter for carrier wave reproduction.

The spectrum 201 of the transmission signal of the video signal
Is lower than the video signal carrier frequency f0.
A frequency f2 or less and a high frequency f0 + f3 or more have characteristics attenuated by the amplitude characteristic of the VSB filter 120 for obtaining a vestigial sideband amplitude modulated wave. The voice signal spectrum 202 that is FM-modulated and transmitted exists around the voice signal carrier f1. On the other hand, as shown in spectrum 203, the spectrum of the signal to be multiplex-transmitted is suppressed by the spectrum suppression processing circuit 114 so as to suppress low-frequency components lower than the frequency f4, and the low-pass filter 115 removes unnecessary high-frequency components higher than the frequency f5. ing.
This is the so-called processed spectrum of the baseband signal. The spectrum after being modulated by the modulation circuit 117 has a characteristic symmetric about f0 as shown in a spectrum 204. The amplitude characteristic of the equalizer 118 is the frequency f
Since the IF Nyquist filter characteristic 206 of the television receiver having a gradient from 0-f6 to f0 + f6 and the characteristic symmetrical about the video carrier frequency f0 are centered, the spectrum of the output signal of the equalizer 118 has a high frequency as shown in the spectrum 205. It is attenuated. Spectrum 2
An addition circuit 119 adds the video signal modulation wave indicated by 01 and the multiplexed signal modulation wave indicated by the spectrum 205. Further, the audio signal modulated wave represented by the spectrum 202 is added by the adding circuit 121 and transmitted.

FIG. 3 is a vector diagram showing the phase relationship between the video signal modulated wave and the signal modulated wave multiplexed and transmitted. Reference numeral 301 denotes a vector of a video signal carrier, 302 and 303 denote vectors of multiplexed signal modulated waves, and 304 and 305 denote composite vectors thereof. At a frequency within ± f2 with respect to the video carrier frequency f0, the video signal modulation wave is a general amplitude modulation wave and is indicated by a vector 301. In a signal multiplexed and transmitted in a frequency band equal to or lower than f2, orthogonal carriers are modulated into amplitudes A and -A corresponding to digital codes 1 and 0, and are represented by vectors 302 and 303, respectively. Assuming that the amplitude of the carrier of the video signal is 1, the combined and transmitted signal E is E = cosωct ± Asinωct (1). Here, ωc is the angular frequency of the video carrier, and t is the time. This is represented by the combined vectors 304 and 30 in FIG.
When viewed corresponding to 5, E = √ (1 + A2) cos (ωct− (± θ)) (2) Here, θ = tan ~ 1 (A) (3).

Here, the interference of the signal reproduced and received by the television receiver with the multiplexed signal will be considered. When the video signal detection circuit is a synchronous detection circuit, the video signal detection output signal is cosω regardless of the value of A.
Since the signal is only the coefficient of ct (only the video signal), there is no interference in principle. When the video signal detection circuit is an envelope detection circuit, the disturbance to the video signal detection output signal changes depending on the value of A, and the lower the A, the less the disturbance. Assuming that A is 0.1 as an example, E = √ (1 + A2) ≒ 1.005 (4) This is 0.005 to 1 (about -46 d
This indicates that the signal B) has an effect. We believe that this interference does not pose a practical problem since the S / N ratio of the video signal can be 40 dB or more. On the other hand, interference from the video signal to the digitally encoded audio is eliminated by using a synchronous detection circuit in the detection circuit for reproducing the digitally encoded audio. We also consider here the transmission SN ratio of the multiplexed signal, whether the multiplexed signal is reproduced at the receiver due to the low level of the multiplexed signal. Video signal SN ratio is 40
In the case of dB, since the transmission bandwidth of the digitally encoded audio signal is about 1/4 of the bandwidth of the video signal, the transmission SN ratio of the digitally encoded audio signal is 46 dB, and A is 0.1 , The transmission SN becomes 26 dB. The relationship between the SNR of the digital signal and the bit error rate is 10 to 4 at a SNR of 17.4 dB for a general binary signal.
Therefore, the signal having the SN ratio of 26 dB is a signal of good quality that can be sufficiently reproduced.

The amplitude characteristic of the band-pass filter of the carrier recovery circuit for pseudo-synchronous detection of a television receiver having a video signal detection circuit of the pseudo-synchronous detection method has a frequency f0 ± f7 as shown by an amplitude characteristic 207. Is a characteristic that allows the light to pass through. The modulated spectrum 205 of a signal multiplexed and transmitted after processing such as spectrum suppression processing has a frequency f0 ± f4.
Since the signal component is suppressed in the frequency band within, the detected video signal is less disturbed by the multiplex signal.

As described above, according to the present embodiment, the signal to be multiplexed is spectrally processed and the multiplex level is set to be smaller than the video signal, so that the interference with the current terrestrial television broadcast can be reduced. There is.

FIG. 4 is a block diagram of a television transmission signal reproducing apparatus embodying the present invention. 401 is an antenna, 402 is a high-frequency amplifier, 403 is a frequency converter, 404 is an IF Nyquist filter, 405 is an intermediate frequency amplifier, 406 is a video signal detector, 407 is a video signal amplifier, 408 is a color difference signal demodulator, 409 is a primary color signal demodulation circuit, 410 is a cathode ray tube, 411 is an audio intermediate frequency amplification circuit, 412 is an audio FM detection circuit, 413 is an audio signal output terminal, 414 is a band pass filter, 415
Is a synchronous detection circuit, 416 is a carrier recovery circuit, 417 is a spectrum suppression processing signal recovery circuit, 418 is a digital signal processing circuit, 419 is a digital / analog conversion circuit (hereinafter abbreviated as DAC), and 420 is digitally encoded and transmitted. Output terminal for reproduced audio signal.

The television signal input from the antenna 401 is amplified by the high frequency amplifier circuit 402 and then converted by the frequency conversion circuit 403 into an intermediate frequency for demodulation. The converted signal passes through the IF Nyquist filter 404 and is amplified by the intermediate frequency amplifier 405. Tuning is performed by changing the oscillation frequency of a local oscillator inside the frequency conversion circuit 403. A video signal is converted from a signal amplified by the intermediate frequency amplification circuit 405 to a video signal detection circuit 406.
Detected at The obtained video signal is supplied to the video signal amplifying circuit 4
07, and a color difference signal is obtained by the color difference signal demodulation circuit 408 from the signal. The signal and the video signal amplification circuit 407
Are converted into three primary color signals of RGB by a primary color signal demodulation circuit 409. The three primary color signals are applied to the cathode ray tube 410, and an image is obtained on the cathode ray tube. The audio band signal among the output signals of the intermediate frequency amplifier 405 is amplified by the audio intermediate frequency amplifier 411. The output signal is audio FM
The signal is FM-detected by a detection circuit 412 and becomes an audio signal. The audio signal is output from an audio signal output terminal 413. The above description is that of the current terrestrial television broadcast playback device.

The part related to the present invention will be described below.
From the signal converted to the intermediate frequency by the frequency conversion circuit 403, a signal in a digitally encoded audio signal band multiplex-transmitted by the band-pass filter 414 is extracted. The extracted signal is synchronously detected by the synchronous wave detection circuit 415 with the carrier reproduced by the carrier wave regeneration circuit 416, and the orthogonally multiplexed signal is detected and demodulated. The demodulated signal is processed by a spectrum suppression processing signal reproduction circuit, and is restored to a signal before the spectrum suppression processing. After that, the digital signal processing circuit 418 detects and corrects an error occurring during the transmission of the signal. The corrected digital signal is returned to an analog signal by the DAC 419. The analog signal is output from the output terminal 420.

As described above, according to the present embodiment, since the reproducing device corresponding to the transmission signal generating device is provided, the multiplexed signal can be demodulated.

FIG. 5 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention.
1 have the same functions. 114 is a spectrum suppression processing circuit, 501 is a delay circuit, and 502 is a subtraction circuit. A specific example of the spectrum suppression processing circuit 114 is shown by a delay circuit 501 and a subtraction circuit 502.

FIG. 6 is a waveform diagram for explaining the operation of FIG.
Reference numeral 601 denotes input general binary digital data;
02 is the output waveform of the delay circuit 501, 603 is the subtraction circuit 5
02 is an output waveform. Here, the delay time τ is the same as or shorter than the minimum inversion interval time T of digital data.

Τ ≦ T (5) When τ is equal to T, the required transmission band does not increase. FIG. 6 shows an example. The input general digital data 601 is subtracted from the output waveform 602 output from the delay circuit 501 in the subtraction circuit 502 to become an output waveform 603. Waveform 603 is digital data 601
At the rising edge such as times T1, T5, and T6, and lasts for a time τ, at times T3, T6, and the like, it becomes Low at time T and continues for a time τ, and at other times from time T2 to time T3 or from time T4 to time T5. The period is the midpoint. The low frequency component of the waveform 603 is smaller than that of the data 601. The carrier wave output from the phase shift circuit 116 is modulated by the modulation circuit 117 so that no modulation is performed at the midpoint of + A at the High and -A at the Low of the waveform 603, so that the modulated wave of the multiplexed signal after modulation is obtained. In the vicinity of the carrier frequency is suppressed.

According to the present embodiment, since the components near the carrier frequency of the multiplex-transmitted signal are suppressed, the effect of the multiplex-transmitted signal on the current terrestrial television broadcast can be reduced.

FIG. 7 is a block diagram showing another example of a television transmission signal reproducing apparatus embodying the present invention. FIG.
The same reference numerals indicate the same functions. 417 is a spectrum suppression processing signal reproduction circuit, 701 is a ternary identification circuit, 70
2 is a ternary / binary conversion circuit, 703 is a code identification circuit, 704
Is a clock recovery circuit. Since the signal transmitted as shown by the waveform 603 in FIG. 6 has three values,
The reproducing device is provided with a ternary / binary conversion circuit and the like. The waveform detected by the synchronous detection circuit 415 is a ternary identification circuit 70.
The data is converted into ternary digital data of +1, 0, -1 by 1 and converted into binary digital data by the ternary / binary conversion circuit 702. The digital signal is sent to the code identification circuit 70
3 is strobed by the clock recovered by the clock recovery circuit 704.

FIG. 8 shows a specific example of the ternary identification circuit 701 shown in FIG. FIG. 9 is a waveform diagram for explaining the operation of FIGS. 701 is a ternary identification circuit, 702 is a ternary / binary conversion circuit, 801 is an input terminal, 802 is a capacitor, 8
03 is a resistor, 804 is an amplifier, 805 and 806 are voltage comparison circuits, 807 and 808 are reference voltage sources, 809 is a set / reset circuit, and 810 is a binary digital data output terminal. 901 is a waveform, 902 and 903 are reference power supply voltage values, 904 is an output waveform of the voltage comparison circuit 805, 905
Is an output waveform of the voltage comparison circuit 806, 906 is an output waveform of the set / reset circuit 809, 907 is a clock timing pulse obtained by the clock recovery circuit 704, 908
Is an output waveform of the code identification circuit 703.

The output signal of the synchronous detection circuit 415 is applied to an input terminal 801. The DC voltage of the signal is cut off by the capacitor 802 and the resistor 803, and the signal is amplified by the amplifier 804 to become a signal as shown in a waveform 901. The signal of this waveform 901 is the voltage comparison circuit 805 and the voltage comparison circuit 8
06, the voltage is compared with the reference voltage V1 or the reference voltage V2.
5 and the signal indicated by the waveform 905 is obtained at the output of the voltage comparison circuit 806. These signals are applied to the set / reset circuit 809, and the signal indicated by the waveform 906 is obtained at the output of the set / reset circuit 809. This signal is obtained at the output terminal 810 as an output signal of the ternary / binary conversion circuit 702. At time T1, the waveform 901 exceeds the reference voltage V1, the waveform 904 becomes High, the set / reset circuit 809 is set, and the waveform 906 becomes High.
h. At time T3, the waveform 901 falls below the reference power supply V1, and the waveform 904 becomes Low.
6 remains as it is. At time T4, the waveform 901 falls below the reference voltage V2 and the waveform 905 goes high, so that the set / reset circuit 809 is reset and the waveform 906 goes low. At time T6, the waveform 901 exceeds the reference voltage V2, and the waveform 905 becomes Low.
Remain low, the waveform 906 remains low.
At time T7, the waveform 901 exceeds the reference voltage V1 and the waveform 904
Becomes High, and the waveform 906 becomes High. With the above operation, binary digital data as shown by a waveform 906 can be demodulated. The binary digital data represented by the waveform 901 is converted by the code identification circuit 703 into a clock timing pulse 90 reproduced by the clock reproduction circuit 704.
7, and becomes binary digital data of a waveform 908. The waveform 906 is strobed into a waveform 908 at the time indicated by the arrow of the clock timing pulse 907 indicated by the times T2, T4, T8 and the like. Strobe data at the time indicated by the arrow, as shown in FIG.
And the time farthest from the reference voltage V2, and the time when the probability of occurrence of an error due to noise or the like is the smallest.

As described above, according to the embodiment shown in FIGS. 7 to 9, the transmission signals shown in FIGS. 5 and 6 can be reproduced.

FIG. 10 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention. 1 have the same functions. 114 is a spectrum suppression processing circuit, and 1001 is a digital modulation circuit. The output signal of the digital signal processing circuit 113 is subjected to digital modulation such as a digital FM modulation method by a digital modulation circuit 1001 to suppress low frequency components of the signal. As a result, according to the present embodiment, as in the case of FIG. 1, there is an effect that interference with the current terrestrial television broadcast can be reduced. A detailed description of a digital modulation method such as a digital FM modulation method in which low-frequency components are suppressed is described in "Modulation Method of Digital Magnetic Recording" of Nikkei Electronics P126 to P164 of 1978, 12, and 11. .

FIG. 11 is another example of a block diagram of a transmission signal reproducing apparatus for television signal transmission according to the present invention. 4 have the same functions. 417 is a spectrum suppression processing signal reproduction circuit, 1101 is a code identification circuit, 1102 is a clock reproduction circuit, and 1103 is a digital demodulation circuit. The output signal of the synchronous detection circuit 415 is sent to the code identification circuit 1101 by the clock recovery circuit 11.
The digital signal is strobed at a timing with a low probability of occurrence of an error caused by noise or the like by the clock signal reproduced in step 02. This digital signal is returned to digital data before digital modulation processed on the transmission side in the digital demodulation circuit 1103. By the operation of the spectrum suppression processing signal reproduction circuit 417, the signal which has been subjected to the spectrum suppression processing and transmitted can be reproduced. As described above, according to the present embodiment, there is an effect that the transmission signal shown in FIG. 10 can be reproduced.

FIG. 12 shows the spectrum suppression processing circuit 1 of FIG.
FIG. 14 is a block diagram of another specific example 14; 114 is a spectrum suppression processing circuit, 1201 is an input terminal, 1202 is a time axis compression circuit, 1203 is a timing generation circuit, 120
4 is an inverter, 1205 is a delay circuit, 1206 is a changeover switch, and 1207 is an output terminal.

As a specific example of the spectrum suppression processing circuit,
The continuous data applied to the input terminal 1201 is converted into intermittent data by the time axis compression circuit 1202,
4. The same data inverted between the intermittent data is obtained by the delay circuit 1205 and the changeover switch 1206. As a result, the same data is inverted and transmitted at certain intervals, so that low-frequency components near the DC of the averaged signal and components near the frequency indicated by the reciprocal of the interval at the fixed interval are suppressed. Is done.

FIG. 13 is an example of a transmission data string for explaining the operation of FIG. 1301 is a data string of the input terminal 1201, 1302 is an output data string of the time axis compression circuit 1202, 1303 is an inverter 1204 and a delay circuit 1205
, An output data string 1304 of the changeover switch 1206, and 1305 a timing waveform.

Data string 1 applied to input terminal 1201
301 is time-axis-compressed into a data string 1302 by the timing of the time axis compression circuit 1202 and the timing generation circuit 1203. Data string 1 from time T1 to time T5
301 is 1/1 of the data string 1302 from time T3 to time T5.
Compressed for two periods.

The intermittent data 1302 is stored in the inverter 12
04 and the delay circuit 1205 invert the data, delay by the time τ to form a data string 1303.
The data sequence 1302 and the data sequence 1303 are switched by the changeover switch 1206 according to the timing waveform 1305 at times T1, T3, T5, and T6, and the data sequence 1304 is obtained. The data sequence 1302 and the data sequence 1303 are intermittent data, and since the data sequences of the respective data sequences do not coincide with each other, the data sequence 1304 can be obtained by digital addition. Therefore, in this case, the changeover switch 1206 can also be configured by a digital addition circuit (OR circuit).

FIG. 14 is a simulation diagram showing a transmission pattern. This figure shows how the multiplexed signal appears on the television screen when the timing waveform 1305 in FIG. 13 is synchronized with the horizontal synchronizing signal of the television signal. The data from a1 to a5 in the first horizontal scanning period, the data from a1 bar to a5 bar in the second horizontal scanning period, the data from b1 to b5 in the third horizontal scanning period, the fourth During the horizontal scanning period, data from b1 bar to b5 bar is transmitted. As a result, data in two horizontal scanning periods such as the first and second horizontal scanning periods and the third and fourth horizontal periods have the same phase.

Here, we consider the video color subcarrier of the current television broadcast. FIG. 15 is a vector diagram showing a change of a video carrier by a color subcarrier. FIG.
FIG. 15A is a video carrier vector diagram when there is no orthogonal component multiplexing, and FIG. 15B is a video carrier vector diagram when there is orthogonal component multiplexing. ωs is the angular frequency of the chrominance subcarrier, and its phase changes from 1 to m, n, o... s.
Further, ωs ′ is the angular frequency of the color subcarrier in the adjacent horizontal scanning period, and is shifted in phase from ωs by π to l ′, m ′, n
, ... s'. A and -A indicate multiplexing on orthogonal components. In the current television broadcast, the phases of the color subcarriers in adjacent horizontal scanning periods are shifted by π due to the relationship between the frequency of the color subcarrier and the horizontal scanning frequency. In the orthogonal multiplexing, the vector diagram in the case where the signal of A is multiplexed in a certain horizontal scanning period having a subcarrier having a phase of ωs and the signal of −A is multiplexed in the horizontal scanning period of adjacent ωs ′ is shown in FIG. b). As shown in FIG. 4, orthogonal multiplex components cause phase fluctuation of the video carrier, and when the detection method of the television video signal is the envelope detection method or the pseudo-synchronous detection method, the video detection output signal includes the video carrier wave. The signal corresponding to the envelope of is obtained. In this case, the phase of the color subcarrier included in the video detection output signal is
In the case shown in FIG. 5 (a), it is l or p ', but in the case shown in FIG.
It is. These phase differences are φ. The change in the phase of the color subcarrier results in a change in the hue on the reproduced video screen.

When the video detection system is synchronous detection, only the directional component of cos ωct shown in the figure is detected.
Even if A is multiplexed, the maximum amplitude phase is 1 or p ', and there is no phase change. In the case of envelope detection or the like, signals to be multiplexed according to the code to be multiplexed are A and -A
When the phase is inverted, the phase direction of the color subcarrier is reversed. A signal multiplexed in adjacent horizontal scanning periods is A
15A, the chrominance subcarriers have exactly the relationship of ωs and ωs ′ shown in FIG. 15B, and the chrominance subcarriers have the same phase change amount and opposite phase. Up and down hue changes on the television screen during adjacent horizontal scanning periods are in opposite phases. A viewer of such a screen hardly feels a change in hue due to the low visual chromaticity sensitivity frequency characteristic and the integration effect of eyes.

That is, a1 to a5 and a5 shown in FIG.
Between the 1 bar to the a5 bar or the b1 bar to the b5 bar and the horizontal scanning period such as the b1 to b5, the person hardly feels the hue change. However, there is no effect of making the hue change hard to perceive at portions where the data are not the same in opposite phases, such as a1 bar to a5 bar and b1 to b5.

Further, when data of the same opposite phase is transmitted in the adjacent horizontal scanning period, a "comb filter" of the correlation (so-called line correlation) in the horizontal scanning period is used for separating the luminance signal and the color difference signal. In such a television receiver, the phase change of the chrominance subcarrier is canceled out in a circuit, and no hue change occurs.

FIG. 16 is a block diagram of a comb filter for extracting a color difference signal for general luminance signal and color difference signal separation. 1
Reference numeral 601 denotes an input terminal, 1602 denotes a delay circuit, 1603 denotes a subtraction circuit, and 1604 denotes an output terminal. Input terminal 1601
Is subtracted by the subtraction circuit 1603 from the color difference signal delayed by one horizontal scanning period by the delay circuit 1602, and is obtained at the output terminal 1604.

FIG. 17 is a waveform diagram for explaining the operation. 17
Numeral 01 denotes a waveform when no multiplexing is performed, 1702 denotes a waveform when the color sub-conveyance of ωs is multiplexed with A, and FIG.
FIG. 15 shows a case where the color subcarrier of ωs ′ is multiplexed with −A.
A waveform 1704 on the left side is an inverted waveform of the waveform 1703. The color subcarrier waveform 1701 without multiplexing has the maximum amplitude at time l. The waveform 1702 of the chrominance subcarrier when the multiplexed signal A is orthogonally multiplexed receives a phase change of φ1, and has a maximum amplitude between time s and time l. In addition, the waveform 1703 of the color subcarrier when orthogonally multiplexed by -A in the adjacent horizontal scanning period receives a phase change of φ2, and the time p ′
And the amplitude becomes maximum between time q ′. Delay circuit 1602
Is subtracted by the subtraction circuit 1603 from the waveform 1702 delayed by one horizontal scanning period after
The inverted waveform 1704 of 703 is added to the waveform 1702, and the waveform after the addition becomes the same as the waveform 1701 when the amplitude is reduced to half. That is, the chrominance subcarrier obtained by the comb filter does not undergo a phase change due to the orthogonally multiplexed signal even if the video signal detection circuit is an envelope detection circuit. In this case, FIG.
Since there is no phase change only when the upper and lower data have the same opposite phase in the adjacent horizontal scanning periods as shown by a1 to a5 and a1 bar to a5 bar shown in FIG. There is no phase change for each scanning period.

As described above, according to this embodiment in which the spectrum suppression processing circuit 114 of FIG. 1 is constituted by the circuit shown in FIG. 12, the hue change of the video signal due to the multiplexed transmission signal is suppressed. This also has the effect of reducing the interference that occurs.

Although the transmission data is assumed to be continuous data in FIG. 12, the time axis compression circuit 1202 is used. However, when the transmission data is intermittent discontinuous data, it becomes unnecessary.

FIG. 18 is a block diagram showing another example of a television transmission signal reproducing apparatus embodying the present invention. 4 have the same functions. 417 is a spectrum suppression processing signal reproduction circuit, 1801 is a code identification circuit,
1802 is a clock recovery circuit, 1803 is a switching circuit, 1
Reference numeral 804 denotes a time base expansion circuit, and 1805 denotes a timing reproduction circuit. The waveform detected by the synchronous detection circuit 415 is converted into a digital code in the code identification circuit 1801 by the clock timing pulse reproduced by the clock reproduction circuit 1802.

Data of a necessary period in the signal returned to the digital code is selected and extracted by the switching circuit 1803 and the timing reproducing circuit 1804. After that, the original transmission data is returned to the original transmission data by the time base expansion circuit 1804.

FIG. 19 shows an example of a data sequence for explaining the operation of FIG. Reference numeral 1901 denotes a timing waveform for synchronization in the horizontal scanning period, 1902 denotes a data sequence of the transmitted and received signals shown in FIGS. 12, 13 and 14, 1903 denotes a timing waveform obtained from the timing waveform 1901, 190
Reference numeral 4 denotes an output data string of the switching circuit 1803, and reference numeral 1905 denotes an output data string of the time base expansion circuit 1804. The transmitted and received data sequence 1 which is the output of the code identification circuit 1801
Reference numeral 902 denotes a switching circuit 180 according to a timing waveform 1903 obtained from the timing waveform 1901 of the horizontal synchronization signal.
3 is repeated from time T1 to T2 in a continuous manner from conduction T2 to T3, thereby forming a data string 1904.
The intermittent data from the time T1 to the time T2 is expanded by the time base expansion circuit 1804 in the data sequence 1904, and the time T1
To T3, and this operation is repeated to form a data string 1905. As a result, the data string before the spectrum suppression processing shown in the data string 1301 in FIG. 13 was reproduced.

As described above, according to this embodiment, there is an effect that the transmission signals shown in FIGS. 12, 13 and 14 can be reproduced.

FIG. 20 is a block diagram of still another example of a television transmission signal reproducing apparatus embodying the present invention.
4 and 18 indicate the same function. 41
7 is a spectrum suppression processing signal reproducing circuit, 2001 is a subtraction circuit, and 2002 is a delay circuit. As shown in FIGS. 12, 13, and 14, the same data is multiplexed and transmitted in opposite phases during two horizontal scanning periods, so that the output signal of synchronous detection circuit 415 is delayed by one horizontal scanning period by delay circuit 2002. When the subtracted signal is subtracted by the subtracting circuit 2001, a double data amplitude is obtained. The white noise added in the transmission system is √2
It only doubles. An interference signal from the video signal due to a ghost or the like of the video signal is canceled out because the video signal has a correlation every horizontal period. The removal of the interference signal from the video signal is performed in the following process. Data X is transmitted at a certain timing in a certain horizontal scanning period, and X is transmitted at a timing after one horizontal scanning period elapses.
Assuming that X bar data is inverted, the received signal X bar and the data X delayed by one horizontal scanning period by the delay circuit 2002 are added to the subtraction circuit 2001 at the same timing and subtracted. Therefore, the output of the subtraction circuit 2001 is X- (X bar) = 2X
.. (6), and a double signal is obtained. Next, it is assumed that interference G from the video signal is added during transmission. Since the video signal has a large correlation in each horizontal scanning period (images such as vertical stripes are particularly strong), the disturbance from the video signal is the disturbance G at both the data X timing and the data X bar timing. The output of the subtraction circuit 2001 is (X + G)-(X bar + G) = 2X (7), and the disturbance G is canceled. However, the canceling effect is small at locations where the correlation between video signals in each horizontal scanning period is small.

FIG. 21 shows an example of a data sequence for explaining the operation of FIG. Reference numeral 2101 denotes a timing waveform for synchronization in the horizontal scanning period, 2102 denotes a data sequence after the transmission signals shown in FIGS. 12, 13 and 14 are received, 2103 denotes an output data sequence of the delay circuit 2002, and 2104 denotes subtraction. Circuit 200
1 is an output data string 2105 is a timing waveform 2101
2106 is the switching circuit 18
03, an output data string 2107 is a time axis expansion circuit 180
4 is an output data string.

The transmitted and received data sequence 2102 output from the synchronous detection circuit 415 is delayed by one horizontal scanning period by the delay circuit 2002 to become a data sequence 2103.
The data at time T0 is delayed to time T1. The data sequence 2102 is subtracted from the delayed data sequence 2103 by the subtraction circuit 2001 to become a data sequence 2104. In response to the timing waveform 2105 obtained from the timing waveform 2101 of the horizontal synchronizing signal, the switch 1803 is turned on from time T1 to time T2 and cut off from time T2 to time T3, so that the data sequence 2104 becomes the data sequence 21
06. Here, the data sequence 2104 and the data sequence 210
In FIG. 6, the double description of the data, such as 2a1 and a1, is omitted.

The data train 2106 expands the data from time T1 to T2 by the time axis expanding circuit 1804.
The data is from time T1 to T3, and this operation is repeated to obtain data 2107. As a result, FIG.
The data string before the spectrum suppression processing shown in the data string 1301 of No. 3 was reproduced.

According to the present embodiment, FIGS.
4 has the effect that the transmission signal shown in FIG. 4 can be reproduced, and also has the effect that interference from the video signal can be reduced. FIG. 22 is a graph showing the effect. The horizontal axis is the level of interference due to ghosts (the right is less). The vertical axis represents a bit error rate representing an error rate accompanying transmission. White noise was added during transmission during the experiment to generate errors. In this case, the ratio of carrier to noise of a signal to be multiplexed and transmitted orthogonally is indicated by CIN. 2201 is a curve connecting measured values when the subtraction circuit 2001 is not used;
Reference numeral 2202 denotes a curve connecting actual measured values when the subtraction circuit 2001 is used. The ghost level at which the same bit error rate is 10 to 4 is about 15 dB to the left of the curve 2202. The fact that the same error rate is obtained when the number of ghosts is increased by about 15 dB indicates that the effect of reducing interference from a video signal due to ghosts and the like is correspondingly obtained. When the ghost level is as small as −50 dB, the curve 2201 has a C / N = 15 dB bit error rate and a curve 2201.
That the bit error rate of C / N = 18 dB is almost the same, the signal amplitude by the subtraction circuit is doubled and the noise
The doubling indicates that the C / N is improved by 3 dB.

FIG. 23 shows the spectrum suppression processing circuit 1 of FIG.
FIG. 14 is a block diagram showing still another example of No. 14. 114 is a spectrum suppression processing circuit, 2301 is an input terminal, 230
2 is a timing generation circuit, 2303 is an inverter, 23
04 is a delay circuit, 2305 is a changeover switch, and 2306 is an output terminal.

As a specific example of the spectrum suppression processing circuit,
The data applied to the input terminal 2301 is
03, the signal is inverted and delayed by the delay circuit 2304. According to the signal generated by the timing generation circuit 2302, the input data and the inverted and delayed data are switched by the changeover switch 230.
5 and is obtained at the output terminal 2306.

FIG. 24 is an example of a transmission data sequence for explaining the operation of FIG. 2401, 2404, and 2406 are timing waveforms, 2402 is an input data sequence applied to the input terminal 2301, 2403 is an output data sequence of the delay circuit 2304, and 2405 is an output data sequence.

Data string 2 applied to input terminal 2301
402 is inverted by an inverter 2303 and a delay circuit 2304, and is delayed by a time τ to become a data string 2403.
Note that the timing waveform 2401 is inverted every time τ. (Times T1, T2, T3) Timing waveform 2404
Is inverted in the middle of the data string. The switch 2305 is connected to the side (a) in the state (a) indicated by High in the drawing (during the period from the time T2 to the time T4). L
In the state (b) indicated by ow (from time T4 to time T5
Switch 2305 is connected to the side (b). The output data sequence 2405 of the changeover switch 2305 is
Obtained at output terminal 2306.

FIG. 25 is a simulation diagram showing a transmission pattern. This figure shows how the multiplexed signal appears on the television screen when the timing waveform 2406 in FIG. 24 is synchronized with the horizontal synchronizing signal of the television signal. The horizontal scanning direction is shown horizontally and the vertical scanning direction is shown vertically. As shown by the frame of a circle in FIG. 25, in the adjacent horizontal scanning period, the upper and lower sides are inverted data every data. Since the data is inverted during the adjacent horizontal scanning period, the multiplexed signal to the quadrature component of the video carrier becomes in opposite phase, and the effect of the multiplexed transmission signal on the hue change of the video signal is reduced. is there.

As described above, according to the present embodiment in which the spectrum suppression processing circuit 114 of FIG. 1 is constituted by the circuit shown in FIG. Since the phase is reversed, there is an effect that interference caused by a change in hue of an image is reduced. Also, in all the horizontal scanning periods, as shown by the frame of a circle in FIG. 25, the phase of each data is opposite to that of the upper and lower adjacent scanning periods in a mesh pattern, so that it is affected by the hue change of the image. The interference is fine, and the effect is that the interference is reduced by the low sensitivity frequency of visual chromaticity.

The transmission signal can be reproduced by the configuration shown in FIG. Since the operation timing is partially different, a data sequence for explaining the operation is shown in FIG. Reference numeral 2601 denotes a timing waveform for synchronization in the horizontal scanning period, 2602 denotes a data sequence after the transmission signals shown in FIGS. 23, 24, and 25 are received, 2603 denotes an output data sequence of the delay circuit 2002, and 2604 denotes a data sequence. Output data string of subtraction circuit 2001, 26
05 is a timing waveform, 2606 is a changeover switch 180
3 is an output data sequence, and 2607 is a time axis expansion circuit 1804.
Is an output data string. The transmitted and received data sequence 2602 output from the synchronous detection circuit 415 is
02, the data string 260 is delayed by one horizontal scanning period.
It becomes 3. If the data at time T1 is at time T2 or at time T
Data 2 is delayed at time T3. The delayed data string 2603 is added to the data string 26 by the subtraction circuit 2001.
02 is subtracted to form a data string 2104. The data sequence 2604 is switched from the time T2 to the time T4 according to the timing waveform 2605 and the conduction is repeated from the time T4 to the time T5.
04 becomes a data string 2606. Here, the data sequence 260
4 and the data string 2606, the double description of the data is omitted, such as 2f1 and f1. Time axis expansion circuit 1804
The data from time T4 to time T5 is
The data sequence 2606 is time-decompressed into a data sequence 2607 so as to be decompressed to data from to T6. As a result, the data string before the spectrum suppression processing shown in the data string 2402 in FIG. 24 was reproduced.

In FIG. 14 and FIG. 25 described above, signals multiplexed and transmitted on the screen of the television are depicted as simulated. In these cases, it is described that the signal to be multiplexed and transmitted has a fixed number of data during the horizontal scanning period and is a synchronized signal. Is obtained. Also, by arbitrarily setting the last data period of the horizontal scanning period or increasing or decreasing the number of data in a certain pair of horizontal scanning periods, the transmission speed of the multiplexed signal does not match the horizontal scanning period of the television. Even in such a case, signal transmission is possible.

FIG. 27 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention. 1 have the same functions. 2701
Denotes a control circuit, 2702 denotes a control signal generation circuit, and 2703 denotes a changeover switch. In this embodiment, the number of increase / decrease in the number of data in the pair of horizontal scan periods described above, a signal indicating the increase / decrease horizontal scan period, a signal indicating whether the top data of the horizontal scan period has the same phase as the upper or lower horizontal scan period data, or A signal indicating the positional relationship with the vertical scanning period and the like are added to the signal to be multiplex-transmitted. An output signal of the digital signal processing circuit 113 and an output signal of the control signal generation circuit 2702 are multiplexed in a time division manner by a changeover switch 2703 controlled by the control circuit 2702. Since the output of the digital signal processing circuit 113 is not transmitted only during a period in which a signal such as a control signal is multiplexed, the data is time-axis-compressed by the digital signal processing circuit 113.

FIG. 28 shows an example of a signal multiplexed by time division in FIG. 2801 is a vertical synchronization signal, 2802
Is a horizontal synchronization signal, 2803 is a multiplexed signal example, 280
4 is an example of a control signal. Television horizontal sync signal 2
In two horizontal scanning periods following the vertical synchronization equalization pulse period 802, the control signals indicated by C and C bar change from time T1 to time T2.
Is transmitted in the normal phase, and is transmitted in the opposite phase during the next horizontal scanning period from time T2 to T3. The control signal is a control signal example 280
As shown by reference numeral 4, it is composed of a 16-bit synchronization signal, a 32-bit control signal, a 48-bit data number information or other information. 2 in one vertical scanning period
The horizontal scanning period is used for transmitting the control signal.
Since four horizontal scanning periods are used for transmitting this control signal within two vertical scanning periods which are 525 horizontal scanning periods of the current NTSC television, the digital signal processing circuit 1
The time axis compression ratio performed in step 13 is 525/521 or more.

According to the present embodiment shown in FIGS. 27 and 28, the polarity of the signal multiplexed and transmitted orthogonal to the horizontal scanning period adjacent to the horizontal scanning period, and the transmission of the signal multiplexed and transmitted during the horizontal scanning period Since the control signal such as the increase / decrease of the number of data and the increased / decreased horizontal scanning period number and the like are transmitted in a time division manner, there is an effect that the receiver for receiving and reproducing the orthogonally multiplexed transmitted signal operates stably.

FIG. 29 is a transmission pattern diagram schematically showing an example of interleave processing of the digital signal processing circuit 113 in correspondence with the television screen of FIG. The 0th, 1st, and 2nd sampled data of the left channel of the audio signal are denoted by L0, L1, and L2. The 0th, 1st, and 2nd sampled data of the right channel are R0, R
1, denoted by R2. L0 bar, L1 bar, L2 bar, R
0 bar, R1 bar, R2 bar are L0, L1, L2, R0, R
1, Inverted data of R2. As shown in the drawing, the first sampled data L1 is characterized in that the preceding L0 and the subsequent L2 are interleaved such that they are separated from each other by the same horizontal scanning period or more.

According to the present embodiment, since adjacent sampled data before and after is transmitted without interleave processing during the same horizontal scanning period, adjacent horizontal data is reproduced in the reproduction of the audio signal by the receiver shown in FIG. In an image (such as a horizon) in which the upper and lower phases are small in the scanning period, the effect of reducing the interference from the image signal to the orthogonally multiplexed signal by the subtraction circuit 2001 is small, and an error easily occurs in certain sampled data. By interpolating the erroneous sampled data from the preceding and succeeding sampled data, an effect of preventing abnormal sound from being generated is obtained. In the above description, one sampled data is represented by one bit. However, the same effect can be obtained with N-bit data by similarly separating the preceding and succeeding sampled data from the same horizontal scanning period.

FIG. 30 is a block diagram showing still another example of a television transmission signal reproducing apparatus embodying the present invention.
30 is an example of a receiver that receives and reproduces the transmission signals shown in FIGS. 27, 28, and 29. Those having the same reference numerals as those in FIG. 20 indicate the same functions. Reference numeral 3001 denotes a control signal reproduction circuit;
Reference numeral 2 denotes an interpolation control circuit.

The control signal is extracted from the output signal of the code identification circuit 1802 by the control signal reproducing circuit 3001. In response to the control signal, a timing recovery circuit 1805
The switch 1803 is switched by the signal from the control unit 1801, and a necessary signal among the output signals of the code identification circuit 1802 is obtained. The output signal of the changeover switch 1803 is expanded by the time base expansion circuit 1804. The decompressed signal is detected and corrected by a digital signal processing circuit 418 during transmission, and deinterleaved to return to the original transmission data. In the error detection and correction, it is not possible to correct errors intensively generated in a portion of the video signal having a small correlation with an increase in interference from the video signal. The sampling data having an error that cannot be corrected is interpolated from adjacent sampling data in the digital signal processing circuit 418 controlled by the interpolation control circuit 3002.

According to the present embodiment, since the reproduction is performed by the control signal and the interpolation from the adjacent sampling data is performed, the stable reproduction can be performed. FIG. 31 is another example of a block diagram of a transmission signal generation device for television signal transmission embodying the present invention. 1 have the same functions.
114 is a spectrum suppression processing circuit, 3101 is a subtraction circuit, and 3102 is a delay circuit.

The signal applied to input terminal 111 is subtracted by subtraction circuit 3101 from the signal delayed by delay time τ in delay circuit 3102. For example, if the input signal is 20K
If the signal is Hz or less, the signal can be transmitted by setting the delay time τ to about 10 μsec. Since the difference between the input signal and the delayed input signal is obtained by the subtraction circuit 3101, the difference decreases as the frequency of the input signal decreases. As a result, a signal having a low frequency component is obtained at the output of the spectrum suppression circuit 114. Since the carrier is modulated by the signal in the modulation circuit 117, the components near the carrier frequency of the modulated signal are suppressed and transmitted.

According to the present embodiment, the components of the multiplexed signal in the vicinity of the carrier frequency are suppressed, so that the effect of the multiplexed signal on the current terrestrial television broadcast can be reduced.

FIG. 32 is a block diagram showing another example of a television transmission signal reproducing apparatus embodying the present invention. 4 have the same functions. 417 is a spectrum suppression processing signal reproduction circuit, 3201 is an addition circuit, 32
02 is a delay circuit.

The output signal of the synchronous detection circuit 415 is converted by the spectrum suppression processing signal reproduction circuit 417 into the spectrum suppression processing circuit 11 of the transmission signal generator shown in FIG.
4 and is obtained at the output terminal 420. In the spectrum suppression processing signal reproduction circuit 417, the addition circuit 320
The reproduction signal, which is the output signal of No. 1, is delayed by the delay time τ by the delay circuit 3202, and then added by the addition circuit 3201 to the output signal of the synchronous detection circuit 415. Synchronous detection circuit 415
Then, a signal having a difference from the signal transmitted before the time τ transmitted by the transmission signal generation device is output. The signal is added back to the signal before time τ to return to the original signal. Thus, according to this embodiment, the transmission signal shown in FIG. 32 can be reproduced.

FIG. 33 is another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention. 1 have the same functions. 114 is a spectrum suppression circuit, and 3301 is a pre-emphasis circuit.

The signal applied to the input terminal 111 is applied to the pre-emphasis circuit 3301 to reduce the low frequency components of the signal. Since the carrier wave output from the phase shift circuit 116 is modulated by the signal of which the low frequency component has been reduced by the modulation circuit 117, the component near the carrier frequency of the modulated signal is suppressed and transmitted.

According to the present embodiment, since components near the carrier frequency of a multiplex-transmitted signal are suppressed, there is an effect that interference with the current terrestrial television broadcast given by the multiplex-transmitted signal can be reduced.

FIG. 34 is a block diagram showing another example of a television transmission signal reproducing apparatus embodying the present invention. 4 have the same functions. 417 is a spectrum suppression processing signal reproduction circuit, and 3401 is a de-emphasis circuit.

The output signal of the synchronous detection circuit 415 is supplied to the de-emphasis circuit 340 in the spectrum suppression processing circuit 417.
1, the inverse processing of the pre-emphasis circuit 3301 of the transmission signal generator shown in FIG. 33 is performed, and the signal is returned to the original signal. As described above, according to the present embodiment, FIG.
Can be reproduced.

FIG. 35 is a block diagram showing another example of a television transmission signal reproducing apparatus embodying the present invention. 4 have the same functions. 3501 is a bandpass filter, and 3502 is a frequency conversion circuit.

The difference from FIG. 4 is that the bandpass filter 3501
The frequency of the intermediate frequency signal detected from the signal multiplexed and transmitted orthogonally by the frequency conversion circuit 3502 is set lower than the frequency of the intermediate frequency signal detected from the video signal. The intermediate frequency signal output from the frequency conversion circuit 403 is detected, and the video signal is demodulated. The frequency of the intermediate frequency signal is often 58.75 MHz in the case of a receiver for Japanese terrestrial television broadcasting. The output signal of the frequency conversion circuit 430 is further converted by the frequency conversion circuit 3502 and then detected by the synchronous detection circuit 415. The frequency of the intermediate frequency signal detected by the frequency conversion circuit 3502 to, for example, about 5 MHz is reduced.

According to the present embodiment, since the frequency of the signal used in the synchronous detection circuit 415 is low, the carrier recovery circuit 4
The phase error caused by the delay or the like caused by passing through the circuit of the carrier reproduced in step 16 can be reduced, and there is an effect that the signal multiplexed and transmitted orthogonally and stably is reproduced.

FIG. 36 is a block diagram showing another example of a television transmission signal reproducing apparatus embodying the present invention. 4 and 35 indicate the same function.
Reference numeral 3502 denotes a frequency conversion circuit, 3601 denotes a reference signal generation circuit, 3602 denotes a mixing circuit, 3603 denotes a voltage-controlled local oscillation circuit, and 3604 denotes a low-pass filter.
Is constituted by a mixing circuit 3602 and a voltage controlled local oscillation circuit 3603.

Detection is performed by utilizing that the modulation by the multiplexed signal is orthogonal to the modulation by the video signal. The phase difference between the output signal of the band-pass filter 414 and the output signal of the reference signal generation circuit 3601 is determined by the synchronous detection circuit 41.
5 is detected by the low-pass filter 3604 and fed back to the voltage-controlled local oscillation circuit 3603. Thereby, the carrier of the intermediate frequency signal output from the band-pass filter 414 and the output signal of the reference signal generation circuit 3601 are synchronized, and a signal multiplexed and transmitted orthogonal to the output of the synchronous detection circuit 415 is obtained.

According to the present embodiment, the reference signal generating circuit 36
Since negative feedback is performed so that the frequency of the intermediate frequency signal detected to the signal frequency of 01 coincides with the frequency of the intermediate frequency signal, the tuning frequency shift of the band-pass filter 414 due to the frequency drift of the local oscillation signal of the frequency conversion circuit 403 is small, and the like. There is an effect that signals multiplexed and transmitted orthogonally are demodulated stably.

FIG. 37 shows the spectrum suppression processing circuit 1 of FIG.
14 shows yet another example.

FIGS. 23 to 25 show a case where odd data is shown as an example of seven data in one horizontal scanning period as a transmission data string. However, in the case of even data, FIG.
As shown in FIG. FIG. 38 is a diagram for explaining an operation such as an example of a transmission data string, and FIG. 39 is an example of a simulation pattern of transmission data of the present invention. 114 is a spectrum suppression processing circuit, 230
2 is a timing generation circuit, 3701 is a timing input terminal, 3702 is a timing generator, and 3703 is an exclusive OR (hereinafter abbreviated as EOR) 3801, 380.
4, 3807 is a timing waveform in the timing generation circuit 2302, 3802 is an input data string of the input terminal 2301, 3803 is an output data string of the delay circuit 2304, 38
05 is a timing waveform of the output of the timing generation circuit 2302, 3806 is one embodiment of the transmission data string of the present invention, 37
03 is EOR. The same reference numerals as those in FIG. 23 denote the same functions.

The difference from FIG. 23 is that the timing generation circuit 230
2 is provided with an exclusive OR 3703, and the timing generation circuit 2 is provided by timing waveforms 3801 and 3804.
A timing waveform 3805 is obtained from the output of the reference numeral 302 to control the changeover switch 2305. EOR3703
Is for inverting the control timing of the changeover switch 2305 every horizontal scanning period, so that a transmission data string 3806 is obtained, which becomes a pattern on the television screen of the transmission data simulated in FIG.

Also in the above-described embodiment, similarly to FIGS. 23 to 25, there is an effect that it is possible to reduce the influence of the multiplexed signal on the hue change of the image.

[0097] Interference can be reduced by transmitting the same multiplexed signal twice in opposite phases, but if the transmission band of the multiplexed signal is fixed, the transmission capacity is reduced by half. Restoration can also be performed by a partial response method or the like that compresses the transmission band by actively using intersymbol interference such as a binary code. In addition,
For the partial response method, see page 137 to 1 of the Ohm Company's version of the modern digital communication method issued in September 1981.
Details are omitted since they are shown on page 42 and the like.

FIG. 40 shows a demodulation operation when receiving transmission data sequence 3806. 4001 is a timing waveform for synchronization in the horizontal scanning period, 4002 is a data string transmitted and received, 4003 is a data string output from the delay circuit 2002, 4004 is a data string output from the subtractor 2001, 40
05 is a timing waveform, 4006 is a timing waveform inverted every horizontal scanning period, and 4007 is a timing waveform 4
005 and a timing waveform obtained from the timing waveform 4006, 4008 is a data sequence holding the value of the switch 1803, 4009 is a timing waveform, and 4010 is an output data sequence of the time axis decompression circuit 1804.

The received data sequence 4002 is
02 results in a data string 4003. Data string 4003
Is subtracted from the data string 4002 by the subtractor 2001 to obtain a data string 4004. Timing waveform 40
XOR of the timing waveform 4006 and the timing waveform 4006 (the same operation as the EOR 3703 in FIG. 37), the switch 1803 is turned on above the obtained timing waveform 4007, and the value of the turning-on period is held during the cutoff period of the switch 1803. A data string 4008 is obtained. This can be configured with a digital circuit that is latched above the timing waveform 4007. By latching this data string 4008 at the falling edge of the timing waveform 4009, a data string 4010 is obtained at the output of the time base extension circuit 1804. This data string 4010 matches the original data string 3802 on the transmitting side shown in FIG. The data string 4008
In the data string 4010, double display such as 2f1 is omitted.

As can be seen from the above description, it is possible to reduce the interference by transmitting the same multiplex signal twice in opposite phase, but if the transmission band of the multiplex signal is fixed, the transmission capacity becomes 1
Since the number is reduced to / 2, it is also possible to perform restoration using a multi-level method or a partial response method in which transmission band compression is performed by positively utilizing intersymbol interference such as duobinary code.

Since the partial response method is described in pages 137 to 142 of the Ohmsha Modern Digital Communication System issued in September 1981, details thereof are omitted.

In FIGS. 14, 25 and 39,
The modulation direction of the multiplexed signal is schematically shown in correspondence with the screen of the television video signal. In these cases, the multiplex signal is
Although the description has been given of a synchronized signal in which a certain number is included in the horizontal scanning period, when the transmission speed of the multiplexed signal and the horizontal scanning period are not synchronized, the horizontal scanning period of the multiplexed signal and the horizontal scanning period of the video signal are almost equal. If they match, a similar low-frequency effect on the video signal can be obtained. Further, it is also possible to absorb the data by making the last data time of the horizontal scanning period arbitrarily, or increasing or decreasing the number of data in a pair of horizontal scanning periods.

FIG. 41 shows another embodiment of the transmission signal reproducing apparatus according to the present invention. 20 and 35 indicate the same function.

The difference from FIG. 20 is that the bandpass filter 350 is used to lower the frequency for demodulating the digitally encoded and multiplexed audio signal below the frequency for video signal demodulation.
1 and a frequency conversion circuit 3502.

According to the present embodiment, the frequency conversion circuit 403
The demodulation of the video signal is performed at the intermediate frequency of the output (58.75 MHz generally used in Japanese terrestrial television), and the intermediate frequency of the output of the frequency conversion circuit 4102 having a lower frequency (for example, about 5 MHz) The demodulation of the audio signal transmitted after being digitally encoded by the carrier demodulation is performed by the carrier recovery circuit 41 used in the synchronous detection circuit 415.
The phase error due to the circuit delay time or the like of the carrier reproduced in step 6 is reduced by lowering the frequency, so that there is an effect that the digitally encoded and transmitted audio signal can be demodulated stably.

FIG. 42 shows still another embodiment of the transmission signal reproducing apparatus of the present invention.

Elements having the same reference numerals as in FIGS. 20, 36 and 41 indicate the same functions. Frequency conversion circuit 350 shown in FIG.
2 is a mixing circuit 3602 and a voltage controlled type local oscillation circuit 36
03.

The difference from FIG. 41 is that in FIG. 41, a synchronous detection circuit 415 detects a signal reproduced in a carrier wave recovery circuit 416, orthogonally modulated with a carrier video signal, digitally encoded, and transmitted in synchronization with an audio signal transmitted. In contrast, FIG. 42 utilizes the fact that the modulation by the digitally encoded audio signal and the modulation by the video signal are orthogonal,
The phase difference between the reference signal generation circuit 3601 and the intermediate frequency signal including the carrier is determined by the synchronous detection circuit 415 and the low-pass filter 3.
By detecting the signal at 604 and feeding it back to the voltage-controlled local oscillation circuit 3603, the carrier of the intermediate frequency is synchronized with the output of the reference signal generation circuit, and the output of the synchronous detection circuit 415 is used as a detection output.

According to the present embodiment, the reference signal generating circuit 36
Since this is a negative feedback loop in which the intermediate frequency for demodulation matches the frequency of 01, the frequency shift and the demodulation frequency drift of the band-pass filter 414 due to the frequency drift of the frequency conversion circuit 403 and the like are small, and the embodiment shown in FIG. Further, there is an effect that the signal can be demodulated stably.

FIG. 43 shows still another embodiment of the transmission signal reproducing apparatus of the present invention. 42 have the same functions as those in FIG. Reference numeral 4301 denotes a sample and hold circuit. The difference from FIG. 42 is that the sample and hold circuit 430
1 enters the loop of the carrier recovery circuit, and the timing recovery circuit 1805 controls the sample and hold circuit 4301.

According to this embodiment, when the amplitude of the video signal becomes large, such as the horizontal synchronization pulse feedback of the video signal among the outputs of the synchronous detection circuit 415, the system is stabilized by setting the period to the hold state.

FIG. 45 shows still another embodiment of the transmission signal reproducing apparatus of the present invention. 44 have the same functions as those in FIG. 4501 is a synchronous detection circuit, and 4502 is a phase shift circuit. The difference from FIG.
Is output from the reference signal generation circuit 3601 using a synchronous detection circuit 4501 and a video signal which is synchronously detected by a signal obtained by shifting the phase of the output signal of the reference signal generation circuit 3601 by π / 2.

As a result, the time required for the input signal to reach the synchronous signal detection circuit 4401 and the time required for the synchronous detection circuit 415 almost match, and the time required for the delay circuit 4402 can be reduced.

According to this embodiment, since the time of the delay circuit 4402 can be reduced, the effect of enabling more stable reception is increased.

FIG. 46 shows still another embodiment of the transmission signal reproducing apparatus of the present invention. 42 have the same functions as those in FIG. Reference numeral 4601 denotes a control signal reproducing circuit. FIG.
2 is that the received control signal is
01, and controls the switch 1803 and the time base expansion circuit via the timing reproduction circuit 1805 according to the control signal.

According to this embodiment, since the timing is generated by the control signal, there is an effect that more stable reception and reproduction can be performed.

FIG. 47 is still another example of a block diagram of a transmission signal generator for television signal transmission embodying the present invention. 1 have the same functions. 11
4 is a spectrum suppression processing circuit, and 4701 is a ternary conversion circuit.

The digital code output from the digital signal processing circuit 113 is converted by a ternary conversion circuit 4701 from a binary digital signal of +1, 0 to a ternary digital signal of +1, 0, -1. A video signal carrier that is 90 degrees shifted in phase with digitally encoded audio of a low-pass fill suitable for the spectral band is modulated. In order to prevent the influence of the characteristics of the IF Nyquist filter of the receiver on the orthogonality, the signal passes through an equalizer 118 having the inverse characteristic of the IF Nyquist filter, and is added to a carrier modulated by a video signal using an adder 119. As a result, the video carrier is modulated in an orthogonal relationship between the video signal and the digitally encoded audio signal.

FIG. 48 shows an example of the ternary conversion circuit 4701. 4801 is a binary digital data input, 4802 is a delay circuit, 4803 and 4804 are inverters, 480
5,4806 is an AND circuit, 4807 is an inverter,
Reference numeral 808 denotes an addition circuit, and 4809 denotes a ternary digital data output. The operation of FIG. 48 will be described with reference to the timing chart of FIG. 49A shows a binary digital data waveform, FIG. 49B shows an output of the delay circuit 4802, and FIG.
D circuit 4805 output, (d) inverter 4807 output, (e) ternary digital data waveform (adder 480
8 outputs). (A) The binary digital data shown in the figure is delayed by a time τ by a delay circuit 4802 (b)
The timing waveform shown in FIG. Here, the time τ is a time shorter than one data length T. 2 in AND circuit 4805
The inverted of the value digital data (a) and the output of the delay circuit (b) is ANDed, and the rising edge of the binary digital data (a) is detected as shown in FIG. Similarly A
The ND circuit 4806 detects the falling edge of the binary digital data (a) by inverting the binary digital data and the AND of the delay circuit output (b), and inverts the falling edge by the invert 4807 to obtain the waveform shown in FIG. Get. The addition circuit 4808 adds the waveform shown in FIG. 9C and the waveform shown in FIG. 9D to obtain ternary digital data shown in FIG.
Comparing the figures (a) and (e), the ternary digital data is high at the rising edge of the binary digital data.
(+1), a pulse of Low (−1) is generated with a pulse width τ at the falling edge, and the remaining potential is an intermediate potential (0) between High and Low. By converting the binary digital data into the ternary digital data in this manner, the low-frequency component of the baseband digital signal can be suppressed. Modulation by the modulated audio signal modulation circuit 117 yields a signal whose spectrum near the carrier frequency is suppressed. According to the ternary conversion circuit of FIG. 48, it is possible to suppress the low-frequency component without reducing the transmission capacity when the transmission band is considered to be constant, and the circuit configuration of FIGS. There is an effect that the components can be reduced.

FIG. 50 shows another example of the ternary conversion circuit 4701. 48 denote the same function, 5001 denotes a clock input terminal, 5002 denotes an inverter, 5003
5004 is a D-flip-flop. FIG. 50 is FIG.
The delay time τ of the eight delay circuits 4802 is one data length T, and the other operations are the same as those in FIG. Therefore, only the operation different from that of FIG. 48 will be described with reference to the timing chart of FIG. In FIG. 51, (a)
Is a binary digital data waveform, (b) is a clock signal,
(C) is the output of the D-flip-flop 5003, (d) is the output of the D-flip-flop 5004, (e) is the output of the AND circuit 4805, (f) is the output of the inverter 4807,
(G) shows a ternary digital data waveform (adder circuit 4808).
Output). (A) Binary digital data shown in FIG.
(See FIG. 51 (C)).
Next, the data is further delayed by T / 2, which is half the data length, by the D-flip-flop 5004. As a result, the output of the D-flip-flop 5004 is one data longer than the binary digital data of the binary digital data input 4801. The signal is delayed by a certain T (see FIG. 51D).
The following operation is the same as in FIG. 48, and ternary digital data having a pulse width of 1 data length T is output from the ternary digital data output 4809 (see FIG. 51 (g)).

According to the ternary conversion circuit of FIG. 50, the fundamental wave component of the output ternary digital data is lower than the case where the delay time τ of the delay circuit 402 of FIG. 48 is shorter than one data length T. Since the frequency becomes the frequency, the baseband band is narrowed, and as a result, the transmission bandwidth after modulation is also narrowed.

FIG. 52 shows still another embodiment of the transmission signal generator of the present invention. 1 and 10 denote the same functions, and reference numeral 5201 denotes a multi-level modulation circuit. The operation of the digital modulation circuit 1001 is the same as that of FIG.
In the embodiment of FIG. 1, low-frequency components can be suppressed by performing digital modulation, but if the transmission capacity is fixed as compared with the embodiment of FIG. 1, the transmission band will be widened. Therefore, the multi-level modulation circuit 5201 uses a multi-level modulation scheme,
It is possible to restore the transmission capacity by compressing the transmission band using a partial response method that actively uses intersymbol interference such as duobinary code. The multi-level modulation circuit 5201 is input to the low-pass filter 115,
The following is the same operation as in FIG.

FIG. 53 shows an example of a receiver which can receive a signal transmitted from the signal transmission apparatus of FIG. 52.
The same reference numerals denote the same functions, and 5301 denotes a multi-level demodulation circuit.

The receiving circuit of FIG. 53 receives the transmission signal from FIG. 52, and the signal detected and demodulated by the synchronous detection circuit 415 is input to the multilevel demodulation circuit 5301 to demodulate digitally modulated digital data. The following is the same as the operation in FIG. The details of the partial response method are omitted since they are shown in pages 137 to P142 of "Modern Digital Communication System" published by Ohmsha in September 1981. According to the embodiment of FIG. 52, there is an effect that the low frequency component can be suppressed without reducing the transmission capacity.

FIG. 54 is an example of a circuit having the functions of a ternary identification circuit 701, a ternary / binary conversion circuit 702, and a code identification circuit 703. 7 or 8 indicate the same function, and reference numeral 5400 denotes a ternary code identification circuit having the functions of the ternary identification circuit 701 and the code identification circuit 703. 5
Reference numeral 401 denotes a sample and hold circuit (hereinafter abbreviated as an S / H circuit), and reference numeral 5402 denotes a clock signal. 54 will be described with reference to FIG.

FIG. 55 (a) shows ternary digital data,
(B) is a clock signal, (c) is the output of the S / H circuit 5401, (d) is the output of the comparator 805, (e) is the output of the comparator 806, and (f) is the binary digital data (R
S-flip-flop 809 output). The S / H circuit 5401 has a clock signal 54 as shown in FIG.
Sample at the rising edge of 02 and hold that value until the next sample. The clock signal 5402 reproduced by the clock reproduction circuit 704 is a signal having one data length T as one cycle, and the rising edge of the clock is located at a point where the code error rate is small (a so-called maximum opening of an eye pattern). The output of the S / H circuit 5401 is as shown in FIG. 55C, and identifies the ternary digital data input at the input terminal 801 as a ternary digital code synchronized with the clock signal 5402. Thereafter, the ternary digital code is input to the comparators 805 and 806, and the ternary digital code is identified as +1, 0, -1 and converted into a binary digital code (FIG. 55 (f)) in the same manner as the operation described with reference to FIG. . By using the circuit of FIG. 54, even when noise as shown by 5501 in FIG.
If it deviates from the sampling point of the H circuit 5401, it does not affect the demodulated binary digital code at all, and has the effect of not deteriorating the code error rate characteristics.

FIG. 56 shows another example of a circuit having the functions of the ternary discriminating circuit 701, ternary / binary converting circuit 702, and code discriminating circuit 703. 54 denote the same functions, 5601 denotes an S / H circuit, 5602 denotes a window comparator, 5603 and 5604 denote addition circuits, and 5605.
Is an intermediate level detection signal.

Since the basic operation of FIG. 56 is the same as that of FIG. 54, a portion that operates differently from FIG. 54 will be described with reference to FIG.

In FIG. 57, (a) is ternary digital data, (b) is a clock signal, and (c) is an S / H circuit 54.
(D) is the output of the window comparator (intermediate level detection signal 5602). Now, the signal received by the antenna 401 has been distorted due to a spatial transmission path or other causes. Therefore, the ternary digital data input to the input terminal 801 is at an intermediate level as shown in FIG. It is assumed that the pulse of the High level is higher than the pulse of the Low level. At this time, the ternary digital data becomes a signal containing a DC component,
When the DC is cut by 2 and the operating point is determined by the resistor 803, the signal is such that the intermediate level does not become 0 volt as shown in FIG. When this signal is sampled at the rising edge of the clock signal 5402 in FIG. 57B using the S / H circuit 5401 and the value is held until the next sample point, the waveform becomes as shown in FIG. The signal has a V offset. The signal in FIG. 57C is input to the comparators 805 and 806, and is also input to the window comparator 5602 and the S / H circuit 5601. The window comparator 5602 is shown in FIG.
An intermediate level portion is detected from the signal (c), and the High level is output as shown in FIG. In addition,
The intermediate level detection signal 5605 is supplied to the comparators 805, 8
06 output is possible. S / H circuit 5
601 receives the output of the window comparator 5602, samples the output of the window comparator 5602 during the High period, and holds the output during the Low period.

By operating as described above, the S / H circuit 56
01 can extract the intermediate level offset ΔV of the ternary digital data. Here, reference voltage source 807
Output V1 of the reference voltage source 808 is 0 volt (G
ND) is set as a reference, and if there is an offset of ΔV in the intermediate level of the ternary digital data, an error will occur accordingly. Therefore, the optimum reference voltage can be given to comparators 805 and 806 by adding the error component ΔV to reference voltage source 807 output V1 and reference voltage source 808 output V2 using addition circuits 5603 and 5604, respectively. As described above, according to the circuit configuration of FIG. 56, Hig for the intermediate level of the ternary digital data
The influence of the imbalance of the pulse heights of h and Low can be canceled, and ternary identification of a ternary digital signal can be performed using an optimal reference voltage. The error voltage canceling circuit shown in FIG. 56 can be used for the ternary identification circuit shown in FIG.

FIG. 58 shows another embodiment of the transmission signal reproducing apparatus according to the present invention. 7 denote the same functions, and differ from FIG. 7 in that the input of the clock recovery circuit 704 is obtained from the ternary / binary conversion circuit 702. According to this configuration, there is an effect that the clock recovery circuit 704 can be configured by a digital circuit.

FIG. 59 shows another embodiment of the transmission signal reproducing apparatus according to the present invention. 7 denote the same functions, and reference numeral 5901 denotes a video signal AGC circuit; and 5902, a digital audio AGC circuit. When the strength of the radio wave input by the antenna 401 is high or low, the ternary identification circuit 7
01 also fluctuates, and as a result, the reference voltage source 80 of the comparators 805 and 806 constituting the ternary identification circuit 701
The problem that the values of the generated voltages V1 and V2 at 7,808 are not optimal reference voltages can be considered in the embodiment of FIG. Figure 59
In FIG. 7, an AGC circuit is provided in the digital audio system of the receiver shown in FIG. The video signal AGC
The circuit 5901 is used in a conventional television receiver, and is added here for convenience of explanation. The video signal AGC circuit 5901 determines the strength of the input radio wave using the detected video signal, and controls the gains of the high frequency amplifier circuit 402 and the intermediate frequency amplifier circuit 405 accordingly. Since the strength of the radio wave of the video signal is proportional to the strength of the digitally coded audio signal modulated in the orthogonal relationship with the video signal, the digital signal AGC is also applied using the AGC control voltage of the video signal AGC circuit 5901. be able to. The AGC circuit 5902 includes a video signal AGC circuit 59
In response to the AGC control signal of 01, the gain is controlled so that the input level of the ternary identification circuit 701 is constant. According to the present embodiment, there is an effect that AGC can be applied to digital data after detection with a simple circuit configuration, and that the operation band of the AGC circuit 5902 can be a baseband band.

FIG. 60 shows another embodiment of the transmission signal reproducing apparatus according to the present invention. The same reference numerals as those in FIG. 59 indicate the same functions, and reference numeral 6001 denotes a digital audio AGC circuit.
FIG. 60 also shows an AGC circuit provided in the digital audio system of the receiver shown in FIG.
The point that the GC control voltage is used is the same as in FIG.
The point that the circuit is inserted between the BPF 414 and the synchronous detection circuit 415 is different from the example of FIG. According to the embodiment of FIG. 60, the AGC of the digital audio system can be applied with a simple circuit configuration, and the AGC circuit 6001 is controlled so that the input level of the synchronous detection circuit 415 is constant. When the optimum operation level of the synchronous detection circuit 415 is set, the synchronous detection circuit 415 can always operate in the best state. In addition, BPF414
It is also conceivable that an AGC circuit is provided in front of the circuit, or AGC is applied to both the conventional television receiver circuit and the digital audio circuit only by changing the gain of the high frequency amplifier circuit 402.

FIG. 61 shows another embodiment of the transmission signal reproducing apparatus of the present invention. This embodiment is also a digital voice AGC.
Circuit. The same reference numerals as those in FIG. 59 indicate the same functions, and reference numeral 6101 denotes an envelope detection circuit. FIG. 61 generates an AGC control signal of the AGC circuit 5902 using the envelope detection circuit 6101. The operation of the envelope detection circuit 6101 will be described with reference to FIGS. The envelope detection circuit of FIG.
It is configured using a part. The same reference numerals as those in FIG. 54 denote the same functions, 6201 denotes an S / H circuit, 6202 denotes an AGC control signal, and 6203 denotes an AGC control signal output terminal. The operation as the ternary identification circuit in FIG. 62 is the same as that in FIG. 54, and the envelope detection operation will be described with reference to FIG. 63, (a) is ternary digital data input from the input terminal 801, (b) is a clock signal 5402 (c) is the output of the S / H circuit 5401, (d) is the output of the comparator 805, and (e) is S. / H circuit 6201
The output is the AGC control signal 6202. Now, assuming that the ternary digital data has positive and negative pulse heights varying according to the strength of the radio wave input to the antenna 401 as shown in FIG. 63 (a), the ternary digital data is sampled and held by the clock signal 5402. FIG. 6 also shows three-valued digital data.
The pulse height changes as shown in FIG. Comparator 8
The section of the pulse High extracted in FIG.
(D), and S / H during the High period of this signal
The circuit 6201 performs a hold operation during the period of the sample operation Low. As a result, the S / H circuit 6201 sets the high-level envelope of the ternary digital data as shown in FIG.
And this signal can be used as the AGC control signal. Similarly, if the output of the comparator 806 is used for the control signal of the S / H circuit 6201, the low-level envelope of the ternary digital data can be detected.
The value obtained by taking R is input to the S / H circuit 6201,
If the output of the H circuit 6201 is full-wave rectified, both the High level and the Low level of the ternary digital data can be used for the AGC control signal.

According to the embodiment shown in FIG. 61, AGC is performed while observing the output of the digital audio system, so that the digital audio system can operate in an optimum state.
There is an effect that the operation band of the GC circuit 5902 can be set to the band of the baseband digital data.

FIG. 64 shows another embodiment of the transmission signal reproducing apparatus according to the present invention. Those having the same reference numerals as those in FIG. 60 or FIG. 61 indicate the same functions. In this embodiment, the digital audio system A
Regarding the GC circuit, the point that an envelope detection circuit 6101 is used is the same as in FIG. 61, but the insertion point of the AGC circuit is between the BPF 414 and the synchronous detection circuit 415 in FIG.
This is different from the example of FIG. According to the embodiment shown in FIG. 64, the AGC is performed while observing the output of the digital audio system, so that the operation can be performed in an optimal state for the digital audio system, and the AGC is performed so that the input level of the synchronous detection circuit 415 becomes constant. Since the circuit 6001 is controlled, there is an effect that the synchronous detection circuit 415 can always operate in the best state as in FIG.

The examples of the AGC circuit shown in FIGS. 59, 60, 61, and 64 have been described with reference to the embodiment shown in FIG. 7. However, the embodiments shown in FIGS. It can also be used. Also, comparators 805 and 805 constituting ternary identification circuit 701 using the AGC control signal
By controlling the reference voltage at 806, the slice level can be optimized according to the strength of the input level. FIG. 65 shows the embodiment, and the same reference numerals as those in FIG. 8 indicate the same functions.
6501 is a data slice level signal input terminal,
This is the same as the AGC control signal. Reference numerals 6502 and 6503 denote reference voltage control circuits.
01 is adjusted so that the reference voltages of the comparators 805 and 806 become optimal. According to the example of FIG. 65, there is an effect that ternary identification can be performed at an optimum slice level.

FIG. 66 shows another embodiment of the transmission signal reproducing apparatus according to the present invention. 7 or 35 indicate the same function. The difference from FIG. 7 is that a filter 3501 and a frequency conversion circuit 3502 are provided in order to lower the demodulation frequency of the digitally encoded and multiplexed audio signal below the frequency for video signal demodulation.

According to the present embodiment, the frequency conversion circuit 403
The demodulation of the video signal is performed at the output intermediate frequency (58.75 MHz is generally used in Japanese terrestrial broadcasting television), and the lower frequency intermediate frequency (for example, about 5 MHz) of the output of the frequency conversion circuit 3502 is output. The demodulation of the audio signal transmitted after being digitally encoded by the carrier demodulation is performed by the carrier recovery circuit 41 used in the synchronous detection circuit 415.
The phase error due to the circuit delay time or the like of the carrier reproduced in step 6 is reduced by lowering the frequency, so that there is an effect that the digitally encoded and transmitted audio signal can be demodulated stably.

FIG. 67 shows still another embodiment of the transmission signal reproducing apparatus of the present invention. The received signals are the same as those in FIG. 7, and those having the same reference numerals as those in FIG. 7 or FIG. 36 indicate the same functions. The frequency conversion circuit 3502 of FIG.
602 and a local oscillation circuit 3603 of a voltage control type.

FIG. 66 is different from FIG. 66 in that a signal transmitted orthogonally to a video signal by using a carrier reproduced by a carrier reproduction circuit 416 is detected by a synchronous detection circuit 415 in FIG. 67 includes a reference signal generator 3601 and a carrier by utilizing the fact that the modulation by the digitally encoded audio signal and the modulation by the video signal are orthogonal, and that the DC component of the modulation by the digitally encoded audio signal is small. Synchronous detection circuit 4 detects the phase difference from the intermediate frequency signal
15 and detected by the low-pass filter 3604 and fed back to the voltage-controlled local oscillator 3602, thereby synchronizing the carrier of the intermediate frequency with the output of the reference signal generator and making the output of the synchronous detection circuit 415 a detection output. It is in.

According to this embodiment, the reference signal generator 360
Since this is a negative feedback loop in which the intermediate frequency for demodulation matches the frequency of 1, the frequency shift and demodulation frequency drift of the bandpass filter 414 due to frequency drift and the like of the frequency conversion circuit 403 and the like are small, and the embodiment shown in FIG. This has the effect of stably demodulating.

The examples of FIGS. 66 and 67 have been described with reference to the embodiment of FIG. 7, but can also be used for other embodiments of the receiving apparatus.

FIG. 68 shows another example of a circuit having the functions of a ternary identification circuit 701, a ternary / binary conversion circuit 702, and a code identification circuit 703. 7 or 8 indicate the same function, and reference numerals 6801 and 6802 denote latches.
68 will be described with reference to FIG. In FIG. 69,
(A) is ternary digital data, (b) is the output of the comparator 805, (c) is the output of the comparator 806, (d)
Is the clock signal, (e) is the output of the latch 6801, (f)
Represents the output of the latch 6802, and (g) represents the binary digital data (the output of the RS-flip-flop 809). The operation until the outputs of the comparators 805 and 806 are obtained is the same as in FIG. The outputs of the comparators 805 and 806 are supplied to the clock recovery circuit 704 by latches 6801 and 6802.
The digital signal is latched at the timing shown in FIG. 69 (d) using the clock signal reproduced in step (1), and becomes a digital signal synchronized with the clock signal. Thereafter, the outputs of the latches 6801 and 6802 are input to the RS-flip-flop 809 as shown in FIG. 68, and the signals identified as digital signals by the same operation as in FIG.
Demodulate the value digital code. According to FIG. 68, the circuit configuration is simple and the ternary digital data is
Even when unnecessary noise such as 6901 and 6902 in (a) is mixed, if it is not at the rising edge of the clock signal, it does not affect the demodulated binary digital code at all and does not degrade the code error rate characteristics. effective.

FIG. 70 shows another example of the ternary identification circuit 701 and ternary / binary conversion circuit 702. FIG. 7 or FIG.
The same reference numerals denote the same functions.
2 is a gate and 7003 is a gate control circuit. Figure 71
70 is a timing chart for explaining FIG. 70, (a) is ternary digital data, (b) is the output of the comparator 805, (c) is the output of the comparator 806, (d) is the gate control signal, and (e) is The gate 7001 output, (f) is the gate 7002 output, and (g) is the binary digital data (RS
-Flip-flop 809 output). The operation until the outputs of the comparators 805 and 806 are obtained is the same as in FIG. The outputs of the comparators 805 and 806 are gate 7
001 and 7002 are input and gated. The gate signal is generated by the gate control circuit 7003 using the clock obtained from the clock recovery circuit 704, and as shown in FIG.
It is assumed that the rising edge of the 06 output normal data is captured. As a result, the outputs of the comparators 805 and 806 are output by the gates 7001 and 7002, respectively, as shown in FIG.
The signal is gated as shown in (f) and sent to the RS flip-flop 809. Thereafter, the same operation as in FIG.
Demodulates digital data. According to the example of FIG. 70,
The ternary digital data is added to 7101 and 71 in FIG.
Even when an unnecessary noise such as 02 is mixed, it does not affect the demodulated binary digital code at all unless it is during the gate ON of the gate signal, and has the effect of not deteriorating the code error rate characteristics. Note that the gate control circuit 7003
In the above, a plurality of gate signals having different gate pulse intervals are provided, and by monitoring the bit error rate and the like, it is possible to determine which pulse interval is to be selected and to optimize the bit error rate. it can. Further, the gate pulse timing can be changed by monitoring the bit error rate or the like, so that the bit error rate can be optimized.

FIG. 72 shows another example of the ternary identification circuit 701 and ternary / binary conversion circuit 702. 7 or 8 indicate the same function, and 7201 and 7202
Denotes a memory circuit, 7203 denotes a memory control circuit, and 7204 denotes a digital signal processing circuit. The example of FIG. 72 is also an example having a noise removing function when unnecessary noise is mixed into the ternary digital data, similarly to FIG. A clock signal of n times the data transmission period is reproduced from the clock reproducing circuit 704, and using this, the memory control circuit 7203 divides the outputs of the comparators 805 and 806 for each clock to obtain digital data, and the memory circuits 7201 and 72
02 is stored. Thereafter, the digital signal processing circuit 72
At step 04, a set pulse indicating a high portion or a reset pulse indicating a low portion of the ternary digital data close to the normal data sample point is selected, and unnecessary noise mixed into the ternary digital data is removed. After that, the data is converted into binary digital data by the RS flip-flop 809. According to the embodiment of FIG. 72, various digital processes are performed, and there is an effect that unnecessary noise mixed in the ternary digital data can be removed. The optimum data sample point can be selected by adjusting the normal data sample point with reference to the code error rate and the like. Further, the comparator 80
5,806 outputs are input to the memory control circuit 7203 and the digital signal processing circuit 7204, and when a plurality of set pulses appear before the reset pulse comes, and when a plurality of reset pulses appear before the set pulse comes. Sometimes you can choose a legitimate data sample point.

[0147]

According to the present invention, it is possible to demodulate and reproduce a transmission signal transmitted while making a large-capacity digital signal compatible with a current television broadcast while reducing interference between the two signals.

[Brief description of the drawings]

FIG. 1 is a block diagram illustrating an example of a transmission signal generation device according to the present invention;

FIG. 2 is a spectrum diagram of a transmission signal according to the present invention;

FIG. 3 is a vector diagram of a transmission signal according to the present invention;

FIG. 4 is a block diagram showing an example of a transmission signal reproducing apparatus according to the present invention;

FIG. 5 is a block diagram showing another example of the transmission signal generator of the present invention;

FIG. 6 is a waveform chart for explaining the operation of the device of FIG. 5;

FIG. 7 is a block diagram showing another example of the transmission signal reproducing apparatus of the present invention;

FIG. 8 is a block diagram showing an example of a ternary identification circuit shown in FIG. 7;

FIG. 9 is a waveform diagram illustrating the operation of FIGS. 7 and 8;

FIG. 10 is a block diagram of still another example of the transmission signal generator according to the present invention;

FIG. 11 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 12 is a block diagram of another example of the spectrum suppression processing circuit of the transmission signal generation device according to the present invention;

FIG. 13 is a schematic diagram showing a data string for explaining the operation in FIG. 12;

FIG. 14 is a schematic diagram of an example of the transmission data pattern of the present invention shown in FIGS. 12 and 13;

FIG. 15 is a vector diagram of an image color subcarrier for explanation of the present invention;

FIG. 16 is a block diagram showing a general configuration of a comb filter.

FIG. 17 is a waveform chart for explaining the operation;

FIG. 18 is a block diagram showing still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 19 is a schematic diagram showing a data sequence for explaining the operation in FIG. 18;

FIG. 20 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 21 is a schematic diagram showing a data sequence for explaining the operation in FIG. 20;

FIG. 22 is a characteristic diagram showing characteristics of the device of FIG. 20;

FIG. 23 is still another block diagram of the spectrum suppression processing circuit of the transmission signal generation device according to the present invention;

FIG. 24 is a schematic diagram showing a data sequence for explaining the operation of FIG. 23;

FIG. 25 is a schematic diagram of an example of the transmission data pattern of the present invention shown in FIGS. 23 and 24;

FIG. 26 is a schematic diagram of a data sequence for explaining an operation;

FIG. 27 is a block diagram of still another example of the transmission signal generation device according to the present invention;

FIG. 28 is a waveform chart showing an example signal for explanation in FIG. 27;

FIG. 29 is a schematic diagram of an example of an interleaving process of the transmission signal generation device according to the present invention;

FIG. 30 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 31 is a block diagram of still another example of the transmission signal generation device according to the present invention;

FIG. 32 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 33 is a block diagram of still another example of the transmission signal generation device according to the present invention;

FIG. 34 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention.

FIG. 35 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 36 is a block diagram showing still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 37 is a block diagram of still another example of the spectrum suppression processing circuit of the transmission signal generation device according to the present invention;

FIG. 38 is a schematic diagram of a data sequence for explaining the operation in FIG. 37;

FIG. 39 is a schematic diagram of an example of the transmission data pattern of the present invention shown in FIGS. 37 and 38;

FIG. 40 is a schematic diagram of a data sequence for describing an operation;

FIG. 41 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 42 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 43 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 44 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention.

FIG. 45 is a block diagram showing still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 46 is a block diagram showing still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 47 is a block diagram of still another example of the transmission signal generation device of the present invention.

FIG. 48 is a block diagram of an example of a ternary conversion circuit of the transmission signal generation device according to the present invention;

FIG. 49 is a waveform chart for explaining the operation of the circuit in FIG. 48;

FIG. 50 is a block diagram of another example of the ternary conversion circuit of the transmission signal generation device according to the present invention;

FIG. 51 is a waveform chart for explaining the circuit operation of FIG. 50;

FIG. 52 is a block diagram of still another example of the transmission signal generation device according to the present invention;

FIG. 53 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 54 is a block diagram of another example of the ternary identification circuit of the transmission signal reproducing apparatus according to the present invention;

FIG. 55 is a waveform chart for explaining the circuit operation of FIG. 54;

FIG. 56 is a block diagram of still another example of the ternary identification circuit and the like of the transmission signal reproducing device according to the present invention;

FIG. 57 is a waveform chart for explaining the operation of FIG. 56;

FIG. 58 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 59 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 60 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 61 is a block diagram showing still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 62 is a block diagram of an example of an envelope detection circuit of the transmission signal reproducing device according to the present invention;

FIG. 63 is a waveform chart for explaining the operation;

FIG. 64 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 65 is a block diagram of still another example of a ternary identification circuit and the like of the transmission signal reproducing device according to the present invention;

FIG. 66 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 67 is a block diagram of still another example of the transmission signal reproducing apparatus according to the present invention;

FIG. 68 is a block diagram of still another example of the ternary identification circuit and the like of the transmission signal reproducing device according to the present invention;

69 is a waveform chart for explaining the operation of the circuit in FIG. 68;

FIG. 70 is a block diagram of still another example of a ternary identification circuit and the like of the transmission signal reproducing device according to the present invention;

FIG. 71 is a waveform chart for explaining the operation of FIG. 70;

FIG. 72 is a block diagram showing still another example of the ternary identification circuit and the like of the transmission signal reproducing apparatus according to the present invention.

[Explanation of symbols]

 110: video signal carrier generation circuit, 114, 114: spectrum suppression processing circuit, 116: phase shift circuit, 117: modulation circuit, 118: equalizer, 119: addition circuit, 120: transmission VSB filter, 201: transmission signal of video signal 205: spectrum of a transmission signal of a multiplex signal; 301: vector of a carrier of a video signal; 302: vector of a modulated wave of a signal to be multiplexed; 414: a band-pass filter; 415: a synchronous detection circuit; Circuits, 417, 417: spectrum suppression processing signal reproduction circuit, 501: delay circuit, 502: subtraction circuit, 701, 701: ternary identification circuit, 702, 702: ternary binary conversion circuit, 703: code identification circuit, 704 ... Clock regeneration circuit 1001 ... Digital modulation circuit 1101 ... Code identification circuit 1102 clock recovery circuit 1103 digital demodulation circuit 1202 time axis compression circuit 1204 inverter 1205 delay circuit 1206 changeover switch 1801 code identification circuit 1802 clock recovery circuit 1803 changeover Circuit, 1804: time base expansion circuit, 1805, timing recovery circuit, 2001, subtraction circuit, 2002, delay circuit, 2303, inverter, 2304, delay circuit, 2305, changeover switch, 2702, control signal generation circuit, 3001, control signal Reproduction circuit 3101 Subtraction circuit 3102 Delay circuit 3201 Addition circuit 3202 Delay circuit 3301 Preemphasis circuit 3401 Deemphasis circuit 3502 3502 Frequency conversion circuit 3601 Reference signal generation time , 3602 ... mixing circuit, 3603 ... voltage control type local oscillator, 3604 ... low-pass filter.

 ──────────────────────────────────────────────────続 き Continuing on the front page (72) Inventor Akihide Okuda 292, Yoshida-cho, Totsuka-ku, Yokohama-shi, Kanagawa Prefecture Inside the Home Appliance Research Laboratory, Hitachi, Ltd. (72) Inventor Noritaka Hotta 292, Yoshida-cho, Totsuka-ku, Yokohama-shi, Kanagawa Hitachi Image Information System Co., Ltd.

Claims (13)

[Claims] A first modulated wave obtained by amplitude-modulating a first carrier with a video signal and a second carrier having a substantially quadrature phase relationship with the first carrier are separated by one horizontal line of the video signal. A television receiver for receiving a vestigial sideband transmission signal in which a second modulated wave whose amplitude is modulated by a digital signal whose polarity is inverted every period corresponding to a scanning cycle is multiplexed, A first reproducing a video signal included in the received transmission signal;
And a second reproduction system for reproducing a digital signal included in the transmission signal, wherein the second reproduction system synchronously detects the transmission signal and demodulates the digital signal. And a reproduction means for performing reproduction processing for reproducing the digital signal output from the demodulation means into a signal before the polarity is inverted. 2. A first modulated wave obtained by amplitude-modulating a first carrier with a video signal, and a second carrier having a substantially quadrature phase relationship with the first carrier are separated by one horizontal line of the video signal. A second modulated wave amplitude-modulated by a digital signal whose polarity is inverted for each period corresponding to a scanning cycle and a third carrier having a higher frequency than the first and second carriers are converted into analog audio signals. A television receiver for receiving a vestigial sideband transmission signal synthesized with a modulated third modulated wave, wherein the first receiver reproduces a video signal included in the received transmission signal.
, A second reproduction system for reproducing a digital signal included in the transmission signal, and a third reproduction system for reproducing an analog audio signal included in the transmission signal, wherein the second reproduction The system includes a demodulation unit that synchronously detects the transmission signal and demodulates the digital signal, and a reproduction unit that performs a reproduction process for reproducing the digital signal output from the demodulation unit into a signal before the polarity is inverted. And a television receiver. 3. A first modulated wave obtained by amplitude-modulating a first carrier wave with a video signal and a second carrier wave having a substantially quadrature phase relationship with the first carrier wave are formed by two horizontal lines of the video signal. A vestigial sideband transmission in which the same data is multiplexed with a second modulated wave whose amplitude is modulated by a digital signal whose polarity is inverted every period corresponding to one horizontal scanning period in a period corresponding to a scanning period. A television receiver for receiving a signal, wherein a first signal for reproducing a video signal included in the received transmission signal is provided.
And a second reproduction system for reproducing a digital signal included in the transmission signal, wherein the second reproduction system synchronously detects the transmission signal and demodulates the digital signal. And a reproduction means for performing reproduction processing for reproducing the digital signal output from the demodulation means into a signal before the polarity is inverted. 4. A first modulated wave obtained by amplitude-modulating a first carrier with a video signal, and a second carrier having a substantially quadrature phase relationship with the first carrier are converted into two horizontal signals of the video signal. A second modulated wave obtained by amplitude-modulating the same data in a period corresponding to a scanning cycle with a digital signal whose polarity is inverted every period corresponding to one horizontal scanning period, and higher than the first and second carrier waves; A television receiver for receiving a vestigial sideband transmission signal in which a third modulated wave obtained by modulating a third carrier having a frequency with an analog audio signal is multiplexed. To reproduce the video signal contained in the first
, A second reproduction system for reproducing a digital signal included in the transmission signal, and a third reproduction system for reproducing an analog audio signal included in the transmission signal, wherein the second reproduction The system includes a demodulation unit that synchronously detects the transmission signal and demodulates the digital signal, and a reproduction unit that performs a reproduction process for reproducing the digital signal output from the demodulation unit into a signal before the polarity is inverted. And a television receiver. 5. The television receiver according to claim 1, wherein the digital signal is a PCM audio signal. 6. The demodulation means comprises: carrier recovery means for recovering the second carrier; and synchronous detection means for synchronously detecting the transmission signal using the second carrier reproduced by the carrier recovery means. 6. The method according to claim 1, wherein
The television receiver according to any one of the above. 7. The reproduction means includes delay means for delaying a signal output from the demodulation means for a period corresponding to one horizontal scanning period, and arithmetic processing means for subtracting an input signal and an output signal of the delay means. The reproduction processing is performed based on an output signal of the arithmetic processing means.
A television receiver according to item 1. 8. The transmission signal input to the demodulation means is a signal converted to an intermediate frequency signal and band-limited to the band of the digital signal. 3. The apparatus for receiving and reproducing a multiplex transmission signal according to claim 1. 9. The transmission signal input to the demodulation means is a signal converted to a signal having a lower frequency than an intermediate frequency signal and band-limited to a band of the digital signal. 8. The apparatus for receiving and reproducing a multiplex transmission signal according to any one of 1 to 7. 10. An apparatus according to claim 1, wherein an output of said reproducing means is corrected by a digital signal processing means.
10. The apparatus for receiving and reproducing a multiplex transmission signal according to any one of claims 9 to 9. 11. The reproduction means according to claim 3, wherein said reproduction means performs said reproduction processing by using a selection means for selecting a signal obtained by subtracting identical data having different polarities from an output signal of said arithmetic processing means. Or the multiplex transmission signal receiving and reproducing apparatus according to 4. 12. The multiplex transmission signal receiving / reproducing apparatus according to claim 11, wherein said reproducing process is performed by extending a time axis of an output signal of said selecting means. 13. A first modulated wave obtained by amplitude-modulating a first carrier with a video signal and a second carrier having a substantially quadrature phase relationship with the first carrier are converted into one horizontal line of the video signal. A television receiver, comprising: receiving a transmission signal of a vestigial sideband system in which a second modulated wave amplitude-modulated by a digital signal whose polarity is inverted every period corresponding to a scanning cycle is multiplexed. Machine.