JPH0951320A - Communication equipment - Google Patents

Abstract

PROBLEM TO BE SOLVED: To realize high speed communication with the digital signals of narrow band width in a transmitter/receiver by providing a specified interference removal circuit as against the judgement output signals of I and Q channels for a demodulation part. SOLUTION: In the receiver RX, the signal is converted into the digital signal by an A/D converter 19 and the distortion of a waveform in a transmission line is removed by a transmission line equalizer 20. The interference removal circuit 21 estimating the interference quantity of present time and removing it derives a signal obtained by subtracting a prediced interference component from a past transmission estimation signal system from the output of the transmission line equalizer 20. A judgement circuit 22 processes the signal and estimates a transmission signal. The estimated transmission signal is decoded into a received information system by a decoder 23 and is outputted from an output terminal 25. A transmission estimation signal is fed back and inputted to the interference removal circuit 21 through a delay circuit 24. Thus, a communication equipment which can realize demodulation with sufficient quality can be obtained even if modulation speed is increased, and flexible correspondence to various conditions accompanied by the increase of interference quantity is realized.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention is used for narrow band high speed communication for realizing high speed multi-channel communication under the condition that the frequency band of a transmission line is limited.

[0002]

Duobinary transmission systems have a bandwidth W /
In the transmission line of 2 [Hz] (0 to W / 2 [Hz]),
This is one of the methods for transmitting a binary symbol sequence at an ideal Nyquist rate W [bit / sec]. In order to realize this without intersymbol interference, it is necessary to provide an ideal low pass filter (LPF: low pass filter) with a bandwidth W / 2 [Hz] in the transceiver. However, in a system constructed in this way, the bit error rate of the system becomes sensitive to clock frequency errors and jitter in the transceiver, transmission line characteristics, etc., and the timing recovery circuit and transmission line equalization circuit are Its implementation is extremely difficult because it must be designed strictly. Therefore, in general, a bandpass filter having a roll-off characteristic is used to allow an increase in the required transmission line bandwidth and to design a system resistant to jitter and the like. However, if a cosine characteristic filter is used instead of an ideal low-pass filter, correct transmission can be realized even though there is interference between adjacent signals on the time axis. Such a transmission method is called a duobinary transmission method or a partial response transmission method, and is widely known as a communication method that positively uses interference.

FIG. 9 is a block diagram showing an example of a device configuration using a conventional duobinary transmission system. As shown in FIG. 9, a transmitter TX is composed of an encoder and a transmission filter, and a receiver RX is composed of a reception filter, a transmission path equalization filter, a decision circuit, and a decoder. First, the operation of the transmitter TX will be outlined. A binary transmission information sequence of "1" and "0" is input from the input terminal 91, and after being encoded by the encoder 92 such as scrambling and differential encoding, "-1", "+1". Is output as a bipolar signal. This is subjected to band limitation by the transmission filter 93 and output as a transmission signal from the transmission signal output terminal 94. On the other hand, in the receiver RX, the signal received via the transmission line 17 is input to the reception signal input terminal 95, and the band is limited by the reception filter 96, and then the transmission line equalizer 97.
Is equalized. Subsequently, the equalized received signal is subjected to signal processing in the determination circuit 98, and the received signal sequence can be obtained from the output terminal 100 by the processing of the decoder 99.

Here, a characteristic G obtained by combining the characteristic G _S (ω) of the transmission filter and the characteristic G _R (ω) of the reception filter.
_S a _{(ω) × G R (ω} ), and G (ω). Bandwidth W /
In the duobinary transmission method of 2 [Hz] (0 to fg / 2 [Hz]), the transmission / reception filter G (ω) is -W / 2 to
A band pass filter having a spectrum of cosine characteristic at W / 2 [Hz] is constructed. That is, when Tg = 1 / W, G (ω) and its impulse response are: G (ω) = 2Tg · cos [(Tg / 2) ω], | ω | ≦ π / Tg (1) G (ω) = 0, | ω |> π / Tg (1) g (t) = (4 / π) [(cos (πt / Tg) / (1- (2t / Tg) ² )] ... ( 2) The impulse response g (t) is also referred to as the duobinary signal.
It is a figure showing (t). The horizontal axis represents the band and the vertical axis represents the signal strength. In addition to g (t), g (t-Tg),
The waveforms of g (t−2Tg) and g (t + Tg) are also shown in a dashed line. A signal at time t (t = n · Tg: n is an integer) of the transmission signal sequence b having a value of “+1” or “−1” is b.
If [t], the transmission signal S (t) is given by S (t) = (i = −∞ to ∞) Σb [iTg] g (t−iTg) (3). Here, when confirming FIG. 5, t =
S at the timing of (n + (1/2)) Tg (n is an integer)
If (t) is sampled and a signal of b [t] is obtained, the sampled value is b [n · t] g.
In addition to ((n + (1/2)) Tg), interference signal b [(n + 1) Tg] g ((n- (1/2)) T of the next symbol.
It can be seen that g) is added, but no other signals are involved. Therefore, only two transmission pulses having the same sampling time should be considered for signal demodulation.

On the other hand, at the position of t = n · Tg, it can be seen that an interference signal due to an infinite number of transmission symbols before and after transmitted is superimposed and appears. Therefore, at the receiver, t = (n + (1/2)) T instead of t = n.Tg.
If the signal sampled at the timing of g is decoded, the transmission signal can be demodulated without being affected by an infinite number of interferences. For example, when the transmission signal sequence b is a binary value of "-1" and "+1", under ideal conditions with no noise, time t = (n + (1/2)) Tg
Received signals are "-K", "0", "+ K" (K is a positive number)
Is one of the three values, and if "+ K", then "+1"
It can be estimated that "-1" has been sent if "-K", and that the value of the different sign from the previous estimated transmission signal has been sent if "0". That is, this means that S (n · Tg) = b [n ·
This is the same as sequentially deriving the solution of the equation of Tg] + b [(n-1) Tg]. In the duobinary transmission method, in order to avoid error propagation during demodulation, normally,
Precoding at the transmitter.

Next, a carrierless amplitude phase modulation method (C
The conventional technique of the AP method) will be described. The CAP method is
In the duobinary signal g (t), the cosine wave cos (2πfct) and the sine wave sin (2πfc) of the frequency fc are added.
This is a communication system using two orthogonal signals each multiplied by t) as an impulse response, and is suitable for processing a transmission / reception signal in a digital circuit. FIG. 8 shows a conventional CAP.
It is a block configuration diagram of a communication device using the method. FIG.
(A) is a block diagram of a transmitter. The transmission information sequence input from the input terminal 71 is encoded by the encoder 72, and the transmission filter 73 obtains a digital waveform signal in which the impulse response is convoluted. Here, the I channel (I-c
The impulse response of g (t) cos (2πfct) is used as the convolutional waveform of the transmission filter 73 of (h), and the convolutional waveform of the transmission filter 73 of the Q channel (illustrated as Q-ch) is g (t). sin (2πf
The impulse response of ct) is used. This is converted into an analog signal by a digital / analog (hereinafter referred to as D / A) converter 74, and a low pass filter 75 (LP
A transmission signal is output from the output terminal 76 by processing in F). Therefore, the superimposed waveform of two orthogonal impulse responses becomes a transmission signal.

On the other hand, in the receiver shown in FIG. 8 (b), a received signal input from the input terminal 77 is converted into a digital signal by an analog / digital (hereinafter referred to as A / D) converter 78, and an adaptive filter is used. The transmission line equalization processing is performed by 79. This is determined by the determination circuit 80,
The bit sequence output is decoded by the decoder 81 and output to the output terminal 82.
To obtain the received signal sequence. In the CAP method, the determination circuit 8
The separation of the orthogonal signal at 0, the removal of the interference component, the simplification of the determination circuit scale, and the like are technically studied subjects, and the determination processing is performed by a matching filter or the like while reducing the modulation speed.

[0008]

In the conventional CAP system, the bandwidth of g (t), which is a duobinary signal, is fg / 2 [H] in order to reduce interference components and simplify the decision circuit.
z], there is a problem that only a modulation speed of about fg / 2 [Hz] can be obtained. Further, in order to reduce the bandwidth, it is necessary to lower the frequency fc of the cosine wave and the sine wave which are multiplied by g (t), but the interference component increases or 2
This causes a problem such as a difference in signal amplitude of the impulse response of the wave. In addition, g (t) cos (2πfc ₁ t),
When g (t) sin (2πfc ₁ t), fc ₁ ≠
Setting fc ₂ is not possible because it is restricted by various conditions. Therefore, there is a trade-off relationship between the compression of the transmission band and the increase of the interference amount, and it is desired to develop a new transmission method that is more effective and can flexibly cope with various conditions.

The present invention has been made against such a background, and an object thereof is to provide a communication device capable of performing high-speed communication with a narrow bandwidth. An object of the present invention is to provide a communication device capable of flexibly coping with various conditions accompanied by an increase in the amount of interference.

[0010]

In the device of the present invention, the CAP
Even in the case where two orthogonal transmission impulses as used in the method are transmitted at a modulation rate fs [symbol / second] that is, for example, twice the bandwidth fg / 2 [Hz] of the impulse, and there is much interference. , The target demodulated signal can be easily obtained by separating the interference signal sample sequence into individual impulse signals.

In the transmitter, g (t) cos is used as the transmission impulse signals of the I channel and the Q channel.
(2πfc ₁ t) and g (t) sin (2πfc ₂ t) are used. However, as described above, g (t) is the bandwidth fg.
It is a duobinary signal of [Hz] / 2. These transmission impulse signals are subjected to multivalued modulation of N values at the modulation rate fs [Hz] for each of the I channel and the Q channel, and the signals obtained by adding these two harmonics are transmitted.

In the receiver, first, the received signal is equalized by the transmission line equalizer and sampled at a rate of fs × 2 [samples / second] or more. The transmission impulse signal transmitted at a certain time affects the sample values at the times before and after that as an interference signal, and the magnitude of the interference signal is a value that can be calculated in advance from the transmission impulse signal and the sampling rate. . Therefore, if the received signal is sampled at a speed of fs × 2 or more, for the interference in the signal section until the influence of the interference is reduced to a negligible level, the individual received signals are sequentially obtained from the sample signal sequence of the interference wave. As a result, accurate transmission is possible even in the presence of interference.

Various configurations are conceivable for the circuit that separates the sampled signal sequence of the interference wave into the individual received signals. As an example thereof, the interference signal component at the present time is extracted from the output signal sequence of the past demodulation determination circuit. There is a configuration for deriving, subtracting that amount from the sample value of the received signal at the current time, inputting it to the determination circuit, and obtaining the next estimated transmission signal. At this time,
The circuit that separates into individual signals can be considered to be a circuit that solves a simultaneous equation that describes the relationship between sample values of interference waves and individual received signals.

The transmission impulse signal used by the device of the present invention is an extension of the transmission signal in the CAP system. When fc ₁ = fc ₂ , it becomes the same transmission impulse signal as in the CAP system, but fc ₁ ≠ fc ₂ In this case, the transmission impulse signal has carriers at different frequencies. However, unlike the CAP method, even in the latter two-wave impulse, the device of the present invention can sufficiently separate and demodulate.

Therefore, even under the condition that the required bandwidth is narrowed by reducing fc and the modulation speed is increased up to fg, transmission of sufficient quality is possible.

As described above, the modulation frequency can be increased to fg, and the utilization efficiency of the transmission line band can be improved. Further, fc ₁ and fc ₂ are reduced to fg / 2 or less to enable a narrow band, and in particular, fc ₁ =
At 0, since the power spectrum density is biased to the low frequency band,
It is suitable for wired transmission where the transmission path loss in the high frequency band is large and it is advantageous to use the low frequency band as much as possible.

That is, the present invention comprises an input terminal for inputting a transmission information sequence, an encoder for inputting and encoding this transmission information sequence, and an I channel signal and a Q channel signal according to the output of this encoder. A transmitter including a modulator that modulates a transmission impulse signal having a period T, an input terminal to which an orthogonal signal arrives, a demodulator that inputs the orthogonal signal and determines a signal value, and reception information according to the signal value The communication device includes a receiver including a decoder that outputs a sequence.

Here, a feature of the present invention is that the demodulation unit weights the signals delayed for each cycle T with respect to the outputs of the I channel signal and the Q channel signal obtained as the determination output. An interference cancellation circuit for adding a signal to the I-channel signal and the Q-channel signal of the quadrature signal is provided.

The modulation rate [Hz: symbol / second] of the transmission impulse signal is the bandwidth of the transmission signal [Hz: cycle /
Seconds] can be larger.

The I channel signal is g (t) cos (2πf
c ₁ t) and the Q channel signal is g (t) sin (2πf
c ₂ t), fc ₁ ≠ fc ₂ can be satisfied.

Further, fc ₁ = 0 can be set.

[0022]

DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of the present invention will be described with reference to FIG. FIG. 1 is a block diagram of an apparatus according to the present invention.

According to the present invention, an input terminal 11 for inputting a transmission information sequence, an encoder 12 for inputting and encoding the transmission information sequence, and an I channel signal and a Q channel signal according to the output of the encoder 12 The transmitter TX including the modulator M that modulates the transmission impulse signal of the period T, the input terminal 18 to which the orthogonal signal arrives, the demodulator D that inputs the orthogonal signal and determines the signal value, and the signal A communication device including a receiver RX including a decoder 23 that outputs a reception information sequence according to a value.

Here, the feature of the present invention is that the demodulation unit M weights the signals obtained by delaying the output of the I-channel signal and the Q-channel signal obtained as the determination output for each cycle T. A signal of the quadrature signal I
It is provided with an interference canceling circuit 21 that adds to the channel signal and the Q channel signal.

[0025]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An embodiment of the present invention will be described with reference to FIG. The configuration of the transmitter TX of FIG. 1 is almost the same as the configuration of the conventional CAP system transmitter shown in FIG. 8B, except that the modulation speed is allowed up to fg and that fc ₁ ≠ fc ₂ . In this case, the I channel (illustrated as I-ch) and the Q channel (illustrated as Q-ch) transmission filters 13 have different output bandwidths.

A transmission information sequence is input to the input terminal 11 of the transmitter TX, encoded by the encoder 12, converted into a multilevel signal, and band-limited by the respective I-channel and Q-channel transmission filters. And modulate the transmitted impulse signal. Then, the adder 14 mixes the I-channel and Q-channel signals to obtain a transmission signal from the output terminal 16 of the D / A converter 15. When the transmission signal passes through the transmission line 17, it is input to the input terminal 18 of the receiver RX. Receiver R
At X, the A / D converter 19 converts the signal into a digital signal, and the transmission line equalizer 20 removes the distortion of the waveform on the transmission line. Here, the interference canceling circuit 21 that estimates and cancels the interference amount at the current time derives a signal obtained by subtracting the interference component predicted from the past transmission estimated signal sequence from the output of the transmission path equalizer 20, This is processed by the determination circuit 22 to estimate the transmission signal. The estimated transmission signal is the decoder 2
The received information sequence is decoded by 3 and output from the output terminal 25. The transmission estimation signal is fed back to the interference canceling circuit 21 through the delay circuit 24.

In the apparatus of the embodiment of the present invention, the transmission filter 13
The transmission impulse signal given by the following equation is used as the impulse response of.

I (t) = g (t) cos (2πfc ₁ t) for I channel (4) Q (t) = g (t) sin (2πfc ₂ t) for Q channel (4) where g (T) = (4 / π) [(cos (πt / Tg) / (1- (2t / Tg) ² )] ... (5), and the spectrum is -fg / 2 to fg / 2 [Hz]. 7 shows an impulse response of a cosine characteristic filter having Tg = 1 / fg.

The band of the transmission line 17 is f _limit / 2 [Hz]
When (0 to f _limit / 2 [Hz]), fg, fc
₁ and fc ₂ must satisfy the following conditions.

Fg <f _limit (6) fc ₁ ≦ f _limit − (fg / 2) (7) fc ₂ ≦ f _limit − (fg / 2) (8) When the modulation speed is fs, fs ≦ fg and fs = f
The highest modulation rate will be obtained with g. FIG. 2 is a diagram showing the spectrum of the transmission signal. As is clear from FIG. 2, the required transmission bandwidth f _max [Hz] (0 to f _max
[Hz]), f _max = max [(fg / 2) + fc ₁ , (fg / 2) + fc ₂ ] ... (9).

When the number of modulation signal points on the IQ signal plane is M and the coding rate of the encoder 12 is "k" when the I channel and the Q channel are respectively modulated with multi-level signals. The information transmission rate R [bit / sec] is R = log ₂ M · fs · k (10). As a first specific example, fg = 51.84 [MHz] fc1 = fc2 = 12.96 [MHz] fs = 51.84 [MHz] M = 64 When k = 1, the required transmission bandwidth is f. _max = 38.8 [MH
z], and the information transmission rate is R = 311.04 [bit / sec]. That is, in the device of the present invention, 622.08 is provided by two pairs of transmission lines having a bandwidth of 38.8 [MHz].
It is possible to realize an information transmission rate of [bit / sec].

Further, as a second specific example, fg = 51.84 [MHz] fc1 = 0 [Hz] fc2 = 3.24 [MHz] fs = 51.84 [MHz] M = 64 With parameters, similar information transmission can be achieved with a transmission line bandwidth of 30 [MHz].

Next, the structure of the transmitter TX is shown in FIG. FIG. 3 is a block diagram of the transmitter TX according to the embodiment of the present invention. As shown in Fig. 3 (a), the actual transmitter TX is a ROM (R
It can be easily configured by a digital circuit using an ead only memory). The D-type flip-flop D-FF as a delay element is necessary for encoding and convolution of the impulse response of the I channel and the Q channel, and the ROM has a target signal value depending on the value indicated by the delay device output. The signal data is recorded in advance so that is output. The signal value output from the ROM is the D / A converter 15
Is converted into an analog signal by a low pass filter (LPF) 85, and unnecessary harmonics are removed. The number of stages of the D-type flip-flop D-FF as the delay element depends on the number of states of the encoder 12 and the convolution interval length of the impulse response. Since the impulse response exists infinitely continuously,
To actually convolve this, the main time interval of the signal, that is, before and after t = 0, is targeted for convolution, and the other impulses are ignored and truncated. The truncated impulse response is restored to some extent by the low-pass filter 85 after D / A conversion, so that the transmitter TX
As for, there is no big problem. In the circuit shown in FIG. 3A, the required ROM capacity increases exponentially with the increase in the number of delay elements, but if there are many delay elements, the linearity of the system is used. Then, as shown in FIG. 3B, the transmitter can be composed of a plurality of small-capacity ROMs. FIG. 3B shows a configuration example of the transmitter TX of the second specific example described above. Low pass filter 8
In order to reduce the load of 207, 207.
The output signal speed is set to 36 [MHz].

Next, the structure of the receiver RX according to the present invention will be described. The receiver RX includes a reception filter (not shown as included in the A / D converter 19 in FIG. 1),
A / D converter 19, transmission line equalizer 20, interference cancellation circuit 2
1, a decision circuit 22, and a decoder 23. The apparatus of the embodiment of the present invention is characterized by the interference elimination circuit 21, and here, this portion will be mainly described. The other constituent blocks are the same as conventional ones.

FIG. 4 is a block diagram of the interference canceling circuit 21 of the embodiment of the present invention. The equalized received signal is input to the circuit at a speed twice the modulation frequency. As can be seen from FIG. 4, the outputs of the determination circuit 22 in the past are held by a plurality of stages of delay devices T, and these outputs are multiplied by a coefficient determined for removing interference to obtain a large number. It is configured to be added to the received signal input to the value level determination circuit 81.

Next, a method of deriving a circuit coefficient for removing interference will be described. Received signals input at a rate twice the modulation rate fs are collected for each symbol period Ts, and ω [k · Ts] and ω [k · Ts + (1 /
2)]. However, Ts = 1 / fs. Since the reception signal is affected by the transmission impulses transmitted before and after, the I-ch transmission signal is changed to bI [t], Q-c.
If the transmission signal of h is BQ [t], then ω [k · Ts],
ω [k · Ts + (1/2)] is, ω [k · T _S] = (i = -∞ to ∞) Σ {I (-iT S) bI [(k + i) T _S] + Q (-iT _S ) BQ [(k + i) T _S ]} (11) ω [(k + (1/2)) T _S ] = (i = -∞ to ∞) Σ {I ((-i + (1/2)) T is given by T _S) bQ [(k + i) T _S] ... (12) - _S) bI [(i + (1/2) k + i) T S ] + Q (().

In practice, the receiver RX cannot consider all the impulses that continue on the time axis indefinitely, so only the main time section of the signal is taken into consideration, and that section is -L≤t≤L. And Therefore, equations (11) and (1
2) is ω [kT _S ] = (i = -L to L) Σ {I (-iT _S ) bI [(k + i) T _S ] + Q (-iT _S ) bQ [(k + i) T _S ]} + Δ ₁ (13) ω [(k + (1/2)) T _S ] = (i =-(L-1) to L) Σ {I ((-i + (1/2)) T _S ) bI [( k + i) T _S ] + Q ((− i + (1/2)) T _S ) bQ [(k + i) T _S ]} + Δ ₂ (14) However, Δ1 and Δ2 are t <−L, and
It represents an error caused by a transmission impulse other than the considered section of t> L. If Δ1 and Δ2 are negligible,
Equations (13) and (14) can be transformed into equations for deriving bI [.] And bQ [.] To obtain each coefficient of the multiplier P shown in FIG.

Derivation of coefficients will be described for the case of the first specific example described above. FIG. 6 is a diagram showing a transmission impulse waveform used in the first specific example, where I (t) = g (t) cos (2πfc ₁ t) Q (t) = g (t) sin (2πfc ₂ t) Show each. However, the X axis in FIG. 6 is normalized by the symbol period (symbol time) Ts. In addition, I (t ± Ts) Q (t ± Ts) is also shown in order to make interference easier to understand. From the figure, it can be seen that the overlap of the signals is large and interferes with other signals over a section of about −2Ts ≦ t ≦ 2Ts, but I (t), Q (at t = 0.5n · Ts The value of t) becomes “0”, and it can be seen that there are many places that do not give interference. Again, Table 1 shows each value of I (t) and Q (t) at t = 0.5n · Ts (n is an integer).

[0039]

[Table 1] From Table 1 as well, with the parameters of the first specific example, t = 0.
It is clear that there are many places where the value of the impulse of 5n · Ts becomes “0”. In such a case, the number of multipliers P in the receiver RX can be reduced, which is advantageous in terms of circuit scale when applied to the device of the present invention.

FIG. 7 is a diagram showing a transmission impulse waveform used in the conventional CAP method. For comparison, referring to FIG. 7 and Table 2, fg = 25.92 [MHz], fc = 1
2.96 [MHz], fs = 12.96 [M symbol /
Second], the transmission impulse waveform used in the CAP method will be similarly shown.

[0041]

[Table 2] It can be seen that the CAP method uses a signal with a small interference amount while suppressing the modulation speed so that the determination circuit can directly determine the signal.

Now, for the derivation of the coefficient in the case of the first specific example, here, for the sake of simplicity, −1.5Ts ≦ t ≦ 1.5.
The interference in the section of Ts will be considered. Equation (13),
The following formula can be obtained by calculating the formula of the received signal from (14) and transforming / arranging it.

BQ [k · Ts] = 3bI [(k−1) Ts] + bQ [(k−2) Ts] − (3π / 4) ω [k · Ts] (15) bI [k · Ts] = 2bI [(k-1) Ts] -bQ [(k-1) Ts] + bQ [(k-2) Ts] + [√2] ω [(k + (1/2)) Ts]-(3π / 4) ω [k · Ts] (16) Expressions (15) and (16) indicate that the interference signal can be removed based on the output of the determination circuit up to 2Ts before, and the respective coefficients of the expression are shown in FIG. It is possible to demodulate the transmission signal of the transmitter TX of the device of the present invention by applying it to the corresponding multiplier P of the interference canceling circuit 21.

In this example, the case where fc ₁ = fc ₂ has been described, but even if fc ₁ ≠ fc ₂ , the interference canceling circuit 21 can be designed by the same procedure.

In the apparatus of the embodiment of the present invention, both the transceiver and
Most of it can be realized by a digital circuit and can be modulated and demodulated by baseband digital signal processing. Therefore, the circuit can be integrated, and an inexpensive and stable device can be provided.

[0046]

As described above, according to the present invention,
It is possible to realize a communication device capable of demodulating with sufficient quality even if the modulation speed is increased. Further, it is possible to flexibly cope with various conditions accompanied by an increase in the amount of interference.

By combining the device of the present invention with multi-level modulation, it becomes possible to compress the transmission bandwidth, which makes it possible to effectively utilize wired transmission with a limited transmission path bandwidth. In addition, since the band can be reduced, unnecessary electromagnetic waves radiated from the cable are reduced in the case of wired transmission. Therefore, even if the cable shield is simple, safe transmission is possible.

Furthermore, since a baseband digital demodulation circuit composed of a delay circuit and a multiplier can be used as a receiving circuit for removing intersymbol interference, the circuit can be integrated, and it is inexpensive and stable. It is possible to provide a device that operates in accordance with.

[Brief description of drawings]

FIG. 1 is a block configuration diagram of an apparatus according to an embodiment of the present invention.

FIG. 2 is a diagram showing a spectrum of a transmission signal.

FIG. 3 is a block configuration diagram of a transmitter according to the embodiment of the present invention.

FIG. 4 is a block diagram of an interference canceling circuit according to an embodiment of the present invention.

FIG. 5 is a diagram showing an impulse response g (t).

FIG. 6 is a diagram showing a transmission impulse waveform used in the first specific example.

FIG. 7 is a diagram showing a transmission impulse waveform used in a conventional CAP method.

FIG. 8 is a block configuration diagram of a communication device using a conventional CAP method.

FIG. 9 is a block configuration diagram showing an example of a device configuration using a conventional duobinary transmission system.

[Explanation of symbols]

 11, 18, 71, 77, 91, 95 Input terminal 12, 72, 92 Encoder 13, 73, 93 Transmission filter 14 Adder 15, 74 D / A converter 16, 25, 76, 82, 94, 100 Output Terminal 17 Transmission line 19,78 A / D converter 20,97 Transmission line equalizer 21 Interference removal circuit 22,80,98 Judgment circuit 23,81,99 Decoder 24 Delay circuit 75,85 Low pass filter 79 Adaptation Filter 81 Multi-level level determination circuit 96 Reception filter D Demodulation unit M Modulation unit P Multiplier RX receiver T delay device TX transmitter

Claims (4)

[Claims] 1. An input terminal for inputting a transmission information sequence, an encoder for inputting and encoding the transmission information sequence, and transmission of a cycle T composed of an I channel signal and a Q channel signal according to the output of the encoder. The transmitter includes a modulator that modulates an impulse signal, an input terminal to which a quadrature signal arrives, a demodulator that inputs the quadrature signal and determines a signal value, and outputs a reception information sequence according to the signal value. In a communication device including a receiver including a decoder, the demodulation unit weights a signal obtained by delaying the output of the I channel signal and the Q channel signal obtained as a determination output at each cycle T. A communication apparatus comprising: an interference canceling circuit for adding the quadrature signal to the I channel signal and the Q channel signal of the orthogonal signal. 2. A modulation rate of a transmission impulse signal [Hz:
2. The communication device according to claim 1, wherein the symbol / second] is larger than the bandwidth [Hz: cycle / second] of the transmission signal. 3. An I channel signal is g (t) cos (2π
fc ₁ t) and the Q channel signal is g (t) sin (2π
The communication device according to claim 1 or 2, wherein fc ₁ ≠ fc ₂ when fc ₂ t). 4. The communication device according to claim 3, wherein fc ₁ = 0.