JPH09266688A - Speed detector for brushless motor - Google Patents

Abstract

PROBLEM TO BE SOLVED: To obtain a speed signal having high accuracy at the time of controlled revolution, and to prevent the generation of noises by circuit oscillation, etc., at the time of main-body revolution. SOLUTION: When the saturation of transistors Q1-Q3 at the time of the revolution of a main body is detected by means d1-d3, a supply current to amplifiers R-T is reduced to a supply current to the amplifiers at the time of the unsaturation (the time of control revolution) of the transistors by a means 8, and a conductive mode to coils Lu-Lw is changed over to a second mode, in which a conductive current is formed in a waveform, in which a waveform inflection point becomes dull, by a circuit 4, and spiky voltage at the time of the changeover of the conduction of the coils is removed. When the saturation of the transistors is not detected (the time of control revolution) by the means d1-d3, on the other hand, the supply current to the amplifiers is made larger than the above-mentioned by the means 8, and the conductive mode to the coils is changed over to a first mode, in which the conductive current is formed in a square shape, by the circuit 4. Induced electromotive voltage generated in the coils is detected by a nonconductive section in the first mode by a circuit 9 at that time, and a speed signal is synthesized on the basis of the pure induced electromotive voltage.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a speed detecting device for a brushless motor.

[0002]

2. Description of the Related Art FIG. 13 shows a schematic structure of a brushless motor. The brushless motor includes a stator X, a rotor Y, and
Position detectors Hu, Hv, H as three position detecting means
w and. The stator X has three-phase drive coils Lu, Lv, Lw that are reciprocally energized in the positive and negative directions, and the rotor Y has a plurality of magnetic poles Z arranged to face the drive coils Lu, Lv, Lw. have. Further, the position detectors Hu, Hv, Hw are arranged at positions shifted in phase by 120 ° in electrical angle, and are composed of, for example, three Hall elements.

As a system for driving this brushless motor, there is a so-called hard switching energization system,
A so-called soft switching energization method is known, and these switching energization methods are described in, for example, JP-A-61-42288. Of these, the hard switching energization method (3 phase 120
FIG. 14 shows a drive circuit portion of a brushless motor adopting a switching energization method).

In the figure, Hall elements Hu, Hv and Hw as position detectors are three-phase drive coils Lu and L.
Depending on the rotation of the rotor Y facing v and Lw, FIG.
A three-phase sine wave-like output signal (voltage waveform) as shown in (g) is generated, and this output signal is supplied to amplifiers (differential amplifiers) R, S, T as shown in FIG. Via the signal synthesis circuit SM. The signal synthesizing circuit SM is digitally constituted by a logic circuit, detects a zero-cross point at which the output signals of the amplifiers R, S, T become 0 V, and detects three-phase 120 °.
It outputs a switching waveform signal (rectangular pulse). Power transistors (switching element group) Q31 to Q3
6 constitutes a final-stage driver, and currents are supplied to the three-phase drive coils Lu, Lv, Lw by the three-phase 120 ° switching waveform signals from the signal synthesizing circuit SM as shown in FIG. 5 (A). And the rotor Y rotates as a result. The current flowing through the drive coils Lu, Lv, Lw is detected by the current detection resistor Rs, and the rotation speed of the rotor Y is detected by the speed detector. The control amplifier (control comparator) A41 takes out the difference between the output voltage (speed command voltage) VCTL of the speed detector and the reference voltage VREF as a current command voltage, and further, the current command voltage and the voltage of the current detection resistor Rs. The current feedback amplifier A42 outputs the difference of The signal synthesis circuit SM is controlled by the output signal of the current feedback amplifier A42, and the output signal voltage changes. The filter capacitors Cu, Cv, Cw are the drive coils Lu,
This is a capacitor provided to reduce mechanical noise and electrical noise generated due to a sudden change in current when switching between Lv and Lw energization.

That is, in the hard switching energization method, the signal synthesizing circuit SM synthesizes a three-phase 120 ° switching waveform signal as shown in FIG. 5A based on the output signals of the amplifiers R, S and T. Power transistor Q3
1 to Q36 are added to the base to energize the drive coils Lu, Lv, Lw.

On the other hand, FIG. 15 shows a drive circuit section of a brushless motor adopting the soft switching energization method (see Japanese Patent Laid-Open No. 61-42288).
In the drive circuit section of the brushless motor adopting this soft switching energization method, the hall elements Hu, H
A three-phase sine wave-like output signal (voltage waveform) as shown in FIG. 11 (g) output from v and Hw is sent to the signal combining circuit SSM via the amplifiers (differential amplifiers) R, S and T. Added. The signal synthesizing circuit SSM uses the output signals of the amplifiers R, S and T 100%, and has an analog three-phase 120 °.
The soft switching waveform signal is synthesized, and the output signals of the amplifiers R, S, and T are logarithmically compressed to synthesize a rectangular wave pulse-like soft switching signal with an inflection point blunted. This soft switching signal is logarithmically compressed again by the amplifier circuits A5 and A6 composed of three differential multipliers, and the power transistors (switching element groups) Q31 to Q are passed through the upper stage pre-driver PD1 and the lower stage pre-driver PD2.
In addition to the base of 36, these power transistors Q31 to Q36 pass currents as shown in FIG. 5C to the drive coils Lu, Lv, Lw. As a result, the rotor Y
Rotates. The filter capacitors Cu, Cv, C
The w, the current detection resistor Rs, the control amplifier A41, and the current feedback amplifier A42 function in the same manner as described with reference to FIG. The midpoint voltage of the drive coils Lu, Lv, Lw is detected by the coil midpoint detector, and the detection voltage of the coil midpoint detector and the reference voltage (1/2 of the power supply voltage Vcc) are differentially amplified. A71 and A72 compare and the output is multiplied by the output signal of the current feedback amplifier A42 by the multipliers M1 and M2. The amplifier circuits A5 and A6 are controlled by the output signals of the multipliers M1 and M2 to change the output signal voltage, and the speed control of the rotor Y, the coil current feedback control, and the coil midpoint voltage feedback control are performed.

That is, in the soft switching energization system, the hall element Hu,
Based on the output signals of Hv and Hw, a soft switching signal with a blunted inflection point is synthesized, and this is combined with the amplifier circuits A5 and A.
The logarithmic compression is performed again at 6, and the drive coils Lu, Lv, and Lw of each phase are energized as an energization waveform as shown in FIG. 5 (C).

Here, as is clear from the current waveforms shown in FIGS. 5A and 5C, in the hard switching energization method (see FIG. 5A), the coil energization is turned off at a high speed. Then, the energization is instantaneously stopped, while in the soft switching energization method (see FIG. 5C), the speed at which the coil energization is turned off is slow and the energization amount is gradually reduced. That is, since the zero cross point of each waveform in FIG. 5 is a non-energized section (energized rest section), in the hard switching energization method, the non-energized section to the drive coil is relatively long, and there is a soft switching energization method. Is
The non-energized section to the drive coil is extremely short.

By the way, in the speed detecting device of the brushless motor, it is necessary to detect the pure induced electromotive voltages generated in the three-phase drive coils Lu, Lv, and Lw in order to obtain a highly accurate speed signal. Therefore, this pure induced electromotive voltage is obtained only in the non-energized sections of the drive coils Lu, Lv, and Lw. Therefore, when a soft switching energization method with an extremely short non-energized section is adopted, it is difficult to detect a pure induced electromotive voltage, so that the on / off of the coil energization is clear and the drive coils Lu, Lv, Lw. The hard switching energization method, which has a long non-energized section, is generally adopted. In addition, the code | symbol P in FIG. 5 has shown the detection timing of the induced electromotive voltage.

[0010]

However, in the above hard switching energization method, as described above,
Since the speed of turning off the coil energization is high, FIG.
As shown in FIG. 5, a so-called spike voltage in which the coil voltage is lower than the ground power source or higher than the power source voltage Vcc is generated.
The magnitude of this spike-like voltage is proportional to the off speed of energization and the magnitude of the current flowing through the coil immediately before turning off. Therefore, for example, at the time of starting the motor or at the time of so-called main body rotation such as being out of control due to overload, because the energization switching is performed with a large current, the noise due to the spike-like voltage becomes particularly large, and circuit oscillation etc. However, there is a problem in that mechanical and electrical noise is generated, which adversely affects a device equipped with a motor and causes various problems.

Here, it is conceivable to provide an output snubber capacitor to reduce the spike voltage, but this is not preferable because the accuracy of the speed signal deteriorates during control rotation.

Further, in the soft switching energization method, as described above, since the coil energization is turned off at a low speed, as shown in FIG. Since the non-energized section of the coil is extremely short, it is difficult to detect a pure induced electromotive voltage.

Therefore, according to the present invention, a pure induced electromotive voltage can be detected during control rotation to obtain a highly accurate speed signal, while a spike-like voltage is not generated during rotation of the main body, and mechanical vibration due to circuit oscillation or the like is eliminated. An object of the present invention is to provide a speed detection device for a brushless motor that can prevent the generation of electrical noise.

[0014]

In order to achieve the above object, a speed detecting device for a brushless motor according to a first aspect of the present invention has a stator having an m-phase drive coil that is reciprocally energized in positive and negative directions, and a magnetic pole. A rotor, m position detecting means for obtaining an m-phase output signal according to the rotation of the rotor,
An amplifier that differentially amplifies each output signal of each position detection unit, a plurality of drive transistors that switch energization to the m-phase drive coil, and energization of the drive transistor is controlled based on the output signal of the amplifier. Drive control circuit, saturation detection means connected to each phase of the drive transistor for detecting saturation of the drive transistor, and current supplied to the amplifier when saturation of the drive transistor is detected by the saturation detection means The value is reduced with respect to the value of the current supplied to the amplifier when the drive transistor is not saturated, and the energization mode switching means for switching the energization mode to the drive coil, and the m as the rotor rotates. The induced electromotive voltage generated in the phase drive coil is detected in the non-energized section of the drive signal of the drive coil, and based on this induced electromotive voltage. It is characterized by comprising a speed signal combining circuit for combining the speed signal Te.

In order to achieve the above-mentioned object, the speed detecting device for a brushless motor according to a second aspect of the present invention, in addition to the first aspect,
The drive control circuit energizes the drive coil in the first energization mode in which the current waveform to the drive coil is a square wave when the drive transistor is not saturated, and when the drive transistor is saturated, the current waveform to the drive coil is curved. The feature is that energization is performed in the second energization mode in which the points are dull.

In order to achieve the above object, a brushless motor speed detecting device according to a third aspect of the present invention is the same as the first aspect.
When the drive transistor is unsaturated, the drive control circuit alternately switches the drive coil conduction mode between a first conduction mode in which the current waveform is a square wave and a second conduction mode in which the waveform inflection point is dull. It is characterized in that the current is supplied to the drive coil when the drive transistor is saturated, and the drive coil is supplied with the second power supply mode.

In order to achieve the above object, a brushless motor speed detecting device according to a fourth aspect of the present invention is the same as the first aspect.
The speed command voltage is used as one input, the reference voltage is used as the other input, and a control comparator that compares them is used, and the output signal of this control comparator and the voltage converted by the current detection resistor connected to each drive transistor are used. A current feedback amplifier for comparing and outputting a control signal; and when saturation detection means detects saturation of the drive transistor, one input of the control comparator is controlled so as not to increase further, and It is characterized by including a control signal processing circuit for outputting a control signal to the drive control circuit.

In order to achieve the above-mentioned object, a speed detecting device for a brushless motor according to a fifth aspect of the present invention, in addition to the first aspect,
A stabilizing circuit including a linear amplifier connected to each output terminal of the m position detecting means and a linear amplifier control circuit for keeping the amplitude of each output signal of the linear amplifier constant is provided. The output signal is output to the amplifier.

In order to achieve the above object, a brushless motor speed detecting device according to a sixth aspect of the present invention is the same as the first aspect.
The speed signal synthesizing circuit is connected to two-input diode OR circuits respectively connected between the respective phases of the output terminals of the drive coil, and to the output terminals of these two-input diode OR circuits and to the diodes of the two-input diode OR circuits. And a three-input diode OR circuit connected in reverse direction.

In order to achieve the above object, a speed detecting device for a brushless motor according to a seventh aspect of the present invention includes a stator having an m-phase drive coil that is reciprocally energized in the positive and negative directions, a rotor having magnetic poles, and this rotation. M position detecting means for obtaining m phase output signals according to the rotation of the child, an amplifier for differentially amplifying each output signal of the m position detecting means, and the m
A plurality of drive transistors for switching energization to the drive coils of the phases; a drive control circuit for controlling energization of the drive transistors based on the output signal of the amplifier; and a drive transistor connected for each phase of the drive transistors. Saturation detecting means for detecting saturation, and when the saturation detecting means detects saturation of the drive transistor, the drive control circuit applies an energization signal with a blunted waveform inflection point to the drive coil to drive the drive transistor. When is not saturated, the drive control circuit alternately switches between a square wave-shaped energization signal and an energization signal with a blunted waveform inflection point to energize the drive coil. , The induced electromotive voltage generated in the m-phase drive coil in accordance with the rotation of the rotor in the non-energized section of the energization signal of the drive coil. Out, it is characterized in that and a speed signal combining circuit for combining the speed signal based on the induced electromotive voltage.

According to the brushless motor speed detecting device of the present invention having such a configuration, each output signal of the m position detecting means for obtaining the m-phase output signal according to the rotation of the rotor is output by the amplifier. Each is differentially amplified, and the drive control circuit controls the energization of the plurality of drive transistors based on the output signal of the amplifier, and these drive transistors switch the energization of the m-phase drive coil. Here, when the main body is rotated, the drive transistor is saturated, and when the saturation of the drive transistor is detected by the saturation detection means, the energization mode switching means
The current value supplied to the amplifier is reduced with respect to the current value supplied to the amplifier when the drive transistor is not saturated, that is, when the control rotation is performed, and the drive control circuit controls the energization mode of the drive coil. As the second energization mode, for example, the energization current waveform to the drive coil is switched to a mode in which the waveform inflection point is blunt, while the saturation of the drive transistor is not detected by the saturation detection means, the energization mode is switched. The current value supplied to the amplifier is increased by the means with respect to the current value supplied to the amplifier when the drive transistor is saturated, and the drive control circuit sets the energization mode to the drive coil to the first mode. Is switched to a mode in which the waveform of the current supplied to the drive coil is a square wave. At this time, the speed signal synthesizing circuit detects the induced electromotive voltage generated in the m-phase drive coil with the rotation of the rotor in the non-energized section of the first energization mode, and the speed is calculated based on the induced electromotive voltage. The signals are combined. Therefore, during control rotation, the induced electromotive voltage is detected in the non-energized section of the first energization mode to detect a pure induced electromotive voltage. During main body rotation, the waveform of the energized current to the drive coil has a blunt inflection point. The second energization mode having a different waveform is eliminated to eliminate the spike-like voltage generated when the energization of the drive coil is turned on and off.

According to another aspect of the brushless motor speed detecting device of the present invention, when the drive transistor is not saturated, that is, at the time of control rotation, the drive control circuit causes the first conduction mode in which the current waveform is a square wave. And a second energization mode in which the waveform inflection point is blunted, the energization mode to the drive coil is alternately switched to be energized, and the speed signal synthesizing circuit causes a three-phase drive coil to be generated as the rotor rotates. The induced electromotive voltage generated is detected in the non-energized section of the first energization mode, and the speed signal is synthesized based on the induced electromotive voltage. Therefore, the induced electromotive voltage is detected in the non-energized section of the first energization mode to detect a pure induced electromotive voltage, and the energization mode to the drive coil is the above-mentioned except the first energization mode during the controlled rotation. The second energization mode is set to reduce the spike-like voltage generated when the energization of the drive coil is turned on and off.

Further, in particular, according to the speed detecting device of the brushless motor of the fourth aspect, the speed command voltage is input to one input of the control comparator and the reference voltage is input to the other input, and the control comparator outputs the speed command voltage and the reference voltage. , The output signal of the control comparator is input to one input of the current feedback amplifier, and the voltage converted by the current detection resistor connected to each drive transistor is input to the other input of the current feedback amplifier. These are compared with each other, and the comparison value, that is, a control signal for controlling the speed is output to the drive control circuit to control the speed. At this time, when the saturation of the drive transistor is detected by the saturation detection means, one input of the control comparator is controlled so as not to increase any more, and the saturation of the drive transistor is prevented.

Further, in particular, according to the speed detecting device of the brushless motor of the fifth aspect, when the drive transistor is saturated,
Even if the value of the current supplied to the amplifier is reduced with respect to the value of the current supplied to the amplifier when the drive transistor is not saturated, the amplitude of each output signal of, for example, m position detecting means is changed by, for example, temperature change. If it fluctuates and becomes too large, the energizing current waveform does not enter the second energizing mode
However, the output signals of the m position detecting means are received by the linear amplifier, and the output signals of the linear amplifier are kept constant in amplitude by the linear amplifier control circuit. When output to
Even if the amplitude of each output signal of the m position detecting means varies,
The input to the amplifier is kept constant, and the conduction current waveform is set to the desired first conduction mode and second conduction mode.

Further, particularly when the speed signal is obtained, for example, when the structure of claim 6 is adopted, the structure becomes relatively simple.

[0026]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a block diagram showing a speed detection device for a brushless motor according to an embodiment of the present invention. In this brushless motor, as shown in FIG. 13, a stator X having three-phase drive coils Lu, Lv, Lw, and a rotor Y having a magnetic pole Z arranged to face the stator X are provided. , Hall element Hu as position detecting means for obtaining a three-phase sinusoidal signal in accordance with the rotation of the rotor Y,
Hv and Hw are provided. These Hall elements H
For u, Hv, and Hw, as shown in FIG.
The power supply voltage Vcc is supplied through the drive circuit section 1 and the drive circuit section 1 is connected. Then, according to the output signals of the Hall elements Hu, Hv, Hw output as the rotor Y rotates, the drive circuit unit 1 drives the drive coils Lu,
The drive currents Iu, Iv, and Iw are output to Lv and Lw,
The rotor Y is configured to rotate.

That is, the drive circuit unit 1 is generally provided with Hall amplifiers R, S, T as amplifiers for differentially amplifying the output signals of the Hall elements Hu, Hv, Hw, and the drive coils Lu, Lv, Lw. Drive currents Iu, Iv, I
A plurality of power transistors (driving transistors; switching element groups) Q3 that output w (switch energization)
1 to Q36, and a drive control circuit 4 for controlling the energization of the power transistors Q31 to Q36 based on the outputs of the Hall amplifiers R, S, T and the control signal processing circuit 3.
And

The Hall amplifier R differentially amplifies the output of the Hall element Hu, the Hall amplifier R differentially amplifies the output of the Hall element Hv, and the Hall amplifier T differentially amplifies the output of the Hall element Hw. Has been done. Between the emitters of these Hall amplifiers R, S, T and the ground power source, npn
Transistors Q16 to Q18 are connected to each other, and an npn transistor Q19 forming a current mirror circuit together with these npn transistors Q16 to Q18 is provided. And this npn transistor Q19
The power source voltage Vcc is connected to the collector and the base of the current source through the current source I0.

The drive control circuit 4 receives the output signals of the hall amplifiers R, S and T and receives the drive coils Lu and
An energization waveform synthesizing circuit 5 for forming a waveform that is a basis of energization waveforms for Lv and Lw is provided. This energization waveform synthesis circuit 5
Basically forms a soft switching conduction waveform as the second conduction mode, but increases the bias current Ic flowing through the npn transistors Q16 to Q18 of the Hall amplifiers R, S, and T, that is, the hall current. When the gains of the amplifiers R, S, and T are set high, an energization waveform close to hard switching as the first energization mode can be formed.

The drive control circuit 4 also multiplies the output of the energization waveform synthesizing circuit 5 and the output of the control signal processing circuit 3 which determines the amplitude of the energization waveform, and the output of the multiplier 6. The pre-driver 7 is provided to the bases of the power transistors Q31 to Q36.

The power transistors Q31 to Q36
Among them, the power transistors Q31 to Q33 are transistors on the power supply voltage Vcc side, and Q34 to Q36 are transistors on the ground power supply side. Then, the drive currents of the respective phases all flow into the current detection resistor Rs.

These power supply voltage Vcc side transistor Q31
Q3 to Q33 are each configured to supply a coil drive current when collector currents from pnp transistors Q1 to Q3 as predrivers flow into the bases. The present apparatus is provided with saturation detection circuits d1 to d3 as saturation detection means for detecting saturation of the pnp transistors Q1 to Q3. Since the saturation detection circuits d1 to d3 have the same configuration, the configuration of the saturation detection circuit d1 will be described here.

The saturation detection circuit d1 is connected to the pnp transistor Q4 which forms a current mirror circuit together with the pnp transistor Q1 and the collector and base of the pnp transistor Q4, and outputs a voltage proportional to the current I6 flowing through the pnp transistor Q4. Generated resistance R3,
A pnp transistor Q7 having an emitter connected to a connection point between the collector of the pnp transistor Q1 and the base of the power transistor Q31 is provided. This pn
The base of the p-transistor Q7 has a pnp of the resistor R3.
The terminal on the opposite side of the transistor Q4 is connected and the output terminal of the pre-driver 7 is connected. The saturation detection circuit d1 is configured as described above.

The pnp transistors Q5 and Q6 in the other two phases correspond to the pnp transistor Q4,
The pnp transistors Q8 and Q9 are pnp transistors Q.
7, the resistors R4 and R5 correspond to the resistor R3, and thus the saturation detection circuits d2 and d3 are configured similarly to the saturation detection circuit d1.

Therefore, when the output current of the pre-driver 7 increases, the base potentials of the pnp transistors Q7 to Q9 are lowered by the resistors R3 to R5 and pn.
When the emitter potentials of the p-transistors Q7 to Q9 are raised to increase the base-emitter voltage Vbe and the pnp transistors Q1 to Q3 reach saturation, the pnp transistors Q7 to Q9 become conductive and the pnp transistor Q7 to Q7.
A saturation detection current I7 as information indicating that the pnp transistors Q1 to Q3 are saturated flows from Q9. Further, an npn transistor Q10 into which the saturation detection current I7 flows into the collector and the base and an npn transistor Q11 forming a current mirror circuit together with the npn transistor Q10 are provided.

The control signal processing circuit 3 is for determining the amplitude of the energizing waveform and preventing saturation of the pnp transistors Q1 to Q3. That is, in the control signal processing circuit 3, as shown in FIG. 2, the speed command voltage VCTL is supplied to the plus side input terminal via the resistor R2 as the saturation prevention means and the reference voltage VREF is supplied to the minus side input terminal. Control amplifier A41 for outputting the difference between these input voltages as a current command voltage, and this current command voltage is supplied to the plus side input terminal and is converted to the minus side input terminal by the current detecting resistor Rs. Current feedback amplifier A, which is supplied with the supplied voltage and outputs the difference between these input voltages to the multiplier 6 as a control signal.
42, and the resistor R2 and the control amplifier A4.
The collector of the npn transistor Q11 is connected to the connection point with 1.

Therefore, when the saturation detection current I7 does not flow in the npn transistor Q10, the saturation detection current I7 does not flow in the npn transistor Q11 either, so that the speed command voltage VCT is applied to the positive side input terminal of the control amplifier A41.
L is input as is. On the other hand, the saturation detection current I7 is n
When the current flows to the pn transistor Q10, the saturation detection current I7 also flows to the npn transistor Q11 and the saturation detection current I7 also flows to the resistor 2. Therefore, the control amplifier A4
Even if the speed command voltage VCTL further increases, a voltage drop due to the resistor R2 does not supply the positive input terminal 1 to a voltage higher than that.

Next, the energization mode switching means 8 connected to the drive circuit section 1 having the above structure will be described. The energization mode switching means 8 is provided with the hall amplifiers R, S, T.
The bias current Ic of the above-mentioned pnp transistors Q1 to Q3 is made large when it is not saturated and becomes small when it is saturated, thereby functioning as a gain varying means for changing the gain of the Hall amplifiers R, S, T. The input to the energization waveform synthesizing circuit 5 of the drive control circuit 4 is changed by the energization waveform synthesizing circuit 5 to switch the energization mode to the drive coil between the hard switching energization mode and the soft switching energization mode. There is.

That is, this energization mode switching means 8 is
An npn transistor Q12 that constitutes a current mirror circuit together with the npn transistor Q10 is provided, and the collector of the npn transistor Q12 is connected to the power supply voltage Vcc via a resistor 6. Power supply voltage Vcc is p
The pnp transistor Q which is connected to the emitter of the np transistor Q13 and constitutes a current mirror circuit
The emitters of 14 and Q15 are respectively connected. A ground power source is connected to the collector and base of the pnp transistor Q14 via a current source I8. The resistor 6 is provided at the base of the pnp transistor Q13.
Is connected to the collector of the npn transistor Q12, and the collector of the pnp transistor Q13 is connected to the connection point of the collector of the pnp transistor Q14 and the current source I8. And the above p
The collector of the np transistor Q15 is connected to the connection point between the current source I0 in the drive circuit section 1 and the collector of the npn transistor Q19.

A speed signal synthesizing circuit 9 for synthesizing speed signals is also connected to the drive coils Lu, Lv, Lw. That is, as shown in FIG. 3, the output coil of the drive coil Lu has the cathode common of the diodes D2 and D3, and the output terminal of the drive coil Lv has the diode D.
The cathode commons of the diodes D1 and D6 are connected to the cathode commons of the diodes D4 and D5 and the anodes of the diodes D1 and D2, the diodes D3 and D4, and the diodes D5 and D6, respectively. A power supply voltage Vcc is connected to the common via current sources I12, I34, and I56. A two-input diode OR circuit D12 including the diodes D1 and D2 and the current source I12, and a two-input diode OR circuit D3 including the diodes D3 and D4 and the current source I34.
The anodes of the diodes D7, D8, D9 are connected to the respective anode commons of the 2-input diode OR circuit D56 composed of 4, the diodes D5, D6 and the current source I56.
A ground power source is connected to the cathode commons of D8 and D9 via a current source I789.

Further, as shown in FIG. 1, resistors R, R, R are respectively connected to the output terminals of the drive coils Lu, Lv, Lw, and these resistors R, R, R are star-connected. One input terminal of the comparator 2 is connected to the generated virtual neutral point. As shown in FIG. 3, the other input terminal of the comparator 2 is connected to the cathode common of a three-input diode OR circuit D789 composed of the diodes D7, D8, D9 and the current source I789. The speed signal synthesizing circuit 9 has the above configuration. In the present embodiment, the drive coil L
The resistors R, R, and R are connected to the output terminals of u, Lv, and Lw, respectively, and the virtual neutral point in which these resistors R, R, and R are star-connected is connected to one input terminal of the comparator 2. However, the coil common of the drive coils Lu, Lv, and Lw, that is, the coil neutral point may be connected to one input terminal of the comparator 2.

Next, the operation of the apparatus thus constructed will be described below. When the Hall elements Hu, Hv, Hw detect the magnetic field of the rotor Y having the magnetic pole Z, the Hall elements Hu, Hv, Hw output voltage waveforms u, v, w shown by g in FIG. The three-phase sinusoidal signals u, v, w are differentially amplified by the Hall amplifiers R, S, T and transmitted to the energization waveform synthesizing unit 5, and the energization waveform synthesizing unit 5 supplies the drive coils Lu, Lv, Lw. A waveform that is the basis of the energization waveform of is formed.

On the other hand, the motor control signal, that is, the speed command voltage VCTL, the reference voltage VREF, and the voltage converted by the current detection resistor Rs are input to the control signal processing circuit 3, and the control signal processing circuit 3 drives the drive coil. L
The amplitude of the energization waveform to u, Lv, and Lw is determined.

The multiplier 6 multiplies the waveform that is the basis of the energization waveform to the drive coils Lu, Lv, and Lw and its amplitude, and transmits the result to the power transistors Q31 to Q36 through the predriver 7. (See the current waveforms indicated by j, k, m, n, q, and r in FIG. 6), the drive coils Lu, Lv, and Lw are reciprocally supplied with the drive currents Iu, Iv, and Iw that are 120 ° out of phase with each other. To be done. Then, as the rotor Y rotates, the drive coils Lu, L
Voltage signals indicated by U (solid line), V (dotted line), and W (dotted line) in FIG. 4 are generated at v and Lw. These voltage signals U, V, W include induced electromotive force.

During motor controlled rotation, the transistors Q1 to Q3 do not become saturated because the overcurrent does not flow through the transistors Q1 to Q3, so that the transistors Q7 to Q9 do not conduct and the saturation detection current I7 becomes It does not flow to the transistor Q10. Therefore, the saturation detection current I7 does not flow in the transistor Q11, the speed command voltage VCTL is directly input to the plus side input terminal of the control amplifier A41, and the control amplifier A41 and the current feedback amplifier A42 perform predetermined speed control. It has become so. That is, for example, when the rotation speed is lower than a predetermined speed, the speed command voltage VCTL increases and the energization waveform amplitude input to the multiplier 6 increases, thereby increasing the drive current and increasing the motor speed, while the rotation speed is predetermined. When the speed is higher than the speed, the speed command voltage VCTL decreases and the amplitude of the energization waveform input to the multiplier 6 decreases, so that the drive current is decreased and the motor speed is decreased. In this way, the rotation speed is controlled to be constant. It is like this.

At this time, since the transistors Q10 and Q11 are off, the transistors Q12 and Q13 are also off, the current of the current source I8 flows through the transistors Q14 and Q15, and the bias currents Ic of the Hall amplifiers R, S and T.
Becomes Ic = I0 + I8. That is, the gains of the Hall amplifiers R, S, and T are set to be high, and the drive coils Lu, Lv, and Lw are supplied with the hard switching conduction waveform as shown in FIG. 5 (A).

At this time, again, the drive coils Lu, Lv, Lw
The voltage signals U, V, W (see FIG. 4) generated at the two-input diode OR circuits D12, D3 of the speed signal synthesis circuit 9 are generated.
4 and D56, respectively, and the two-input diode O
R circuit D12, D34, D56 to 3-input diode O
Two input waveforms (U, W phase, U, V) to the R circuit D789.
Phase, V, W phase), only the waveform with the lower potential is output.

That is, the 2-input diode OR circuit D1
The waveform with the lower potential of the U phase and the W phase is output from 2 as the waveform shown by a in FIG. 4, and the one with the lower potential of the U phase and the V phase is output from the 2-input diode OR circuit D34. 4 is output as the waveform shown by b in FIG. 4, and the waveform of the lower potential of the V phase and the W phase is the waveform shown by c in FIG. 4 from the 2-input diode OR circuit D56. ,
Each is output.

These two-input diode OR circuits D12,
The three-phase signal from D34 and D56 is a three-input diode OR
It is input to the circuit D789. This 3-input diode OR
The circuit D789 outputs only the waveform with the highest potential among the three input waveforms indicated by a, b, and c in FIG.
That is, from the 3-input diode OR circuit D789,
The triangular wave signal indicated by d in FIG. 4 is output to one input terminal of the comparator 2.

The triangular wave signal d from the 3-input diode OR circuit D789 and the coil neutral point voltage shown by e in FIG. 4 are respectively inputted to the comparator 2 and compared therewith, and the speed signal is obtained. The speed signal (FG signal) indicated by f in FIG. 4 is output from the comparator 2 which is the output unit of the synthesizing circuit 9.

That is, during control rotation, the induced electromotive voltage is detected in the non-energized section of the hard switching energization mode in which the waveforms of the energized currents to the drive coils Lu, Lv, and Lw are square waves. Is detected, and a highly accurate speed signal f is obtained based on this pure induced electromotive voltage.

On the other hand, for example, at the time of starting the motor or at the time of so-called main body rotation such as when the motor is out of control due to overload,
Since an overcurrent flows through the transistors Q1 to Q3 and the transistors Q1 to Q3 are saturated, the transistor Q1
7 to Q9 are conducted, and the saturation detection current I7 is applied to the transistor Q.
It flows to the transistor Q11 and the resistor 2 as well as the current to the transistor 10. Therefore, even if the speed command voltage VCTL further increases, the increased amount of the speed command voltage VCTL is offset by the voltage drop due to the resistor R2, and a higher voltage is supplied to the plus side input terminal of the control amplifier A41. There is no such thing. That is, the amplitude of the energization waveform input to the multiplier 6 does not increase any more, the further increase of the drive current is suppressed, and the progress of saturation of the transistors Q1 to Q3 is prevented. ing.

At this time, since the transistors Q10 and Q11 are turned on, the transistors Q12 and Q13 are also turned on, the current of the current source I8 starts to flow to the transistor Q13, and the current does not flow to the transistors Q14 and Q15. Result Bias current I of Hall amplifier R, S, T
For c, Ic = I0. That is, the hall amplifier R,
Since the gains of S and T are set low, the soft switching energization waveform as shown in FIG. 5C is flown to the drive coils Lu, Lv, and Lw.

That is, when the main body rotates, the drive coil L
u, Lv, Lw is energized in a soft switching energization mode in which the waveform of the energizing current is a waveform with a dull waveform inflection point,
As a result, the spike-like voltage generated when the drive coils Lu, Lv, Lw are turned on and off is eliminated (see FIG. 5D). Further, since the induced electromotive voltage is detected in the soft switching energization mode, it is difficult to detect the pure induced electromotive voltage, and the accuracy of the speed signal is lower than that in the hard switching energization mode. There is no problem because a highly accurate speed signal is not required during rotation.

The waveforms of the currents supplied to the drive coils Lu, Lv, and Lw are shown by the symbols Z1 and Z in FIG. 7 as the current increases.
As shown by 2, when the amplitude is increased and the transistors Q1 to Q3 are saturated, a waveform Z3 having the maximum amplitude is obtained. However, if the saturation progress of the transistors Q1 to Q3 is left as it is, it is shown by a symbol Z4. As described above, the waveform expands in the horizontal axis direction and the soft switching conduction waveform becomes the hard switching conduction waveform. However,
In the present embodiment, as described above, further increase of the drive current is suppressed, and the progress of saturation of the transistors Q1 to Q3 is prevented, so that the drive coils Lu, Lv, Lw. The waveform of the energizing current to is a waveform Z3 having the maximum amplitude, and is surely a soft switching energizing waveform.

As described above, in this embodiment, the transistors Q1 to Q3 are saturated when the main body is rotated, and when the saturation of the transistors Q1 to Q3 is detected by the saturation detection circuits d1 to d3, the energization mode switching means 8 causes The current value supplied to the Hall amplifiers R, S, T is reduced and driven with respect to the current value supplied to the Hall amplifiers R, S, T when the transistors Q1 to Q3 are not saturated, that is, during control rotation. The drive coil L is controlled by the control circuit 4.
u, Lv, Lw energization mode, drive coil Lu, L
In order to switch to the soft switching conduction mode in which the waveform of the current flowing to v, Lw becomes a waveform with a blunt inflection point,
During the rotation of the main body, it is possible to eliminate the spike-like voltage generated when turning on and off the power supply to the drive coils Lu, Lv, Lw, and the mechanical and electrical noise due to the circuit oscillation caused by the spike-like voltage during the rotation of the main body. Can be eliminated. Also, the transistors Q1 to Q3
Saturation is not detected by the saturation detection circuits d1 to d3, that is, at the time of control rotation, the current value supplied to the Hall amplifiers R, S and T by the energization mode switching means 8 when the transistors Q1 to Q3 are saturated. The current value supplied to the Hall amplifiers R, S, T is increased, and the drive control circuit 4 causes the drive coil L to increase.
u, Lv, Lw energization mode, drive coil Lu, L
Switching to the hard switching energization mode in which the waveforms of the energization currents to v and Lw are square waves, and at this time, the induced electromotive force is detected by the speed signal synthesizing circuit 9 in the non-energization section of the above-mentioned hard switching energization mode, it is pure. The speed signal f can be synthesized based on the induced electromotive voltage, and a highly accurate speed signal can be obtained during control rotation.

Further, the speed command voltage VCTL is input to one input of the control comparator A41, and the reference voltage section VREF is input to the other input. The control comparator A41 compares these, and one of the current feedback amplifiers A42. The output signal of the control comparator A41 is input to the input of the above, the voltage converted by the current detection resistor Rs is input to the other input, and these are compared by the current feedback amplifier A42. Since the control signal for controlling is output to the drive control circuit 4 to perform the speed control, the speed control can be performed with a relatively simple configuration. Further, the saturation of the transistors Q1 to Q3 is detected by the saturation detection circuit d
1 to d3, the resistance R2 controls one input of the control comparator A41 so as not to increase any more. Therefore, it is possible to prevent saturation of the transistors Q1 to Q3 with a relatively simple configuration. It has become.

Further, since the speed signal synthesizing circuit 9 described above has a simple structure, the speed signal f can be obtained with a relatively simple structure.

By the way, when the transistors Q1 to Q3 are saturated, the current value supplied to the Hall amplifiers R, S and T is set to the current value supplied to the Hall amplifiers R, S and T when the transistors Q1 to Q3 are not saturated. On the contrary,
If the amplitudes of the output signals of the Hall elements Hu, Hv, Hw fluctuate and become too large, for example, due to changes in temperature,
There is a fear that the conduction current waveform will be in the hard switching conduction mode instead of the soft switching conduction mode. FIG. 8 shows a configuration for avoiding such a problem.

In this embodiment, the Hall element H
The stabilizing circuit 10 is connected between u, Hv, Hw and the Hall amplifiers R, S, T. This stabilizing circuit 10
Are linear amplifiers 10a to 10c that receive the output signals of the Hall elements Hu, Hv, and Hw, and the linear amplifier 10a.
a linear amplifier control circuit 10d for keeping the amplitude of each output signal a to 10c constant. This linear amplifier control circuit 10d includes the linear amplifiers 10a to 1 described above.
0c Full-wave rectification circuit for full-wave rectifying each output of the Hall elements Hu, Hv, Hw in order to obtain a DC voltage signal proportional to the amplitude of the output of the Hall elements Hu, Hv, Hw by receiving each output signal A three-phase synthesizing circuit 10i including 10e, 10f, 10g and an adder 10h for adding output signals from these full-wave rectifying circuits 10e, 10f, 10g;
Adder 10h as an output unit of the three-phase synthesis circuit 10i
Of the linear amplifiers 10a to 10c so that the amplitude of the output signal of the linear amplifiers 10a to 10c becomes constant based on the smoothing signal smoothed by the smoothing capacitor C0. And a linear amplifier gain setting circuit Ad that performs setting.

Therefore, when the output signals u, v, w of the Hall elements Hu, Hv, Hw as shown in FIG. 11 (g) are input to the linear amplifiers 10a to 10c, the linear amplifiers 10a to 10c output the signals. Outputs the signals u1, v1, w1 shown in FIG. 9 (a).
When w1 is input to the full-wave rectifier circuits 10e, 10f, 10g, the full-wave rectifier circuits 10e, 10f, 10g
Full-wave rectified signals u2, v2 as shown in FIG.
w2 is output. This full-wave rectified signal u2, v2, w2
Is input to the adder 10h, a DC voltage signal (addition signal) as shown in FIG. 9 (c) is output from the adder 10h.
A smoothing signal as shown in FIG. 9D is input to the linear amplifier gain setting circuit Ad that is smoothed by 0. When this smoothed signal is input to the linear amplifier gain setting circuit Ad, when the output signals of the Hall elements Hu, Hv, Hw are reduced by the linear amplifier gain setting circuit Ad and the DC voltage signal is also reduced. When the gains of the linear amplifiers 10a to 10c are increased and the output signals of the Hall elements Hu, Hv, Hw are increased and the DC voltage signal is also increased, the linear amplifiers 10a to 1c are increased.
The gain of 0c is lowered, so that the linear amplifier 10a
The amplitudes of the output signals of -10c are kept constant.

That is, with the above configuration, even if the amplitudes of the output signals of the Hall elements Hu, Hv, and Hw fluctuate due to, for example, temperature changes, the inputs to the Hall amplifiers R, S, and T are kept constant, and the current is supplied. The current waveform can be set to a desired hard switching conduction mode or soft switching conduction mode. For example, the Hall element Hu,
Even if the amplitudes of the Hv and Hw output signals become too large due to, for example, temperature changes, the Hall amplifier R,
The inputs to S and T are kept constant so that the current waveform is not the hard switching current conduction mode but the desired soft switching current conduction mode.

FIG. 10 is a block diagram showing a main part of a speed detecting device for a brushless motor according to another embodiment of the present invention. This embodiment is different from the previous embodiment shown in FIG. 1 in that the energization mode switching means 8 is replaced with the energization mode switching means 18 shown in FIG.

The energization mode switching means 18 is generally the first
2 is composed of a synthesizing circuit 13, a full-wave rectifying circuit 14, and a mode switching circuit 15.

As shown in FIG. 10, the first combining circuit 13 includes a differential amplifier A connected to the output terminal of the Hall element Hu and a differential amplifier B connected to the output terminal of the Hall element Hv. , A differential amplifier C connected to the output terminal of the Hall element Hw, and a pair of p's connected between the output terminals of the same polarity of the differential amplifiers A, B, and C and the power supply voltage V _CC.
Each current mirror circuit D and E composed of np transistors,
A current mirror circuit F composed of a pair of npn transistors connected between each one of the collectors of the current mirror circuits D and E and the ground power source, and between the collectors of the current mirror circuits D and F. The signal (I1-I2) is output from the connection point to the full-wave rectifier circuit 14 in the subsequent stage.

That is, in the first combining circuit 13, the output signals from the Hall elements Hu, Hv, Hw are received by the differential amplifiers A, B, C, respectively, and the output currents of these differential amplifiers A, B, C are received. Are merged with each other with their respective currents I
1, I2 (see the waveform h shown in FIG. 11) is passed through the current mirror circuits D and E to the current mirror circuit F, and the current I
The difference (I1-I2) between 1 and I2 is the full-wave rectification circuit 1 in the subsequent stage
4 is output.

As shown in FIG. 10, the full-wave rectifier circuit 14 has a push-pull circuit G composed of an npn transistor Q20 and a pnp transistor Q21, and resistors RH1 and RH2 connected to the power supply voltage V _CC . Current mirror circuit H consisting of a pair of pnp transistors
And a resistance voltage dividing circuit including resistances RG1 and RG2 connected in series between the power supply voltage V _CC and the ground power supply. The emitter of the transistor Q20 is connected to the emitter of the transistor Q21, and this connection point is connected to the connection point between the collectors of the current mirror circuits D and F. The bases of the transistors Q20 and Q21 are commonly connected to the connection point of the resistors RG1 and RG2. The collector of the transistor Q20 is connected to one collector and the common base of the current mirror circuit H,
A transistor Q is provided on the other collector of the current mirror circuit H.
21 collectors are connected. Then, from the connection point between the other collector of the current mirror circuit H and the collector of the transistor Q21 to the mode switching circuit 15 in the subsequent stage, the signal I3 is output.
Is configured to output.

That is, in the full-wave rectifier circuit 14,
When I1> I2, the pnp transistor Q21 of the push-pull circuit G is turned on, and the current Ib flowing into the push-pull circuit G is output to the mode switching circuit 15 in the subsequent stage. When I1 <I2, the npn transistor Q20 of the push-pull circuit G is turned on, and the current Ia is supplied from the push-pull circuit G to the current mirror circuit F. The current Ia is also output to the subsequent mode switching circuit 15 via the current mirror circuit H. Then, the above phenomenon is alternately repeated, and the full-wave rectification circuit 14 to the mode switching circuit 15
The so-called full-wave rectified waveform i shown in FIG.

As shown in FIG. 10, the mode switching circuit 15 is connected to the square circuit J connected to the connection point between the other collector of the current mirror circuit H and the collector of the transistor Q21, and the square circuit J. The current mirror circuit K is The squaring circuit J is a well-known circuit including an npn transistor group, and includes a first resistor Rj connected between the transistor group and the power supply voltage V _CC . The current mirror circuit K is composed of a pair of pnp transistors, and has a second resistor Rk connected between the emitter of one pnp transistor and the power supply voltage V _CC . The collector of the transistor Q13 in the previous embodiment shown in FIG. 1 is connected to the connection point between the square circuit J and the current mirror circuit K, and the collector of the other pnp transistor of the current mirror circuit K is connected. , The current source I0 and the npn transistor Q19 in the drive circuit section 1 in the subsequent stage.
The connection point with the collector of is connected. Therefore, when the transistor Q13 is on, the output current I4 'of the squaring circuit J does not flow to the current mirror circuit K side but to the transistor Q13 side, and a current flows from the current mirror circuit K to the drive circuit unit 1. On the other hand, when the transistor Q13 is off, the output current I4 ′ of the squaring circuit J is the current mirror circuit K.
Flowing from the current mirror circuit K to the drive circuit unit 1 and the current I
4 is made to flow in.

That is, in the squaring circuit J of the mode switching circuit 15, the current I3 input to the squaring circuit J is inputted.
By squaring (Ib when I1> I2 and Ia when I1 <I2, and the waveform i shown in FIG. 11) is squared, the tip of the corrugated concave portion where the current value becomes 0 is smoothly curved and convex. Width M of the part in the electrical angle direction (see FIG. 11 (j))
Is transformed into a compressed waveform and the signal I4 'is output.

Here, the output signal I4 'from the squaring circuit J is
Of the first resistor Rj and the waveform width in the electrical angle direction of
It is necessary to set the resistance value of the first resistor Rj so that it has an optimum waveform shape, since it changes depending on. Since there is an inverse relationship between I4 ′ and the current flowing through the first resistor Rj, if the resistance value of the first resistor Rj is too small, the current flowing through the first resistor Rj will increase. After that, the waveform becomes as shown in FIG.
If the resistance value of the first resistor Rj is made too large, the current flowing through the first resistor Rj becomes too small and the waveform becomes as shown in FIG. 12B. Therefore, the optimum resistance value should be set. Is important. A similar operation can be obtained by replacing the first resistor Rj with a current source.

When the motor is controlled to rotate, the saturation detection current I7 is applied to the transistor Q10 as described above.
Predetermined speed control is performed without flowing into. At this time, the transistors Q10 and Q11 are turned off, and the transistor Q
12, Q13 are also turned off, and the output current I of the square circuit J
4 ′ flows to the current mirror circuit K side.

In the current mirror circuit K of the mode switching circuit 15, the current I input to the current mirror circuit K is inputted.
The amplitude of 4 ′ is further changed according to the resistance value of the second resistor Rk, and a sinusoidal current I4 (waveform j shown in FIG. 11) is output to the drive circuit unit 1. Therefore, it is necessary to set the resistance value of the second resistor Rk to an optimum value like the first resistor Rj. That is, these resistors Rj, R
By adjusting k, it is possible to obtain an optimum sinusoidal waveform (waveform j shown in FIG. 11).

The structure of the drive circuit section 1 is similar to that shown in FIG. Therefore, during controlled rotation,
The current I0 + I4 obtained by adding the output I4 of the mode switching circuit 15 and the current of the current source I0 is the Hall amplifiers R, S,
It becomes the bias current Ic of T (Ic = I0 + I4). That is, the current I4 increases (the waveform j shown in FIG. 11).
Convex portion) and the bias current Ic also increase to shift to the hard switching conduction mode, and I4 decreases (FIG. 11).
The concave portion of the waveform j shown in 1) and the original soft switching energization mode are set. Further, the switching between the hard switching energization mode and the soft switching energization mode is made smooth by the sine wave-like signal I4 in an analog rather than digital manner. That is, in FIG. 11, Iu, Iv,
The current indicated by the solid line as Iw is the drive coil Lu, L
Each of v and Lw is energized.

That is, the energization waveforms Iu, Iv, Iw to the drive coils Lu, Lv, Lw indicate the hard switching energization mode as shown in FIG. 5 (E) and FIG. 11 at the time of control rotation. The soft switching conduction mode is mixed, and the hard switching conduction mode is set at the timing P for detecting the induced electromotive voltage for synthesizing the speed signal f. Can be detected. That is, a highly accurate speed signal f can be obtained by the speed signal synthesizing circuit 9 described above.

Further, the switching timing of energization to the drive coils Lu, Lv, Lw (switching timing of non-energization / energization).
Then, since the switching is performed in the state close to the soft switching energization mode, the magnitude of the spike-like voltage is as shown in FIG.
As shown in (F), it can be reduced compared to the conventional hard switching type (see FIG. 5B).

Further, since the mode switching signal I4 for switching between the hard switching energization mode and the soft switching energization mode is an analog waveform like a sine wave, the applicant of the present application filed Japanese Patent Application No. 7-20959. Compared with the brushless motor speed detection device described in (1) in which the mode switching signal is a pulse-shaped rectangular wave, the switching between the hard switching conduction mode and the soft switching conduction mode is made smoother. Therefore, it is possible to eliminate the generation of spike-like voltage due to the switching of the conduction mode.

On the other hand, when the main body is rotating, the saturation detection current I7 flows through the transistor Q10 as described above. That is, the transistors Q10 and Q11 are turned on, the transistors Q12 and Q13 are also turned on, and the output current I4 ′ of the squaring circuit J does not flow to the current mirror circuit K side but to the transistor Q13 side and the mode switching circuit 15 Therefore, no current flows into the drive circuit unit 1, and as a result, the bias current Ic of the Hall amplifiers R, S, T becomes Ic = I0, which is the same as in the previous embodiment.

That is, the gains of the Hall amplifiers R, S and T are set low, and the drive coils Lu and L are set.
v and Lw are shown by solid lines in FIG.
A soft-switching energization waveform indicated by a dotted line as Iu, Iv, and Iw is flown to 1 so that the spike-like voltage can be eliminated when the main body rotates. Of course, at this time, similarly to the previous embodiment, the progress of saturation of the transistors Q1 to Q3 is prevented.

As described above, in the present embodiment, as shown in FIG.
In addition to the effect of the embodiment shown in FIG.
When Q3 is not saturated, that is, during control rotation, the drive control circuit 4 causes the drive coils Lu, Lv, and Lw to be switched between the hard switching conduction mode in which the current waveform is a square wave and the soft switching conduction mode in which the waveform inflection point is blunt. The energization modes are alternately switched to energize, and the speed signal synthesizing circuit 9 detects the induced electromotive voltage generated in the drive coils Lu, Lv, Lw in the non-energized section of the hard switching energization mode to obtain this pure induced electromotive voltage. The speed signals are combined on the basis of the above, and the energization mode to the drive coils Lu, Lv, Lw is set to the soft switching energization mode except for the hard switching energization mode during control rotation.
During control rotation, the spike-like voltage generated when the drive coils Lu, Lv, Lw are turned on and off can be reduced, and mechanical and electrical noise due to circuit oscillation and the like caused by the spike-like voltage during control rotation can be reduced. It can be reduced.

Although the invention made by the present inventor has been specifically described based on the embodiments, the present invention is not limited to the above embodiments, and various modifications can be made without departing from the scope of the invention. Needless to say.

[0082]

As described above, according to the speed detecting device of the brushless motor of the present invention, each output of the m position detecting means for obtaining the m-phase output signal according to the rotation of the rotor by the amplifier. In addition to the basic configuration in which signals are differentially amplified, a drive control circuit controls energization of a plurality of drive transistors based on an output signal of the amplifier, and these drive transistors switch energization to an m-phase drive coil. Then, when the drive transistor is saturated when the main body is rotated and the saturation of the drive transistor is detected by the saturation detection means, the current value supplied to the amplifier is supplied to the amplifier by the conduction mode switching means when the drive transistor is not saturated, that is, The drive current is reduced by the supply current value to the amplifier at the time of control rotation, and the drive is performed by the drive control circuit. The current-carrying mode to the drive coil is switched to a second current-carrying mode, for example, a mode in which the waveform of the current flowing to the drive coil has a dull waveform inflection point, and a spike-like voltage generated when the power supply to the drive coil is turned on and off. On the other hand, when the saturation of the driving transistor is not detected by the saturation detection means (during control rotation), the current value supplied to the amplifier by the conduction mode switching means is
The current value supplied to the amplifier when the drive transistor is saturated is increased, and the drive control circuit sets the energization mode to the drive coil as the first energization mode, for example, the energization current waveform to the drive coil. Is switched to a square wave mode, and at this time, the induced electromotive force generated in the m-phase drive coil with the rotation of the rotor is detected by the speed signal synthesis circuit in the non-energized section of the first energization mode. Since it is configured to synthesize the speed signal based on the purely induced electromotive voltage of the lever, it is possible to eliminate mechanical and electrical noise due to circuit oscillation etc. that occurs due to spike-like voltage when the main body rotates. At the same time, a highly accurate speed signal can be obtained during controlled rotation.

Further, in particular, according to the speed detecting device of the brushless motor in the third or seventh aspect, when the drive transistor is not saturated, that is, at the time of control rotation, the drive control circuit causes the first conduction mode in which the current waveform is a square wave. And a second energization mode in which the waveform inflection point is blunted, the energization mode to the drive coil is switched alternately to energize, and the speed signal synthesizing circuit causes the three-phase drive coil to rotate with the rotation of the rotor. The induced electromotive voltage is detected in the non-energized section of the first energization mode, the speed signal is synthesized based on the pure induced electromotive voltage, and the drive coil is energized except in the first energization mode during control rotation. Since the mode is set to the second energization mode to reduce the spike-like voltage generated when the energization of the drive coil is turned on and off,
In addition to the effects of the present invention described above, it is possible to reduce mechanical and electrical noises due to circuit oscillation and the like that occur due to a spike-like voltage during control rotation, which adversely affects equipment equipped with a motor and causes various problems. Can be prevented.

Further, in particular, according to the speed detecting device of the brushless motor in the fourth aspect, the speed command voltage is input to one input of the control comparator and the reference voltage is input to the other input, and these are input by the control comparator. And the output signal of the control comparator is input to one input of the current feedback amplifier, and the voltage converted by the current detection resistor connected to each drive transistor is input to the other input of the current feedback amplifier. These are compared with each other, and the comparison value, that is, a control signal for controlling the speed is output to the drive control circuit to perform the speed control, while saturation of the drive transistor is detected by the saturation detection means. Since it is configured to prevent saturation of the drive transistor by controlling one input so as not to increase any more, In addition to the effects, a relatively simple configuration, it is possible to prevent saturation of the speed control and the driving transistor.

According to the brushless motor speed detecting device of the fifth aspect, the current value supplied to the amplifier when the drive transistor is saturated is compared with the current value supplied to the amplifier when the drive transistor is not saturated. However, if the amplitudes of the respective output signals of the m position detecting means fluctuate and become too large, for example, due to temperature changes, the energizing current waveform does not become the second energizing mode, and the Although it is in the energization mode, each output signal of the m position detecting means is received by the linear amplifier, and each output signal of this linear amplifier is output to the amplifier while its amplitude is kept constant by the linear amplifier control circuit. , M even if the amplitude of each output signal of the position detecting means fluctuates, the input to the amplifier is kept constant, and the conduction current waveform is set to the desired first conduction mode, second conduction mode. Since those configured so as to conduction mode, in addition to the present invention the above-mentioned effects, it is possible to improve the performance and reliability of the device further.

Further, in particular, according to the speed detecting device for the brushless motor in the sixth aspect, in addition to the effect of the present invention described above, it becomes possible to obtain the speed signal with a relatively simple structure.

[Brief description of drawings]

FIG. 1 is a configuration diagram showing a speed detection device for a brushless motor according to an embodiment of the present invention.

FIG. 2 is a circuit diagram showing a specific configuration of a control signal processing circuit.

FIG. 3 is a circuit diagram showing a speed signal synthesis circuit in detail.

FIG. 4 is a timing chart for explaining the circuit operation of the speed signal synthesis circuit.

FIG. 5 is a current waveform diagram and a voltage waveform diagram that respectively represent the energization state of the drive coil by hard switching, soft switching, and a mixed switching of hardware and software.

FIG. 6 is a current waveform diagram output from the pre-driver to each drive transistor.

FIG. 7 is a waveform diagram showing a current waveform of a drive coil which changes depending on whether the drive transistor is unsaturated or saturated.

8 is a circuit diagram showing a stabilizing circuit attached to the device of FIG. 1. FIG.

FIG. 9 is a timing chart for explaining the circuit operation of the stabilizing circuit.

FIG. 10 is a configuration diagram showing a main part of a speed detection device for a brushless motor according to another embodiment of the present invention.

11 is a timing chart for explaining a circuit operation of each circuit shown in FIG.

FIG. 12 is a waveform chart showing an undesirable example of a signal output from the squaring circuit in the mode switching circuit.

FIG. 13 is an exploded perspective view showing a schematic configuration of a brushless motor.

FIG. 14 is a configuration diagram showing a drive circuit unit of a brushless motor adopting a hard switching energization method.

FIG. 15 is a configuration diagram showing a drive circuit unit of a brushless motor adopting a soft switching energization method.

[Explanation of symbols]

 3 control signal processing circuit 4 drive control circuit 8, 18 energization mode switching means 9 speed signal synthesizing circuit 10 stabilizing circuit 10a to 10c linear amplifier 10d linear amplifier control circuit A41 control comparator A42 current feedback amplifier D1 to D6 2 input diode OR Circuit diode D12, D34, D56 2-input diode OR circuit D789 3-input diode OR circuit d1-d3 Saturation detection means Hu, Hv, Hw Position detection means Lu, Lv, Lw Drive coil Q31-Q36 Drive transistors R, S, T Amplifier R2 Control signal processing circuit resistance (saturation prevention means) Rs Current detection resistance VCTL Speed command voltage VREF Reference voltage X Stator Y Rotor Z Magnetic pole

Claims (7)

[Claims] 1. A stator having an m-phase drive coil that is reciprocally energized in positive and negative directions, a rotor having magnetic poles, and m position detections for obtaining an m-phase output signal according to the rotation of the rotor. Means, an amplifier that differentially amplifies each output signal of the m position detecting means, a plurality of drive transistors that switch energization to the m-phase drive coil, and the drive circuit based on the output signals of the amplifier. A drive control circuit for controlling energization of a transistor, saturation detection means connected for each phase of the drive transistor to detect saturation of the drive transistor, and when saturation of the drive transistor is detected by the saturation detection means, Energizing the drive coil by reducing the current value supplied to the amplifier with respect to the current value supplied to the amplifier when the drive transistor is not saturated. An energization mode switching means for switching the mode, and an induced electromotive voltage generated in the m-phase drive coil with the rotation of the rotor is detected in a non-energized section of an energization signal of the drive coil, and based on the induced electromotive voltage. A speed detection device for a brushless motor, comprising: 2. The drive control circuit performs energization in a first energization mode in which the current waveform to the drive coil is a square wave when the drive transistor is not saturated, and when the drive transistor is saturated, the current to the drive coil is 2. The brushless motor speed detecting device according to claim 1, wherein energization is performed in a second energization mode in which the waveform has a dull waveform inflection point. 3. The drive control circuit is configured so that when the drive transistor is unsaturated, the drive coil has a first current-carrying mode in which the current waveform is a square wave and a second current-carrying mode in which the waveform inflection point is dull. The brushless motor speed detecting device according to claim 1, wherein the drive coil is energized in the second energizing mode when the drive transistor is saturated. 4. A control comparator which compares a speed command voltage with one input and a reference voltage with the other input, and an output signal of this control comparator and a current detection resistor connected to each drive transistor. And a current feedback amplifier that outputs a control signal by comparing the generated voltage, and when the saturation detection means detects saturation of the drive transistor,
The brushless motor according to claim 1, further comprising a control signal processing circuit that controls one input of the control comparator so as not to increase any more and outputs the control signal to the drive control circuit. Speed detector. 5. A stabilizing circuit comprising a linear amplifier connected to each output terminal of m position detecting means, and a linear amplifier control circuit for keeping the amplitude of each output signal of the linear amplifier constant, The speed detecting device for a brushless motor according to claim 1, wherein an output signal of the stabilizing circuit is output to an amplifier. 6. The speed signal synthesizing circuit is connected to each of two phases of output terminals of the drive coil.
An input diode OR circuit; and a three-input diode OR circuit connected to each output terminal of these two-input diode OR circuits and reversely connected to the diodes of the two-input diode OR circuit. The speed detecting device for a brushless motor according to claim 1. 7. A stator having an m-phase drive coil that is reciprocally energized in positive and negative directions, a rotor having magnetic poles, and m position detections for obtaining m-phase output signals according to the rotation of the rotor. Means, an amplifier that differentially amplifies each output signal of the m position detecting means, a plurality of drive transistors that switch energization to the m-phase drive coil, and the drive circuit based on the output signals of the amplifier. A drive control circuit for controlling energization of a transistor, saturation detection means connected for each phase of the drive transistor to detect saturation of the drive transistor, and when saturation of the drive transistor is detected by the saturation detection means, When the drive control circuit energizes the drive coil with an energization signal with a blunted waveform inflection point and the drive transistor is not saturated, the drive control circuit has a square wave shape. An energization mode switching unit that alternately switches an energization signal and an energization signal with a blunted waveform inflection point to switch the energization mode so as to energize the drive coil, and driving of the m-phase as the rotor rotates. A speed signal synthesizing circuit that detects an induced electromotive voltage generated in a coil in a non-energized section of the energization signal of the drive coil and synthesizes a speed signal based on the induced electromotive voltage; Speed detector.