JPH09224064A - Amplifier - Google Patents

Abstract

PROBLEM TO BE SOLVED: To attain the linear amplification of signal for generating a big peak in the envelope power of signal by inputting an input signal to a 1st phase shifting means having non-linear phase shifting quantity frequency characteristics, and inputting its amplified output to a 2nd phase shifting means having frequency characteristics inverse to the phase shifter frequency characteristics of the 1st phase shifting means. SOLUTION: The input signal from an input terminal 1 is supplied through a 1st phase shifting means 21 to an amplifier 22, and the amplified output of the amplifier 22 is outputted through a 2nd phase shifting means 23 to an output terminal 13. Concerning the 1st and 2nd phase shifting means 21 and 23, their respective phase shifting amounts are changed non-linearly to frequencies, and their change characteristics are inverse each other. Thus, even when the signal having the big peak is inputted to the input terminal 1, the phase of each frequency component of that input signal is shifted mutually variously at the phase shifting means 21 and the peak is reduced in the output signal. In such a state, the signal is amplified by the amplifier 22. Therefore, the amplifier 22 can be constituted compact and inexpensive.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention is applied to a common amplifying device for mobile communication, satellite communication, broadcasting system, etc., and has an envelope power peak value (Peak Env) of an input signal.
elope Power: hereinafter referred to as PEP) relates to an amplifying device suitable for amplifying a signal having a remarkably large value compared to the average power.

[0002]

2. Description of the Related Art For example, a transmitter used for communication or broadcasting in a microwave band requires amplification of a multicarrier signal having a band of several MHz to about 10 MHz. The spectrum of the multicarrier signal is schematically shown in FIG.
Centered on N carrier signals, the first to N-th channels # 1 to # 1 in which the spectrum spread by the modulation signal is formed
Consists #N, usually spacing between channels is equal bandwidth B of the multi-carrier signal is a band that covers the first through the N-channel, the center is the center frequency f _o. In this multi-carrier signal, the PEP may significantly increase and reach the number of carriers (N) times the average power, depending on the carrier amplitude and phase conditions of each channel. This causes the required saturation output of the amplifier used in the transmitter to increase, and the transmitter is downsized,
This is a serious obstacle to lower power consumption.

In order to solve such a problem, it has been conventionally proposed that the initial phase of each carrier has a specific relationship. For example, IEEE TRANSACTI
ONS ON COMMUNICATIONS, Vo
l. 41, No. 4. April, 1993, pp. 6
31-635, D.I. R Gimlin and others “On Min
imaging the Peak-to-Avera
ge Power Ratio for the Su
m of N Sinusoids ”, but this method works only if each carrier is unmodulated, that is, if each channel remains a sinusoidal carrier and each carrier is modulated independently. When modulated by a signal, the relationship between the specified initial phases is broken and a large PEP is generated.

A feedforward amplifier is known as one suitable for amplifying the above-mentioned multi-carrier signal with low distortion. The feed forward amplifier is disclosed, for example, in US Pat. No. 5,16, issued Nov. 24, 1992.
6,634. A conventional feedforward amplifier will be briefly described with reference to FIG. The signal Sin input to the input terminal 1 is supplied by the power distributor 2 to the variable attenuator 3, the variable phase shifter 4, and the main amplifier 5.
Are distributed to the main amplification path 14 in which is inserted and the linear path 15 in which the delay means 6 is inserted. These two paths 14,
Each signal passing through 15 is input to the directional coupler 7, the signal of the main amplification path 14 is output as it is to the main signal path 16, and the main amplifier distortion component is output to the distortion amplification path 17 as described later. . The delay means 8 is inserted in the main signal path 16, and the variable attenuation means 9, the variable phase shift means 10, and the auxiliary amplifier 11 are inserted in the distortion amplification path 17. The signals that pass through the main signal path 16 and the distortion amplification path 17 are input to the power combiner 12, combined with the power, and output to the output terminal 13.

The variable attenuation means 3 and the variable phase shift means 4 are
In the power combiner 7, the signal of the main amplification path 14 and the signal of the linear path 15 are added in antiphase, and the distortion component generated in the main amplifier 5 is adjusted to be output to the distortion amplification path 17. Also, the variable attenuation means 9 and the variable phase shift means 10
Indicates that the distortion component contained in the signal of the main signal path 16 and the distortion component amplified by the auxiliary amplifier 11 in the distortion amplification path 17 are added in anti-phase by the power combining means 12, and the distortion is removed at the output terminal 13. The signal is adjusted to be output. The distortion generated in the main amplifier 5 can be roughly classified into distortion due to imperfections in the linear region and distortion due to the saturation characteristic. However, in the feedforward amplifier, all distortion components generated in the main amplifier 5 are input terminal 1 Can be detected by the distortion detector 18 from the power combiner 7 to the power combiner 7, and the distortion can be removed by the distortion remover 19 from the power combiner 7 to the output terminal 13. However, it is essential that the auxiliary amplifier 11 operates linearly.

[0006]

SUMMARY OF THE INVENTION A first object of the present invention is to provide an amplifying device capable of linearly amplifying a signal such as a multi-carrier signal which has a significantly large peak in the envelope power of the signal. It is in. Another object of the present invention is to provide an amplifying device which achieves the first object and which can use an amplifier having a relatively small saturation amplification output and which can be constructed in a small size and at a low cost with a relatively simple structure.

[0007]

According to a first aspect of the present invention, an input signal is input to a first phase shift means having a non-linear phase shift amount frequency characteristic, and an output of the first phase shift means is an amplifier. The amplified output is input to the second phase shifting means having the inverse characteristic of the phase shifter frequency characteristic of the first phase shifting means, and the amplified output of the input signal is obtained as its output.

According to the second aspect of the present invention, the signal separating means obtains the main signal and the difference signal from the input signal, the main signal is supplied to the first signal path, and the difference signal is the first phase shift means. , Said amplifier-second with said second phase shifting means inserted
The signal is supplied to the signal path, and the output of the first signal path and the output of the second signal path are power-combined by the first power combining means and supplied to the output terminal.

The signal separating means is composed of a distortion detecting section in the feedforward amplifier, and the distortion removing section is composed of the first and second signal paths and the first power combining means. Alternatively, the signal separating means is means for separating the input signal into the main signal whose envelope power is limited to a predetermined value or less and the difference signal which is the difference between the main signal and the input signal. , A main amplifier is inserted in the first signal path.
The signal separation means in this case is configured by analog technology or digital technology.

According to a third aspect of the present invention, a first input path is provided between the input terminal and the input terminal of the amplifier,
First switching means for passing an input signal is provided in either of the second input path in which the first phase shift means is inserted, and the first output path and the first output path are provided between the amplifier and the output terminal. Second switching means for passing the amplified output signal is provided in any of the second output paths in which the two phase shift means are inserted, the envelope power of the input signal is detected by the detection means, and the detected envelope power is When the voltage exceeds the threshold value, the control means controls the first and second switching means so that the input signal passes through the second input path and the amplified signal passes through the second output path.

[0011]

DESCRIPTION OF THE PREFERRED EMBODIMENTS FIG. 3 shows the basic structure of the present invention. The input signal from the input terminal 1 is supplied to the amplifier 22 through the first phase shifting means 21, and the amplified output of the amplifier 22 is output to the output terminal 13 through the second phase shifting means 23. The first phase shift means 21 and the second phase shift means 23 respectively change their phase shift amounts in a non-linear manner with respect to frequency, and their change characteristics are opposite to each other as shown by curves 24 and 25 in FIG. 4, for example. Is a characteristic of. As the phase shift amount frequency characteristic of the first phase shifting means 21, for example, the squared phase characteristic of the following equation is used.

Ψ (f) = − α (f−f _O ) ² where α is the coefficient of the square characteristic. In the above equation, the phase shift amount Ψ (f) is a relative value with the phase shift amount at the frequency f _O as a reference (0). This squared phase characteristic can be applied to a chirp filter used in a chirp radar and an initial phase setting method for reducing the ratio (PAPR) of the envelope power peak value (PEP) to the average power of a multi-sine wave carrier signal. . The coefficient α can be expressed as α = πT / B in the chirp filter. Here, T is the period of the chirp signal and B is the frequency band. Further, T is the cycle of the multi-sine wave carrier signal (T = 1 / Δf when the carrier frequency interval is Δf)
In that case, it is an example of a phase setting equation in the initial phase setting method.

With this configuration, even if a signal is input to the input terminal 1 with large peaks 26 and 27 as shown in FIG. 5A, the first phase shifting means 21 changes the frequency components of the input signal. Receive different phase shifts, that is, different frequencies but different phases,
For example, as shown in FIG. 5B, the output signal of the first phase shifter 21 has smaller peaks 26 and 27. In the absence of this large peak, the signal is amplified by amplifier 22 as shown in FIG. 5C. Therefore, the saturation power value required for the amplifier 22 can be reduced, and the amplifier 22 can be made small and inexpensive. The output signal amplified by the amplifier 22 has its frequency components in the second phase shift means 23 and the first phase shift means 21.
The phase shift amount is opposite to that of the phase shift amount received in step 1, that is, the sum of the phase shift amounts received by the first and second phase shift means 21 and 23 for each frequency component becomes equal to each other, and the second phase shift means The relative phase of each frequency component of the output signal of 23 is the same as that of the input signal of the first phase shift means 21, and the output signal of the second phase shift means 23 is the same as the input signal waveform of FIG. 5A as shown in FIG. 5D. The waveforms are almost the same and are amplified by the gain G times.

Next, a concrete example of the first and second phase shifting means 21 and 23 will be described. For example, the second-order delay equalizer circuit including the reactance circuit shown in FIG. 6A has a frequency characteristic of a single-peak delay amount as shown in FIG. 6B. Therefore, as shown in FIG. 6C, such a reactance circuit (second-order delay equalization circuit) Z
_{1 to} Z _n are connected in cascade and each characteristic of these circuits Z _{1 to} Z _n is selected to construct a non-linear phase shift circuit that approximates the characteristic required for the first or second phase shift means 21 or 23. can do.

Dispersion delay lines may be used as the phase shifting means 21 and 23. FIG. 7 shows an example in which the distributed delay line is configured by a surface acoustic wave circuit. In this case, in the surface acoustic wave circuit, two sets 32a and 32b of interdigital electrodes (one pair of comb-teeth-shaped electrodes interdigitated with each other) are formed on the piezoelectric substrate 31, and the interdigital electrodes 32a are formed on the interdigital electrode 32a. The applied signal is converted into a surface acoustic wave on the piezoelectric substrate 31, propagates on the piezoelectric substrate 31, reaches the other interdigital transducer 32b, and is converted into an electric signal. In this case, the interdigital electrodes 32a, 32b
The pitch between the electrodes gradually changes in the surface acoustic wave propagation direction, and the changing directions are opposite between the electrodes 32a and 32b. When the one shown in FIG. 7A is used as the first phase shifting means 21, the one shown in FIG. 7B in which the changing direction of the electrode pitch is reversed is used as the second phase shifting means 23 and is used in the opposite. May be.

Still another example of the phase shifting means 21 and 23 is shown in FIG.
A and B show. In the phase shift means 21, the input signal is branched into n by the signal branch means 33, each of which has a center frequency f ₁ to.
is passed through a bandpass filter 34 ₁ to 34C _n of f _n, it is separated for each frequency band, further the delay line 35 _1-3
The signals are delayed by 5 _n for delay amounts T _{1 to} T _n , and then combined by the signal combining means 36. As shown in FIG. 8B, the phase shift means 23 is also branched into n by the signal branch means 37, separated by band-pass filters 38 _{1 to} 38 _n whose center frequencies are f _{1 to} f _n , and delayed by delay lines 39 _{1 to} 39 _n . Delay amount T _{n to} T
After being given a delay of ₁ , the signals are combined by the signal combining means 41. At this time, the sum of the delay amount of the delay line 35i (i = 1, 2, ..., N) and the delay amount of the delay line 39i is T ₁ + T _n .

In the embodiment shown in FIG. 3, when the PEP of the input signal at the input terminal 1 is relatively small, the first phase shift means 2
In some cases, the PEP of 1 may be large. From this point of view, it is desirable that the present invention works effectively only for a signal in a portion having a high PEP. In the feed-forward amplifier (FIG. 2), the PEP in the input signal of its input terminal 1 is extremely large, that is, the main amplifier 5
When the power exceeds the saturation power of, the distortion is remarkably generated, and the PEP of the difference signal between the output signal of the main amplification path 14 and the output signal of the linear path 15 supplied to the distortion amplification path 17 is significantly increased. Therefore, if the present invention is applied to the distortion amplification path 17, it always operates well. An example in which the present invention is applied to a feedforward amplifier is shown by attaching the same reference numerals to portions corresponding to those in FIGS. The configuration of FIG. 2 is otherwise the same as that of FIG. Here, the first phase shift means 21 and the second phase shift means 22 are phase shift means in which the amount of phase shift changes non-linearly with respect to the frequency. The operation of this amplifier will be described below.

FIG. 10A shows an example of the waveform of the envelope power P of the signal Sin input to the input terminal 1. In a multi-carrier signal, the envelope power peak of the signal depends on the amplitude and phase conditions of each carrier. Value (PEP) is the average power Pa
To the envelope power P of the input signal Sin.
Is the input level P corresponding to the saturation output Ps of the main amplifier 5.
When s'is significantly exceeded, a component of large instantaneous power is input to the distortion amplification path 17.

At the input of the first phase shift means 21, as shown in FIG. 10B, a distortion component caused by imperfections in the linear region of the main amplifier 5 is constantly present, but the envelope power of the input signal Sin is constant. A large peak power is generated when P exceeds the input conversion value Ps' of the saturated output. In order to operate the feedforward amplifier with low distortion, the auxiliary amplifier 11 is required to have a linear operation. Therefore, in order to compensate the distortion having a high level peak power caused by the saturation characteristic of the main amplifier 5 as described above, the auxiliary amplifier 1 is conventionally used.
It was necessary to set the required saturation output of 1 to a considerably large value. However, the first phase shifting means 21 shifts the phase of each frequency component of the input signal shown in FIG. 10B, and the peak power is reduced on the output side as shown in FIG. 10C. That is, the peak power is reduced on the input side of the auxiliary amplifier 11, and the required saturation output of the auxiliary amplifier 11 can be reduced accordingly.

The signal shown in FIG. 10D amplified by the auxiliary amplifier 11 is input to the second phase shift means 22, and the second phase shift means 2 is supplied.
2, the phase relationship of the previous input signal is calculated by the first phase shifting means 21.
Phase compensation is performed to restore the phase relationship of the signal input to
The signal shown in FIG. 10E is input to the power combiner 12.
That is, the output of the second phase shifting means 22 is the first phase shifting means 2
A signal obtained by linearly amplifying the input signal of 1 is obtained. The peak power of the envelope power P is shown in FIG.
For example, P ₁ → P ₂ → GP ₂ → GP ₁ changes.
Here, P ₂ <P ₁ , and G is the gain of the auxiliary amplifier 11.

The high PEP portion of the input signal increases the demands on the amplifier and causes distortion. Therefore,
It suffices to separate and amplify the input signal into a main signal in which the envelope power is limited to a predetermined value or less and a difference signal of the difference between the main signal and the input signal. This is the second aspect of the present invention. That is, as shown in FIG. 11, the input signal of the input terminal 1 is input to the signal separation means 43, and the difference between the main signal whose envelope power is limited to a predetermined value (threshold value Lth) and the difference between the input signal and the main signal. It is separated into corresponding difference signals and output to the main signal path 44 and the difference signal path 45, respectively. 12A, B, C
Shows examples of waveforms of the envelope powers of the input signal, the main signal, and the difference signal, respectively. Main signal path 44 and difference signal path 4
5, a main amplifier 46 and an auxiliary amplifier 47 are inserted in each. Here, as the main amplifier 46, an amplifier having excellent linearity, for example, a feedforward amplifier is used. The signals respectively amplified by the main amplifier 46 and the auxiliary amplifier 47 are linearly combined by the power combining means 48 and output to the output terminal 13. The power combiner 48 is composed of a transformer circuit, a hybrid circuit, or the like.

In this embodiment, the envelope power of the main signal is L
limited to th to prevent an increase in peak power and keep the saturated output power of the main amplifier 46 below the corresponding input power value Lth.
Therefore, the main amplifier 46 can be operated with high efficiency and the main signal can be linearly amplified. On the other hand, when Lth is set to about several times the average power of the input signal, the time when the envelope power level of the input signal exceeds Lth, that is, the ratio of the time when the difference signal occurs to the total time is about 10 ⁻³ . The effect of distortion that occurs when amplifying the difference signal is small.

That is, in the prior art using only one amplifier, the saturated output power of the amplifier was selected to be about 10 times the average power of the output signal, but for example, 5 to 6 of the average power of the input at that time was selected. Double the threshold value Lth. The following electronic computer simulation experiments were conducted to confirm that such effects can be obtained. As the input signal of the first phase shift means 21, N = 32 sine wave carrier signals with initial phases set randomly were input, and 1000 combinations of the initial phases were simulated and evaluated. The square phase characteristic is used as the frequency characteristic of the phase shift amount of the first phase shifting means 21, and the coefficient α is represented by the following equation.

Α = π / ((N-1) Δf ² ) That is, Δf is the carrier frequency interval, and B = (N−
1) Δf and T = 1 / Δf. The peak power generation distributions of the input signal and the output signal of the first phase shift means 21 at this time are obtained, and the results are shown as curves 28 and 29 in FIG. The horizontal axis of FIG. 13 represents the peak power of the multi-carrier signal,
This power is standardized by the average power of the input multicarrier signal, and the vertical axis shows the occurrence probability. The generation of an input signal has a large probability for a large PEP, but the generation of an output signal has a PEP of 0.2 or less, that is, a large PEP does not appear. The average value of the PEP of the output signal is
It is about 1/10 of the average value of the PEP of the input signal. This means that the required saturation output of the auxiliary amplifier 47 is 1
This means that it can be reduced to about 1/0.
Since the input signal is amplified in a state where such a large peak does not appear, the saturation power of the amplifier can be made small and the signal is not distorted.

FIG. 14 shows a concrete configuration example of the signal separating means 43. The input signal from the input terminal 1 is branched into two into a limiter path 52 and a linear path 53 by the first power distribution means 51. A variable attenuation means 102, a variable phase shift means 54, a limiter 55 and a second power distribution means 56 are provided in series on the limiter path 52. The second power distribution means 56 distributes the input signal into two, and supplies one of them to the main signal path 44. The linear path 53 is provided with delay means 57, and the signal passing through the linear path 53 and the other output of the second power distribution means 56 are combined by the power combining means 58 and supplied to the difference signal path 45. A power distribution ratio of the first power distribution unit 51 and the second power distribution unit 56, a power combination ratio of the power combining unit 58,
Regarding the attenuation amount of the variable attenuation means 102, the phase shift amount of the variable phase shift means 54, and the delay amount of the delay means 57, both signals of the limiter path 52 and the linear path 53 input to the power combining means 58 are in opposite phases. Combined and adjusted to produce the desired difference signal.

A PIN diode limiter circuit structure can be used as the limiter 55. Variable damping means 5
2 can be configured by using a PIN diode, and the variable phase shift means 54 can be configured by using a circulator, a varactor diode, or the like, and both commercially available products can be used.
The delay means 57 can be constructed using a delay line. Another specific example of the signal separating means 43 is shown in FIG. Input signal is A / D
The converter 61 converts the digital signal into a digital signal, and the digital signal is limited by the limiting processing means 62.
A larger digital value is output as Lth, and a value smaller than Lth is output as it is as a digital main signal, and this digital main signal is subtracted from the output digital signal of the A / D converter 61 by the difference means 63. The signal is obtained. The digital main signal and the digital difference signal are converted into analog signals by the D / A converters 64 and 65, respectively, and high frequency components of these analog signals are removed by the low-pass filters 66 and 67, respectively. It is output to the signal path 45. Each function of the limiting processing means 62 and the difference means 63 can be processed by a microprocessor.

FIG. 16 shows an example in which a digital multicarrier signal is generated by digital signal processing, a digital main signal and a digital difference signal are further obtained, and these are used as analog main signals and difference signals. Data indicating each carrier frequency f _k (k = 1, 2, ..., N) is set by the frequency setting means 68, and the complex digital carrier signal exp is outputted from the carrier signal generating means 69 k according to the data.
(J2πf _k t) is generated. On the other hand, the complex symbol data sk of the k-th channel is band-limited by the filter 70k, and the complex digital carrier signal exp is multiplied by the multiplier 71k.
Complex multiplication is performed with (j2πf _k t). The k complex multiplication results are added by the adding means 72 to generate a digital complex multicarrier signal u (t). This complex multi-carrier signal u (t) is processed by the limiting processing means 62 and the difference means 69 in the same manner as in FIG. 15, and the digital complex main signal g (t) and the digital complex difference signal d.
(T) and separated.

In-phase component (real part) g of the complex main signal g (t)
_i(T), orthogonal component (imaginary part) g_q(T), complex difference signal d
In-phase component (real part) d of (t)_i(T), orthogonal component (imaginary
Part) d _q(T) are D / A converters 64i and 64, respectively
q, 65i, 65q convert to analog signal,
The analog signals of the low-pass filters 66i, 66q,
High frequencies are removed at 67i and 67q. fill
The output of each of the
Each output of the data 67i and 67q is output by the quadrature modulator 74.
And is quadrature-modulated by the local signal from the local oscillator 75,
Main signal path 44, the difference signal
It is supplied to the path 45.

The above relationship is shown by a formula. If the amplitude component of the baseband signal obtained by band-limiting the symbol data {sk} is ak (t) and the phase component is φk (t),
The complex multicarrier signal u (t) can be expressed by the following equation. u (t) = Σ _{k = 1} ^N _ak (t) exp [j2πf _k + φ _k (t)] = A (t) exp [jΦ (t)] A (t) = | u (t) |, Φ (T) = argu (t) If the threshold amplitude is L, the complex main signal and the complex difference signal can be expressed by the following equations.

When | u (t) | ≦ L, g (t) = u (t) | u (t) |> L and g (t) = Lexp [jΦ (t)] d (t) = u (t ) -G (t) From these complex signals, as in-phase component and quadrature component,
The real part and the imaginary part are output as the in-phase component and the quadrature component of the baseband signal.

G _i (t) = Re [g (t)], g _q (t) = Im [g (t)] d _i (t) = Re [d (t)], d _q (t) = Im [d (t)] The baseband signal is input to the quadrature modulators 73 and 74, the carrier having the frequency f _o is quadrature modulated, and the signals represented by the following equations are output. g (t) = Re [g (t) exp (j2πf _O t)] d (t) = Re [d (t) exp (j2πf _O t)] The signal separation type multicarrier signal generator shown in FIG. Then, the main signal component g (t) and the difference signal component d (t) are sequentially calculated according to the input data of each carrier frequency {fk} and the symbol data {sk}. This calculation is usually done in a microprocessor. {Fk} and {sk}
17, the baseband main signal and the difference signal are calculated in advance and stored in the storage element 76 such as a ROM as shown in FIG. 17, and the main signal is input according to the input of {fk} and {sk}. And the waveform data of the difference signal is stored in the storage element 7.
You may make it read from 6.

As compared with the embodiment shown in FIG. 11, as shown in FIG. 18, delay means 78 is inserted at the input side of the main amplifier 46, and variable attenuating means 79 and variable phase shift means at the input side of the auxiliary amplifier 47. It is also possible to adopt a configuration in which 81 is inserted in series. The delay means 78, the variable attenuation means 79 and the variable phase shift means 81 are provided to correct the amplitude and phase mismatch in the power combining means 48 caused by the transmission characteristics of the main amplifier 46 and the auxiliary amplifier 47. However, these are not limited to the insertion positions shown in FIG. 18, and may be inserted at any positions of the main signal path 44 and the difference signal path 45 where similar effects can be obtained. Further, it is not necessary to provide all the delay means 78, the variable attenuating means 79 and the variable phase shift means 81, and it is only necessary to insert the necessary ones according to the actual circuit characteristics of the main amplifier 46 and the auxiliary amplifier 47.

In the embodiment shown in FIG. 11, as shown in FIG. 19, frequency conversion means 82 may be inserted at the input side of the main amplifier 46 and the auxiliary amplifier 47. The frequency conversion means 82 is composed of mixers 83 and 84 provided for the main signal path 44 and the difference signal path 45, and band-pass filters 85 and 86 and a local oscillator 87 commonly provided. The circuit configuration provided with the frequency conversion means 82 in this way is effective when the circuit provided in the preceding stage of the signal separation means 43 and the input terminal 1 is operated at a low frequency or a low speed. FIG. 20 is an application of the first aspect of the present invention shown in FIG. 3 to the embodiment shown in FIG. The difference signal paths 45 on the input side and the output side of the auxiliary amplifier 47 are provided with the first phase shift means 21 and the second phase shift means 23 used in the embodiment of FIG. 3, respectively. The difference signal separated by the signal separating means 43 has a sharp pulse shape as shown in FIG. 12C, but the first phase shifting means 21
As a result, as described above, the phases of the frequency components of the difference signal, which are in a state close to the same phase, are shifted from each other, the peak power is reduced, and in this state, the frequency components of the amplified output signal are amplified by the auxiliary amplifier 47. Is the second phase shifting means 23
By the same as the state at the input of the first phase shift means 21,
A signal obtained by linearly amplifying the difference signal separated by the signal separation means 43 is obtained.

For the embodiment of FIG. 20 as well, similar to the idea of the embodiment of FIG. 18, as shown in FIG.
The delay means 78, the attenuation means 79 and the phase shift means 81 are provided in the path 44 between the third phase shift amplifier 3 and the main amplifier 46, and the first phase shift means 2 is provided.
When delays, attenuations and phase shifts occur in the first and second phase shift means 23, their influences can be compensated. These insertions are not limited to the insertion positions shown in FIG. 21, and may be inserted at any position in the main signal path 44 and the difference signal path 45 where the same effect can be obtained.
Further, the delay means 78, the attenuation means 79 and the phase shift means 81.
It is not always necessary to insert all of the
The necessary ones are inserted according to the actual circuit characteristics of the first and second phase shifting means 23.

Only a portion of the input signal where PEP is larger than a predetermined value is passed through the path of the first phase shifting means-amplifier-second phase shifting means, and when the PEP is less than the predetermined value, the first and second phase shifting means are operated. An embodiment will be described in which the amplifier is passed through without passing. FIG. 22 shows an example thereof. The signal input to the input terminal 1 is branched by the directional coupler 91, and one of the branched outputs detects the envelope power of the signal 92.
And the detection signal of the detection means 92 is input to the control means 93. The other of the branched outputs of the directional coupler 91 is the first
It is input to the switching means 94, and is switched and supplied to one end of the input path 95a or 95b. This input path 9
The other ends of 5a and 95b are switched and connected to the amplifier 22 by the second switching means 96. In addition, the amplifier 22
Is output to the third switching means 97 and is output to the output path 98a.
Alternatively, it is switched and supplied to one end of 98b. One of the other ends of the output paths 98a and 98b is switch-connected to the output terminal 13 by the fourth switching means 99. The first phase shift means 21 is provided on one of the input paths 95a, and the other output path 9 is provided.
The second phase shifting means 23 is inserted in each 8a. Here, the first phase shift means 21 and the second phase shift means 23 are the same as those in FIG.
Is similar to that described in the embodiment.

When the envelope power of the input signal is detected by the detection means 92 and the detection output is input, the control means 93 receives the PEP value L of the input signal when the value L is less than or equal to the threshold value Lth.
The first to fourth switching means 94, 96, 97, 99 are switching-controlled so that the signal passes through the input path 95b and the output path 98b. When the value L of PEP is larger than Lth, the first to fourth switching means 94, 96, 97, 9 are arranged so that the signal passes through the input path 95a and the output path 98a.
9 is switched and controlled. The control means 93 uses, for example, the frequency interval Δ.
The input / output path is switched at a cycle corresponding to the reciprocal of f. That is, when the PEP of the input signal during one cycle (= 1 / Δf) before the switching timing exceeds the threshold value Lth,
The first to fourth switching means 94, 96, 97, 99 are switching-controlled so as to pass through the input path 95a and the output path 98a. Further, when the PEP becomes less than or equal to the threshold value Lth, the first to fourth switching means 94, 96, 97, 9 are so arranged as to pass through the input path 95b and the output path 98b.
9 is switched and controlled.

The detecting means 92 is composed of, for example, a diode, a capacitor, a resistor and the like. Control means 93
Is an A / D converter, microprocessor, ROM,
A RAM, a D / A converter, or the like can be used, or an analog circuit using an operational amplifier, a resistor, or the like can be used. First to fourth switching means 94, 96, 9
Reference numerals 7 and 99 are semiconductor switches, for example. The operation of this amplifier will be described below.

23A and 23B show examples of the waveform of the envelope power of the signal input to the input terminal 1. FIG.
3A is an example when the value L of PEP is less than or equal to the threshold value Lth, and the signal passes through the input path 95b and the output path 98b, and the amplification operation similar to that of the conventional amplification device is performed. The threshold value Lth is set according to the saturation output of the amplifier 22, and when the PEP value L is Lth or less, the distortion generated based on the saturation characteristic of the amplifier 22 is, for example, 60 dB or more in the signal-to-distortion power ratio. To be set. Figure 23B is PE
This is an example of the case where the value L of P is larger than Lth, and the signal passes through the input path 95a and the output path 98a to be amplified. In this case, as in the case already described, the phase of each frequency component of the input signal is relatively shifted by the first phase shifting means 21, the PEP is reduced and supplied to the amplifier 22, and the output of the amplifier 22 is the second. The phase shift means 23 makes the relative phase of each frequency component similar to that of the input signal at the input terminal 1, and at the output terminal 13, a linearly amplified output is obtained even for a large PEP signal.

In the above description, the first to fourth switching means 94, 96, 97, 99 have their input signals changed to PEs.
Although it is assumed that the control is performed depending on whether P has a portion equal to or greater than Lth, the control can be performed according to the instantaneous fluctuation of the envelope power detected by the detection means 92. in this case,
Normally, the input path 95b and the output path 98b are selected, and when the power level detected by the detection means 92 exceeds the threshold value Lth, the input path 95a and the output path 98a are switched and selected, and after a predetermined time, the original signal path is selected. First to fourth switching means 94, 96, 97, 99 so as to return to 95b, 98b.
It is configured to perform switching control of. The predetermined time is about the width of the peak of the envelope power, and may be almost the reciprocal of the bandwidth of the multicarrier signal.

When an attenuation bias and a delay (the same amount of delay added to each frequency component) occur in the first phase shift means 21 or the second phase shift means 23, the same attenuation amount and delay amount are used. The attenuating means and the delaying means that they respectively have are inserted in the input path 95b or the output path 98b. For example, as shown by the dotted line in FIG. 22, the attenuation means 101 and the delay means 1 are connected to the input path 95b.
11 to the output path 98b, the attenuation means 102 and the delay means 1
Insert 12.

As shown in FIG. 24, the second switching means 96 and the fourth switching means 99 in FIG. 22 may be replaced with path coupling / branching means 103 and 104, respectively. The input paths 95a and 95b are connected by the path combining / branching means 103 and connected to the amplifier 22. First switching means 94
As a result, regardless of which of the input paths 95a and 95b is selected, the signal that has passed through the selected input path is input to the amplifier 22. Route coupling / branching means 1
Also for 04, the output path 98 is output by the third switching means 97.
When any of the output paths a and 98b is selected, the signal passing through the selected output path is output to the output terminal 13
Output. Note that the switching means in FIG. 24 may be replaced with the path coupling / branching means only for one of the switching means. As shown in FIG.
In the first switching means 94, the third switching means 97 may be replaced with path coupling / branching means 105 and 106, respectively.
This replacement may be performed only for one of the switching means.

In the above description, the input signal is not limited to the multi-carrier signal, but the present invention is suitable for amplifying a signal including a peak having a significantly large envelope power.

[0043]

As described above, according to the present invention, the peak value of the input signal is reduced by the first phase shift means, and the peak value of the input signal is supplied to the amplifier in that state. Therefore, the saturation output power of the amplifier is relatively small. An inexpensive and compact one can be used, and the output of this amplifier is returned by the second phase shifting means to an envelope power waveform similar to that of the input signal, i.e. a linearly amplified output signal is obtained.

By applying the present invention to the auxiliary amplifier of the feedforward amplifier, the required output power of the main amplifier can be made relatively small. Further, the input signal is limited to a predetermined value of the envelope power or more, and a difference signal between the main signal and the input signal is created.
The required saturated output power of the main amplifier can be made relatively small by combining the powers after the amplifications by the auxiliary amplifiers. In this case, by providing the first and second phase shifting means before and after the auxiliary amplifier, the required output power of the auxiliary amplifier can be reduced and an output signal with excellent linearity can be obtained.

Further, when the envelope power of the input signal is less than the threshold value, the input signal is directly supplied to the amplifier, and when the envelope power is more than the threshold value, the input signal is supplied to the amplifier through the first phase shift means, and the output of the amplifier is changed. Similarly, the required saturation output of the amplifier can be made small by passing it through the second phase shift means.

[Brief description of drawings]

FIG. 1 is a diagram showing an example of a spectrum of a multicarrier signal.

FIG. 2 is a diagram showing a configuration of a conventional feedforward amplifier.

FIG. 3 is a block diagram showing the basic configuration of the present invention.

FIG. 4 is a diagram showing an example of phase shifter frequency characteristics of the first and second phase shift means in FIG.

FIG. 5 is a diagram showing an example of a signal envelope power waveform in each part of FIG.

6A is a diagram showing a second-order delay equalization circuit made up of a reactance circuit, B is a diagram showing an example of delay amount frequency characteristics of the circuit of FIG. 6A, and C is a phase shift means by serial connection of the circuit of FIG. 6A. It is a block diagram which shows the example which comprised.

7A is a plan view showing an example in which the first phase shifting means 21 is composed of a dispersion type delay line, and B is a plan view showing an example in which the second phase shifting means 23 is composed of a dispersion type delay line. FIG.

8A is a block diagram showing another configuration example of the first phase shift means 21, and FIG. 8B is a block diagram showing a configuration example of the second phase shift means 23 corresponding to FIG. 8A.

FIG. 9 is a block diagram showing an example in which the present invention is applied to a feedforward amplifier.

FIG. 10 is a diagram showing a waveform example of envelope power of a signal of each part in FIG.

FIG. 11 is a block diagram showing another embodiment of the present invention.

FIG. 12 is a diagram showing a waveform example of envelope power of a signal of each part of FIG. 11.

FIG. 13 is a diagram showing a simulation result of a probability distribution of each envelope power peak power of an input signal and an output signal in the first phase shift means, for explaining the effect of the present invention.

14 is a block diagram showing a specific example of the signal separation means 43 in FIG.

15 is a functional configuration diagram showing another specific example of the signal separation means 43 in FIG.

16 is a block diagram showing a specific functional configuration example of a signal separation unit 43 and a multicarrier signal generation unit in FIG.

FIG. 17 is a block diagram showing another example of a multicarrier signal generation unit.

FIG. 18 is a block diagram showing a modification of the embodiment of FIG.

FIG. 19 is a block diagram showing still another modification of the embodiment of FIG.

FIG. 20 is a block diagram showing still another modification of the embodiment of FIG.

FIG. 21 is a block diagram showing still another modification of the modification of FIG. 20.

FIG. 22 is a block diagram showing an embodiment of the third aspect of the present invention.

23 is a waveform chart for explaining the operation of the embodiment in FIG.

FIG. 24 is a block diagram showing another embodiment of the third aspect of the present invention.

FIG. 25 is a block diagram showing still another embodiment of the third aspect of the present invention.

Claims (26)

[Claims] 1. A first phase shifter having an input terminal connected to an input terminal and having a non-linear phase shift amount frequency characteristic, and an amplifier having an input terminal connected to an output terminal of the first phase shifter. An amplifier having an input terminal connected to an output terminal of the amplifier, an output terminal connected to an output terminal, and second phase shifting means having a phase shift frequency characteristic substantially opposite to that of the first phase shifting means; apparatus. 2. The amplifying device according to claim 1, wherein a main signal and a difference signal are obtained from the input signal input to the input terminal, the difference signal is supplied to the first phase shift means, and the main signal is supplied to the first phase shift means. A first power combiner that combines the main signal that has passed through the first signal path and the output signal of the second phase shifter and supplies the combined signal to the output terminal. . 3. The amplifying device according to claim 2, wherein the signal separating means distributes the input signal to the first and second branch terminals, and the power distributing means distributes the input signal to the first and second branch terminals. The main amplification path and the linear path whose ends are connected, and the respective other ends of the main amplification path and the linear path are connected, and the signal that has passed through the main amplification path is supplied to the first signal path as the main signal. A directional coupler that takes the difference between the signal that has passed through the main amplification path and the signal that has passed through the linear path and supplies the difference signal to the first phase shift means, and the main amplifier inserted in the main amplification path. And a first delay means inserted in the linear path, including a second delay means inserted in the first signal path, wherein the first phase shift means, the amplifier, and the second phase shift means are included. The difference signal and the first signal which have passed through the distortion amplification path formed by series connection.
The distortion component in the signal that has passed through the signal path is added in antiphase by the first power combining means. 4. The amplifying device according to claim 3, wherein a variable attenuation means and a variable phase shift means are inserted into at least one of the main amplification path and the linear path, and the first signal path and the distortion amplification path are connected. At least one of the variable attenuation means and the variable phase shift means is inserted in at least one side. 5. The amplifying means according to claim 2, wherein the signal separating means limits the input signal so that its envelope power is not more than a predetermined value, the main signal and the input signal. And a main amplifier inserted in the first signal path. 6. The amplifying device according to claim 5, wherein the signal separating means distributes the input signal to first and second branch terminals, and a first branch terminal of the first power distributing means. A limiter for removing the envelope power of the input signal from the predetermined value or more, and a second power distribution means for branching the output signal of the limiter into two and outputting one of them as the main signal. The delay means connected to the second branch terminal of the first power distribution means, the output signal of the delay means and the other branch output signal of the second distribution means are combined in reverse phase shift and output as the difference signal. Second power combining means for 7. The amplifying device according to claim 5, wherein the signal separating means converts the input signal into a digital signal, and an A / D converter which has a predetermined value or more in the digital signal to the predetermined value. Means for changing the digital main signal, means for subtracting the digital main signal from the digital signal to obtain a digital difference signal, and first and first means for converting the digital main signal and the digital difference signal into analog signals, respectively. 2D / A converter,
The first and second D / A converters are composed of first and second low-pass filters that limit the band of each analog signal and output the main signal and the difference signal, respectively. 8. The amplifying device according to claim 5, wherein the input signal is a multi-carrier signal, and frequency data indicating a frequency of each carrier signal of first to N-th channels (N is an integer of 2 or more) is set. Frequency setting means, carrier signal generating means for generating each of the complex digital carrier signals of the first to Nth channels having a frequency according to the set frequency data, and the complex signals of the first to Nth channels. Multiplication means for complex-multiplying the digital carrier signal and the complex symbol data of the first to N-th channels, and addition means for adding N complex multiplication results to obtain a digital signal of the multi-carrier signal; Means for obtaining a digital main signal in which the value of the envelope power of the digital multi-carrier signal is equal to or more than a predetermined value is changed to the predetermined value; Means for obtaining the digital difference signal by subtracting the digital main signal from the digital multi-carrier signal, means for converting the real part and the imaginary part of the digital main signal into analog signals, and a high frequency signal orthogonal to these analog signals Means for modulating to obtain the main signal, means for respectively converting the real part and the imaginary part of the digital difference signal into analog signals, and means for quadrature modulating the high frequency signal with these analog signals to obtain the difference signal. The signal separating means is constituted by the above. 9. The amplifying device according to claim 5, wherein the first signal path, the first phase shift means, the amplifier,
It includes a transmission characteristic adjusting means that is inserted into at least one of the second signal paths formed by the second phase shifting means and that aligns the transmission characteristics of the first and second signal paths. 10. The amplification device according to claim 1, wherein the input terminal is switched and connected to one end of a first input path and one end of a second input path in which the first phase shift means is inserted. A second switching means for switching and connecting the other end of the first input path and the other end of the second input path to the input end of the amplifier; and one end of the first output path at the output end of the amplifier. Second above
Third switching means for switching and connecting one end of the second output path in which the phase shift means is inserted, another end of the first output path, and another end of the second output path for switching connection to the output terminal. Fourth switching means, detecting means for detecting the envelope power of the input signal, and comparing the detected envelope power with a threshold, and when the envelope power exceeds the threshold, And a control means for connecting the first and second switching means to the second input path and the third and fourth switching means to the second output path, respectively. 11. The amplifying device according to claim 1, wherein the input terminal is switched and connected to one end of a first input path and one end of a second input path in which the first phase shift means is inserted. A first path coupling means for connecting the other end of the first input path and the other end of the second input path to the input end of the amplifier; and one end of the first output path at the output end of the amplifier, Second above
Second switching means for switching and connecting one end of the second output path in which the phase shift means is inserted, and second for connecting the other end of the first output path and the other end of the second output path to the output terminal. The path coupling means, the detecting means for detecting the envelope power of the input signal, the detected envelope power are compared with a threshold value, and when the envelope power exceeds the threshold value, the first switching is performed. Control means for connecting the means to the second input path and the second switching means to the second output path, respectively. 12. The amplifying device according to claim 1, wherein the input terminal is connected to one end of a first input path and one end of a second input path in which the first phase shift means is inserted. A first switching means for connecting the other end of the first input path, the other end of the second input path to the input end of the amplifier, and one end of the first output path for the output end of the amplifier and the first The second path branching means that connects one end of the second output path in which the two phase shift means is inserted, the other end of the first output path, and the other end of the second output path are switched to the output terminal. The second switching means to be connected, the detecting means for detecting the envelope power of the input signal, the detected envelope power and the threshold value are compared, and when the envelope power exceeds the threshold value, The first switching means is connected to the second input path, and the second switching means is connected to the second output path. Control means connected to the road respectively. 13. The amplification device according to claim 10, wherein the control means determines the switching means from the state of the comparison result of the envelope power and the threshold value, and switches the switching means to the first input path and the first input path. It is means for keeping connected to one output path, or to be connected to the second input path and the second output path. 14. The amplifying device according to claim 10, wherein when the control means connects the switching means to the second input path and the second output path, the switching means is operated after a predetermined time. It is means for returning to the connection with the first input path and the first input path. 15. The amplifying device according to claim 10, wherein the attenuation means has an attenuation amount substantially equal to the attenuation amount of the first phase shifting means and the attenuation amount of the second phase shifting means. It is inserted in the input path and the first output path. 16. The amplifying device according to claim 10, wherein the delay means has a delay amount substantially equal to the bias delay amount of the first phase shift means and the bias delay amount of the second phase shift means. It is inserted in the first input path and the first output path. 17. The amplifying device according to claim 1, wherein the first and second phase shifting means are composed of a delay equalizing circuit including a reactance circuit. 18. The amplifying device according to claim 1, wherein the first and second phase shifting means are composed of a dispersion type delay line composed of a surface acoustic wave circuit. 19. The amplifying device according to claim 1, wherein the first and second phase shift means include n (n) input signals.
Is a integer of 2 or more), signal branching means for branching into the n paths, band pass filters of different center frequencies respectively inserted into the n paths, and band pass filters inserted into the n paths. It comprises delay means for delaying the passed signals by mutually different times, and power combining means for power combining the output signals from the n paths and outputting them. 20. The amplifying device according to claim 1, wherein the input signal is a multicarrier signal. 21. A main signal obtained by limiting an input signal input to an input terminal so that an envelope power is equal to or less than a predetermined value,
A signal separating means for separating and outputting a difference signal which is a difference between the main signal and the input signal; a main amplifier for amplifying the main signal; an auxiliary amplifier for amplifying the difference signal; An amplifier device comprising: a power combiner that combines the output and the output of the auxiliary amplifier and outputs the combined power to an output terminal. 22. The amplifying device according to claim 21, wherein the signal separating means distributes the input signal to first and second branch terminals, and a first branch terminal of the first power distributing means. A limiter for removing the envelope power of the input signal from the predetermined value or more, and a second power distribution means for branching the output signal of the limiter into two and outputting one of them as the main signal. A delay circuit connected to the second branch terminal of the first power distribution unit, the output signal of the delay circuit and the other branch output signal of the second distribution unit are combined in opposite phase and output as the difference signal. It comprises a second power combining means. 23. The amplifying device according to claim 22 or 6, wherein at least one of the variable attenuator and the variable phase shifter is connected in series with at least one of the limiter and the delay circuit. 24. The amplifying device according to claim 21, wherein the signal separating means is an A / D converter for converting the input signal into a digital signal, and the digital signal has a value equal to or more than a predetermined value thereof to the predetermined value. Means for changing the digital main signal, means for subtracting the digital main signal from the digital signal to obtain a digital difference signal, and first and first means for converting the digital main signal and the digital difference signal into analog signals, respectively. 2D / A converter,
The first and second D / A converters are composed of first and second low-pass filters that limit the band of each analog signal and output the main signal and the difference signal, respectively. 25. The amplifying device according to any one of claims 5 to 7 and 21 to 24, wherein each frequency of the main signal and the difference signal is respectively on an input side of the main amplifier and an input side of the first phase shift means. A frequency conversion means for increasing the frequency is provided. 26. The amplification device according to claim 21, wherein the input signal is a multicarrier signal, and frequency data indicating a frequency of each carrier signal of first to Nth channels (N is an integer of 2 or more) is set. Frequency setting means, carrier signal generating means for generating each of the complex digital carrier signals of the first to Nth channels having a frequency according to the set frequency data, and the complex signals of the first to Nth channels. Multiplication means for complex-multiplying the digital carrier signal and the complex symbol data of the first to N-th channels, and addition means for adding N complex multiplication results to obtain a digital signal of the multi-carrier signal; Means for obtaining a digital main signal in which the power envelope of the digital multi-carrier signal is changed to a predetermined value or more , Means for obtaining the digital difference signal by subtracting the digital main signal from the digital multi-carrier signal, means for converting the real part and the imaginary part of the digital main signal into analog signals, and a high frequency signal by these analog signals. Means for obtaining the main signal by quadrature modulation, means for converting the real part and the imaginary part of the digital difference signal into analog signals, and means for obtaining the difference signal by orthogonally modulating the high frequency signal with these analog signals. And the signal separating means.