JPH0837476A - Data transmitter-receiver - Google Patents

Abstract

PURPOSE:To reduce the deterioration in the reception sensitivity, to reduce the transmission power or to extend the arrival range. CONSTITUTION:Transmission data are separated by each prescribed bit number by a packet composing device 16, a unique word and an error check bit are added to the data and data packets are formed. A transmitter 10 sends a spread spectrum signal (a) obtained by applying digital modulation to the data packet and applying spread modulation to the result. The spread spectrum signal (a) is given to band pass means 21A-21B, in which partial band components only are extracted to be an intermediate signal bi. Automatic gain adjustment devices 30A-30B, detectors 22A-22B, inverse gain adjustment devices 31A-31B detect the intermediate signal bi and adjust the gain to obtain a detection signal co. Clock recovery devices 25A-25B and decoders 23A-23B discriminate a discrimination data string d'm from the detection signal co. A clock recovery device 25C and a decoder 23C discriminate the discrimination data string d'm from a synthesis detection signal (c) synthesized from the detection signal ci by a synthesizer 32. Moreover, unique word detectors 26A-26C, packet detectors 27A-27C and error detectors 29A-29C are provided.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a data transmission / reception device for performing data transmission using a spread spectrum signal, for example.

[0002]

2. Description of the Related Art A spread spectrum communication system has good transmission characteristics in a multipath environment and has a capability of eliminating an interfering signal. Therefore, it is suitable for use in wireless LAN and other in-house wireless data communication and power line carrier data communication. It is attracting attention as a suitable method. One of the frequency bands of the radio waves provided to the wireless LAN is the industrial scientific medical frequency band (IS
M band) is currently assigned. The ISM band is a frequency band used by devices that use strong electromagnetic waves such as microwave ovens, so that a transmitter / receiver used in a wireless LAN can normally transmit data even under extremely high levels of interfering waves. Is required.

As a data transmission / reception apparatus which has taken measures against interference in response to such a request, for example, Japanese Patent Application No. 5-21282.
There is 8. The configuration and operation of this conventional data transmitting / receiving apparatus will be described below with reference to the drawings.

FIG. 12 is a block diagram showing an example of the above data transmitting / receiving apparatus.

In FIG. 12, 10 is a transmitter, and 20 '.
Is a receiver, 16 is a packet assembler, 11 is a differential encoder, 12 is a phase modulator, 14 is a spreading modulation multiplier, 1
5 is a clock generator, 13 is a spread modulation signal generator, 21
A to 21B are bandpass means, 22A to 22B are detectors,
23A to 23B are decoders, 25A to 25B are clock regenerators, 26A to 26B are unique word detectors, and 27A.
27B are packet extractors, 29A to 29B are error detectors, and 24 is a selection selector. Then, FIG. 6 is a schematic diagram of a spectrum of a signal of each unit.

In the transmitter 10 of FIG. 12, the data d, which is a bit string, is formed into a packet by the packet assembler 16, and the burst spread spectrum signal a corresponding to each packet is output as a transmission signal. That is, the transmission data is first divided into a predetermined number of bits, a preamble, a unique word, etc. are added to this information data, and a CRC (Cyclic Redundacy C
heck) An error detection bit such as a code is added to form a data packet. These data packets are taken in in synchronization with the symbol clock CK of the period T, differentially encoded by the differential encoder 11, modulated by the phase modulator 12, and burst-like primary modulation of the symbol period T. It becomes the signal p. For the primary modulation method, for example, the case of two-phase differential phase modulation will be described as an example. The primary modulation signal p has the same phase as the previous symbol when the data d is 1, and the previous symbol when the data d is −1. The opposite phase (2 of ± 1)
Value data). The spread modulation signal generator 13 generates a spread modulation signal q having the same period as that of the symbol clock CK in synchronization with the symbol clock CK. A chirp signal or the like is used as the spread modulation signal q. The spread modulation multiplier 14 multiplies the primary modulation signal p and the spread modulation signal q to obtain a spread spectrum signal a.

The spread spectrum signal a thus obtained enters the receiving device 20 'through the transmission path, is band-limited by the band pass means 21A to 21B, and is the intermediate signal b.
Becomes FIG. 6 exemplifies the outline of the spectrum of the received spread spectrum signal a and the case where the band pass means 21A to 21B can take three bands (B1 to B3). The bandpass means 21A-21B are
Each of them is composed of a band pass filter, and each corresponds to all or a part of the pass bands B1 to B3. The intermediate signal b thus obtained
Is delayed-detected by each of the detectors 22A to 22B, and a detection signal ci is obtained. As disclosed in Japanese Patent Application No. 5-212828, by arranging the center pass bands of the band pass means 21A to 21B and the center input frequencies of the detectors 22A to 22B at 1 / T intervals, the detection signal ci has a phase inversion. Negative and positive pulses are generated depending on the presence or absence. The clock regenerators 25A to 25B generate a regenerated symbol clock from the detected signal ci, and the decoders 23A to 23B sequentially sample / detect the detected signal ci using the timing.
After identification, it is determined to be 1 in the case of positive and −1 in the case of negative depending on the polarity of the sign of the identification point, and the determination data string d ′ _m
Is output. The unique word detectors 26A-26B are
The judgment data string d' _m and the fixed pattern of the unique word are collated at any time, and if a match is detected, a frame signal is output. The packet extractors 27A to 27B extract a decoded data packet composed of information data and error detection bits based on the timing of the frame signal, and the error detector 29
Hand over to A-29B. Each of the error detectors 29A and 29B detects a bit error in the decoded data packet on the basis of the error detection bit and delivers the result to the judgment selector 24, and also the information data in the decoded data packet together with the judgment selector. Hand over to 24. The judgment selector 24 selectively connects only the information data of the system in which no bit error is detected, and outputs the decoded data as the final output of the receiving device 20 ′.

Now, consider the case where the interference wave j shown in FIG. 6 is added to the transmission path. According to the data transceiver shown in FIG. 12, the spread spectrum signal a to be transmitted is
Since the band pass means 21A to 21B for passing only the partial bands B1 to B3 are provided, in the example of FIG. 6, one or more of the band pass means 21A to 21B are set to the pass band B1. If so, the intermediate signal b input to the detector of the system is not affected by the interfering wave j, and normal reception is performed. Therefore, in the other systems, if reception is not normally performed and even if the error detector detects a bit error, if there is at least one system that can avoid the influence of the interfering wave j as described above, The system error detector does not detect bit errors,
The judgment selector 24 selects the information data of that system and outputs it as decoded data, so that normal reception is continued.

With the above-described structure, the signal components of a plurality of bands, which are partial within the band of the spread spectrum signal, are detected at the same time. Therefore, when a very strong interfering wave exists in the signal band, these deterioration factors It is possible to selectively use the signal component in the band with the better reception state while avoiding the influence of, and it is possible to reduce the deterioration of the error rate due to a strong interference wave.

[0010]

However, in the above configuration, a part of the transmission spread spectrum signal band is always selected and used, so that a signal loss occurs. In other words, even if there is no interference,
Since most of the signal components are discarded, there is a problem that reception sensitivity is reduced.

The present invention solves the above-mentioned problems, and in the environment where there is interference, interference is eliminated as usual, but in an environment where there is no interference, the decrease in reception sensitivity is further reduced as compared with the prior art. The purpose is to reduce the transmission power or increase the reach.

[0012]

In order to solve the above problems, a data transmitting / receiving apparatus of the present invention divides transmission data into a predetermined number of bits to form a data packet including a unique word and an error detection bit, and a carrier wave. And a transmitter for outputting a burst spread spectrum signal obtained by multiplying a primary modulation signal obtained by digitally modulating the data packet with a spread modulation signal having a wider band than the primary modulation signal; A receiving device that demodulates a signal and outputs decoded data, wherein the receiving device is
Within the band of the spread spectrum signal, only the signal components of different partial bands are extracted to obtain a detection signal,
A plurality of systems of band pass means, an automatic gain adjuster, a detector, an inverse gain adjuster, and a plurality of systems of clock regenerators, decoders, and unique words for extracting decoded data from the detected signals of the plurality of systems, respectively. A detector, a packet extractor, an error detector, a combiner for combining the detection signals of the plurality of systems into one, and a clock regenerator, a decoder and a unique word detector for extracting decoded data from the combined detection signals. And a packet extractor and an error detector, and the error detector connects the outputs of systems not containing bit errors to obtain the demodulated data.

[0013]

According to the present invention, with the above-described structure, the partial signal components of a plurality of bands within the band of the spread spectrum signal are detected and selected at the same time. Therefore, the ability to avoid a strong interfering wave localized in the signal band. In addition, in a non-jamming environment, it is possible to combine a plurality of detection signals to reduce a decrease in reception sensitivity, reduce transmission power, or increase a reach distance.

[0014]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS A data transmitting / receiving apparatus according to an embodiment of the present invention will be described below with reference to the drawings.

FIG. 1 is a block diagram of a data transmitting / receiving apparatus according to the first embodiment of the present invention, and the configuration of this embodiment will be described with reference to FIG.

In FIG. 1, 10 is a transmitter, 20 is a receiver, 16 is a packet assembler, 17 is a primary modulator,
Reference numeral 14 is a spreading modulation multiplier, 15 is a clock generator, and 13 is a clock generator.
Is a spread modulation signal generator, 21A to 21B are bandpass means, 30A to 30B are automatic gain adjusters, and 22A to 22B.
Is a detector, 31A to 31B are inverse gain adjusters, 32 is a combiner, 23A to 23B and 23C are decoders, and 25A to 2
5B and 25C are clock regenerators, 26A to 26B and 27C are unique word detectors, 27A to 27B and 27C are packet extractors, 29A to 29B and 2C.
9C is an error detector, and 24 is a decision selector.

Further, FIG. 2 shows the automatic gain adjuster 30 of FIG.
1 is a configuration example of A to 30B, in which 301 is an amplifier and 302 is
Is an envelope detector and 303 is a low-pass filter. 3 is a block diagram showing a configuration example of the detectors 22A to 22B, 221 is a symbol delay device, 222 is a multiplier, and 223 is a low-pass filter. An example of the configuration of the primary modulator 17 in the transmission device 10 is the differential encoder 11 and the phase modulator 12 in the transmission device 10 of FIG.
Equivalent to.

Further, FIG. 4, which is a code configuration diagram of an example of a data packet output from the packet assembler 16, will be described below.
The operation of the present embodiment will be described with reference to FIG. 5, which is an explanatory diagram of an example of a data packet observed in the determination data string output from the decoders 23A to 23B, and FIG. 6 which is a schematic diagram of the spectrum of the signal of each unit. explain.

The transmitter 10 of FIG. 1 is the same as that of FIG. 12 described in the item of "Prior Art". That is, the transmission data is first divided into a predetermined number of bits, becomes information data 93, and is added with the preamble 91, the unique word 92, and the error detection bit 94, as shown in the example of FIG. Make up 64. The data packets 61 to 64 are input to the primary modulator 17,
It becomes a burst-like primary modulation signal corresponding to each packet. As the primary modulation method, for example, (differential) phase modulation such as 2, 4, or 8 phase is used, and its basic configuration and operation are the differential encoding of FIG. 12 described in the section “Prior Art”. The configuration and operation of the device 11 and the phase modulator 12 are similar. Since the steep rise and fall of the burst causes the transmission spectrum width to widen, a ramp waveform whose envelope curve changes smoothly may be added to the leading and trailing edges of the burst. The primary modulation signal is the same as in the conventional example of FIG. 12 described in the item “Prior Art”,
Further, it is multiplied by the spread signal q and output from the transmission device 10 as a burst-shaped spread spectrum signal a corresponding to each packet.

The unique word 92 is, as will be described later,
It is a fixed bit pattern string inserted to find a corresponding data packet in the decoding process in the receiving device 20. On the other hand, the error detection bit 94 is
It is a variable bit pattern string inserted to check whether a bit error has occurred in the information data 93 and the error detection bit 94 itself. Error detection bit 94
Is actually a parity code or CRC (Cyclic R
edundacy Check) code is used.

The operation will be described in more detail below by taking the case where the primary modulation method is two-phase differential phase modulation as an example.

In the transmitter 10, the m-th data d _m (in the case of ± 1, binary) of the data packets 61 to 64 output from the packet assembler 16 are fetched in synchronization with the symbol clock CK of the cycle T. , Is modulated by the primary modulator 17, and as its output, a primary modulation signal p which is a two-phase phase modulated wave having a symbol period T is obtained. The spread modulation signal generator 13 generates a spread modulation signal q having the same period as that of the symbol clock CK in synchronization with the symbol clock CK. Spread modulation signal q
Is, for example, a pseudo-random pulse waveform of constant amplitude generated by the pseudo-random sequence. The spread modulation multiplier 14 multiplies the primary modulation signal p and the spread modulation signal q to obtain a spread spectrum signal a.

Now, the data after the differential encoding is represented by δ _m
If the (case of ± 1, 2 _{values), d m = δ m ×} δ m-1 (1) and represented. Therefore, assuming that the frequency of the carrier wave is f _c , Re
When [...] is a real part, the spread spectrum signal a to be transmitted is expressed by the following equation. a (t) = Re [δ m · q (t) · expj (2πf c t)] (2) spread spectrum signal a through the transmission path, the receiving apparatus 2
0, the band is first band-limited by the bandpass means 21A to 21B, and becomes the intermediate signal bi. FIG. 6 shows an outline of the spectrum of the received spread spectrum signal a and three bands that the band pass means 21A to 21B can take (B1 to B1).
B3) is an example. The bandpass means 21A to 21B are each configured by a bandpass filter, and each corresponds to all or part of the passbands B1 to B3. The pass band is not limited to three as shown in FIG. 6, but may be two or more. Also, similarly, the bandpass means 2
1A to 21B may be plural as shown in FIG. 1, and two cases are included as a typical example. The intermediate signal bi thus obtained is used for the automatic gain adjusters 30A to 3A.
It is amplified by 0B, and the gain is adjusted so that the output level becomes constant, and the detector input signal bo is obtained. Automatic gain adjuster 3
0A to 30B are, for example, an amplifier 301 having a gain G1 for amplifying the intermediate signal bi and outputting it as a detector input signal bo as shown in FIG. 2, and an envelope curve for obtaining the gain control voltage v1 of the amplifier 301 from this output signal bo. It is composed of a detector 302 and a bandpass filter 303. In FIG. 2, the envelope level of the intermediate signal bi input to the amplifier 301 is | Ai | ² , and the detector input signal b which is the output of the amplifier 301.
If the envelope level of o is | Ao | ² , the gain control voltage v1
Becomes v1 = | Ao | ² (3), and the gain G1 of the amplifier 301 becomes G1 = | Ao | ² / | Ai | ² (4). Here, the gains of the automatic gain adjusters 30A to 30B are G1A to G2B. Automatic gain adjusters 30A-30B
Then, the detector input signals bo whose gains have been adjusted to constant levels are input to the detectors 22A to 22B, respectively. The detectors 22A to 22B detect the detector input signal bo to obtain the detected signal ci. Detectors 22A-22B
For example, a delay detector 22 as shown in FIG. 3 is used.

The baseband waveform of the spread spectrum signal a has the same shape when the phases of the primary modulation signals are the same in each symbol section, and the positive and negative sides are inverted when the phases of the primary modulation signals are opposite. Has become. The detector input signals bo is the output of the band pass means 21A~21B is, bo (t) = Re [ δ m · q '(t) · expj (2πf c t)] (5) and expressed ([delta] _m = ± 1). Although the baseband waveform of the detector input signal bo is band-limited by the bandpass means 21A to 21B and has a waveform considerably different from the shape of the spread spectrum signal a (that is, the complex envelope in the equation (2)). The term of q representing the line is replaced with q ′ in the case of being band-limited in the equation (3)), and has substantially the same shape when the phases of the primary modulation signals are equal in each symbol section, If the phase of the primary modulation signal is opposite, the shape will be inverted positive and negative (as can be seen from equation (5), bo is δ
The sign of _m reverses the sign of the waveform). Strictly speaking, in the vicinity of the boundary with the adjacent symbol, the shape will not be exactly the same due to the influence of the adjacent symbol, and intersymbol interference will occur, but the band of the intermediate signal is compared with the symbol repetition frequency. If it is made large, the intersymbol interference is small, so that it does not become a problem.

In the differential detector 22 of FIG. 3, first, the symbol delay device 221 delays the detector input signal bo by the symbol period T to obtain the differential detector input signal bo _d . The spread modulation signal q is a repetitive waveform with a period T,
'When also noted that a repeating waveform of the period _{T, b od (t) =} b o (t-T) = Re [δ m-1 · q' approximately q (t) · expj (2πf c t ) .Expj (-2πf _c T)] (6). Now, by accurately setting T so as to satisfy expj (−2πf _c T) = 1 (7) or adjusting the phase of the output signal of the symbol delay unit 221, the delay detector input signal is eventually obtained. bo _{d is, b od (t) = Re} [δ m-1 · q '(t) · expj (2πf c t)] becomes (8). Of the multiplier 222, the low-pass frequency component extracted by the low-pass filter 223, that is, the detection signal c
i is the harmonic component e which is obtained by performing the multiplication of Equation (5) and Equation (8).
except the section components xpj (4πf _c t), by using equation (1), ci (t) = δ m · δ m-1 | q '(t) | 2 = d m | q' (t ) | ² (9) is obtained. From equation (9), it can be seen that the data is decoded by determining the polarity of the detection signal ci. That is, when there is no change in phase from the previous symbol, pulses with the same shape are multiplied to generate positive pulses, and when the phase is inverted from the previous symbol, pulses with inverted positive and negative shapes are multiplied. Produces a negative pulse. Therefore, the detection signal ci becomes a negative pulse and a positive pulse depending on whether or not the phase is inverted. The inverse gain adjuster uses this detection signal ci
Is adjusted in level by gains G2A to G2B which are inversely proportional to the gains G1A to G1B of the automatic gain adjusters 30A to 30B, and the gain correction detection signal co is output to the clock regenerators 25A to 25B and the combiner 32. Clock regenerator 25
A to 25B generate a reproduction symbol clock from the gain correction detection signal co, and the decoders 23A to 23B sequentially sample / identify the gain correction detection signal co by using the timing, and then code the identification point. Depending on the polarity of the above, it is determined to be 1 when it is positive and −1 when it is negative, and the determination data string d ′ _m is output. On the other hand, the combiner 32 outputs the combined detection signal c obtained by combining the plurality of gain correction detection signals co output by the inverse gain adjusters 31A to 31B to the decoder 23C and the clock regenerator 25c. The clock regenerator 25c generates a regenerated symbol clock from the combined detection signal c and uses the timing to generate a reconstructed symbol clock.
Outputs the determination data string d' _m similarly to the decoders 23A to 23B.

Although the case of two-phase phase modulation has been described here, the detection process is the same in the case of multi-level modulation such as four-phase and eight-phase modulation. The difference is that the delay detector 2
2. The configuration of the inverse gain adjuster 31 and the combiner 32 is two systems with orthogonal axes added, and the decoders 23A to 23B and 23
In C, the gain-corrected detection signal Co and the combined detection signal c are discriminated and determined to obtain a determination symbol data string, and then parallel-serial conversion is performed to output a decoded data string d ′ _m that is a bit string. is there. (Eg WRBenn
et, JR Davey, "Data Transmission," Lattes).

Now, the determination data string includes the data packets 61'-64 'of FIG. 5 having the same structure, which correspond to the data packets 61-64 of FIG. The unique word detectors 26A to 26B and 26C include a determination data string d' _m ,
The fixed pattern of the unique word is collated at any time, and if a match is detected, a frame signal is output. Based on the timing of this frame signal, the packet extractors 27A to 27B and 26C receive the information data 93 'and the error detection bit 9
The decoded data packet 95 'composed of 4'is extracted and delivered to the error detectors 29A to 29B and 29C. The error detectors 29A to 29B and 29C each detect a bit error in the decoded data packet 95 'based on the error detection bit 94', and pass the result to the decision selector 24, and at the same time, detect the bit error in the decoded data packet 95 '. Information data 93 '
It is also delivered to the judgment selector 24. Judgment selector 24
Is the information data 9 of the system in which no bit error was detected.
Only 3 ′ is selectively connected and output as the decoded data of the final output of the receiving device 20.

Therefore, as in the conventional example, when the interfering wave j shown in FIG. 6 is added to the transmission line, one or more of the band pass means 21A to 21B is passed through the pass band B1.
If set to, the intermediate signal b at the input of the detector of that system is not affected by the interfering wave j, and normal reception is performed. Therefore, in other systems, if reception is not normally performed and even if the error detector detects a bit error, as described above, if there is at least one system that can avoid the influence of the interference wave j,
The error detector of that system does not detect a bit error, and the decision selector 24 selects the information data 93 ′ of that system and outputs it as decoded data, so that normal reception is continued. Further, as one of the systems that can be selected by the judgment selector 24, there is also a system that obtains decoded data from the combined detection signal c obtained by combining a plurality of gain correction detection signals co, so if there is no interference, The reception sensitivity can be increased as compared with the conventional example.

The operation of obtaining decoded data from a combined detection signal obtained by combining a plurality of detection signals, which is a feature of the present invention, will be described in more detail below. In the following, as an example, it is assumed that the bandpass means in the embodiment of FIG. 1 has two systems, 21A and 21B.

The roles of the inverse gain adjusters 31A and 31B will be described. If the detection signals ci output from the detectors 22A and 22B are directly combined by the combiner 32 without performing the gain correction by the inverse gain adjusters 31A and 31B, the following problems occur. For example, consider a case where the intermediate signal biB, which is the output of the bandpass means 21B, becomes almost a noise component due to the level reduction due to multipath fading. In this case, even if the level of the intermediate signal biA of the bandpass means 21A is a level at which data can be received without error, the intermediate signals biA and biB are amplified to the same level by the automatic gain adjusters 30A and 30B, respectively, and detected. Since the signals are combined by the combiner 32 through the converters 22A and 22B, the detection signal ciB of the system of the pass band means 21B, which is only the amplified noise component, is added to the detection signal ciA of the system of the pass band means 21A. Therefore, the S / N ratio of the combined detection signal c combined by the combiner 32 deteriorates rather than the detection signal ciA. In the present invention, in order to avoid such a problem, the automatic gain adjuster 21A,
With the gain inversely proportional to the gain of 21B, the detected signals ciA, ciB
Inverse gain adjusters 31A and 31B for adjusting the level of are provided.

That is, the automatic gain adjusters 30A and 30B
Of the gain of G1 by the gain control signals v1A and v1B, respectively.
When A and G1B are obtained, the gain control signals v1A and v1B are used to set the gains G2A and G2B of the inverse gain adjusters 31A and 31B, respectively.
G2A = k / G1A (10) G2B = k / G1B (11), where k is a proportional constant, so that the combiner 32 causes the intermediate signals biA, b
It is possible to combine gain-corrected detection signals coA and coB that are equivalent to the detection signal obtained when a signal obtained by amplifying iB with equal gain is detected. Therefore, if there is no interfering wave,
Even if the reception signal of some systems becomes a noise component due to the level reduction due to fading, the above-mentioned problem does not occur, and if the signal waves of both systems are good, the combined detection signal c has a better S / N ratio than the detected signals ciA and ciB. Therefore, when there is no interference, the reception sensitivity is higher than that of the conventional receiving device 20 ′ that selectively uses only a part of the extracted band, and the transmission power can be reduced or the reach can be increased. Although the case where the number of reception systems is two has been shown in the embodiment of FIG. 1 so far, the above description is similarly applied to the case where there are three or more systems.

In the above description, the spread modulation signal q has a pseudo-random pulse waveform of a constant amplitude generated by a pseudo-random sequence, but the present invention is not limited to this, and other noise-like signals or those shown in FIG. Such a chirp signal may be used. The waveform in the detection process when the chirp signal is used is shown in FIG.

Although the period of the spread modulation signal q is equal to the symbol period T of the primary modulation signal p, it may be 1 / n (n is a natural number) of the symbol period T of the primary modulation signal p, or The delay detection may be performed by using the symbol period T of the modulation signal p times n (n is a natural number) and using the symbol delay device 221 having a delay time of n times the symbol period T.

Further, the bandpass means 21A-21B may be constituted by the bandpass filter 211, the frequency mixer 212 and the local oscillator 213 like the bandpass means 21 shown in FIG. In this case, the input signal is
After being converted by the frequency mixer 212 into a frequency band having a difference from the local oscillation signal which is the output of the local oscillator 213,
A part of the frequency components of the spread spectrum signal a, which has been band-limited by the band pass filter 211 and whose frequency has been converted, is extracted and output as the intermediate signal bi. The local oscillator 213 is usually composed of a PLL (Phase Locked Loop) synthesizer, and varies the frequency of the local oscillation signal at a frequency interval that is an integer multiple of the symbol rate 1 / T, or of the band pass means 21A to 21B. Local oscillator 213
Generates a local oscillation signal having a frequency different by this frequency interval. Equivalently, by changing the frequency of the local oscillation signal, the components of the original spread spectrum signal a having different frequency components can be extracted as the intermediate signal bi. When the center frequencies of the intermediate signals of the band pass means 21A to 21B are selected to be the same and the frequencies of the local oscillation signals are made different to obtain different pass bands, the band pass filters 211 and the detectors 22A to 22B are respectively obtained. Can use the same one, and the detector 22
A to 22B have an advantage that they can be easily realized because it is sufficient to guarantee the operation in a relatively narrow frequency range corresponding to the pass band.

Assuming that the frequency of the local oscillation signal is f _L , the equation (5) is as follows: b _o (t) = Re [δ _m · q ′ (t) · expj {2π (f _c −f _L ) t }] (5 ′), and the equation (6) is expressed as follows: b _od (t) = b _o (t−T) = Re [δ _m−1 · q ′ (t) · exp {2π (f _c −f _L ) t} · expj {-2π (f _c -f _L ) T}] (6 ′), and instead of expression (7), exp j {−2π (f _c −f _L ) T} = 1 ( By accurately determining T or adjusting the phase of the output signal of the symbol delay unit 221 so as to satisfy 7 ′), the equation (8) representing the differential detector input signal _bod is eventually expressed by _od (t) = Re [δ _m−1 · q ′ (t) · expj {2π (f _c −f _L ) t}] (8 ′). Similarly, in the multiplier 222, the low-pass frequency component extracted by the low-pass filter 223, that is, the detection signal ci, executes the multiplication of the formula (5 ′) and the formula (8 ′) to obtain the harmonic component. Excluding the component terms of expj {4π (f _c -f _L ) t}
It can be seen that the result of the equation (9) is obtained by using the equation (1), and similarly, the determination data is obtained by determining the polarity of the detection signal ci. Note that by modifying Equation (7 '), a k is an _{integer, f L = f c - k} × of (1 / T) (12) result is obtained, the frequency of the local oscillation signal, symbol rate The frequency interval must be an integral multiple of 1 / T.

FIG. 7 is a block diagram of a transmitting / receiving apparatus according to the second embodiment of the present invention. In this example,
The transmitter 10 is the transmitter 1 of the first embodiment shown in FIG.
0 has the same configuration and operation as that of 0, but the spread modulation signal q is shown in FIG.
The chirp waveform is as shown in. Further, the configuration and operation of each part of the receiving device 201 are almost the same as those of the receiving device 20 of the first embodiment, but the difference from the first embodiment is that the receiving device 201 in FIG. The difference lies in that delay compensators 33A to 33B for giving a predetermined compensation delay amount are connected to the detectors 22A to 22B. Since the other parts are exactly the same as the receiving device 20 in the first embodiment, the description of the operation is omitted and the delay unit 33 is omitted.
The second embodiment of the present invention will be described below, focusing on the operations A to 33B.

When the spread modulation signal q has a chirp waveform or the like as shown in FIG. 8, the bandpass means 2 is used as shown below.
Due to the pass band characteristics of 1A to 21B and the characteristics of the spread modulation signal q, the peak positions are deviated between the detection signals ciA to ciB. Therefore, it is desirable to perform timing correction before combining the detected signals. FIG. 9 illustrates this timing correction, and is a waveform diagram showing a detection process when the spread modulation signal q has a chirp waveform when the receiving device 201 has three types of pass bands. FIG. 9 shows intermediate signals bi1 to bi1 in accordance with the detection process corresponding to the pass bands B1 to B3 of the band passing means 21A to 21B, respectively.
bi3 and detected signals ci1 to ci3 are shown. The in-symbol waveform of the spread spectrum signal a is a chirp waveform, and in the example of FIG. 9, the first part of each symbol section is composed of low frequency components, and the higher the frequency components are toward the rear of each symbol section. Composed. Intermediate signal bi
Since 1 has a low frequency component extracted from the original spread spectrum signal a, the amplitude is large in the first half of the symbol section but small in the latter half. vice versa,
Since the intermediate signal bi3 is obtained by extracting a high frequency component from the original spread spectrum signal a, the amplitude is small in the first half of the symbol section and large in the latter half. Further, the intermediate signal 2 has a large amplitude at the center of the symbol section and a small amplitude at both ends. Detection signal ci1
Ci3 is a pulse train according to this amplitude change, and the peaks of the pulse have a shape located in the first half, the center, and the second half of the symbol section, respectively. This peak position is determined by the frequency sweep parameter of the spread modulation signal q and the characteristics of each bandpass means. Therefore, when the spread modulation signal q has the chirp waveform as shown in FIG. 8 as in the present embodiment, the timing correction (t23, t13 in FIG. 9) corresponding to the above shift of the peak position is performed.
Etc.) is preferably performed. Delay compensators 33A to 33
B is provided for performing this timing correction. In addition, in FIG. 7, the delay correctors 33A to 33A to
33B to the detector signals 22A to 22B and the inverse gain adjuster 3
Although it is inserted between 1A to 31B, the delay compensators 33A to 3B
The insertion position of 3B is not limited to this. For example, the delay compensators 33A to 3A are provided at the output terminals of the inverse gain adjusters 31A to 31B.
It is also possible to connect 3B. FIG. 10 is a block diagram of a transmitting / receiving apparatus according to the third embodiment of the present invention. In this embodiment, the transmitter 10 is the same as the transmitter 10 of the first embodiment shown in FIG. The configuration and operation of each unit of the receiving device 202 are almost the same as those of the receiving device 20 of the first embodiment, but the difference from the first embodiment is that the receiving device 202 in FIG. When 29A to 29B detect a bit error, the passband of the corresponding bandpass means 21A to 21B is changed, or the passband width is changed narrowly, or the passband is changed at the same time. The point where it changes narrowly is different. When the total band including the pass bands of the band pass means 21A to 21B is a part of the band of the spread spectrum signal a which is the transmission signal, the error detectors 29A to 29B determine whether there is interference, and When an interference is detected by detecting an error, the interference can be effectively avoided by changing the pass band of the corresponding band pass means to a band not used for reception. For example, there are many pass bands (three or more), and the band pass means 21A to 21B to the error detectors 29A to 29B.
Even if the receiving systems up to are only two systems allocated to two of these passbands, the probability that these two systems will be disturbed at the same time is low, and when either one is disturbed, By allocating the received system to the unused band, the hardware scale is not so large and efficient interference avoidance can be realized. The change of the pass band of the band pass means 21A to 21B shown in FIG. 10 is realized by, for example, switching and selecting a plurality of band pass filters. In that case, the respective bandpass means 21A to 21B
When a plurality of band pass filters are switched and selected, part or all of the band pass filters may be shared by some or all of the band pass means. Further, when the bandpass means 21A to 21B are equivalently realized as shown in FIG. 11, the local oscillator 213
May normally be configured by a PLL (PhaseLocked Loop) synthesizer, and the frequency may be changed at frequency intervals that are integer multiples of the symbol rate 1 / T according to the condition of Expression (12).

On the other hand, the larger the pass band width of the band pass means 21A to 21B, the larger the partial band of the spread spectrum signal a which is the transmission signal can be used.
The reception sensitivity is improved. However, on the other hand, the larger the pass bandwidth, the more often the disturbance is received. As in the present embodiment, when the presence or absence of interference is determined by the error detectors 29A to 29B and the interference is detected by detecting the bit error, if the pass band width of the corresponding band pass means is narrowly changed, the interference occurs. When there is no signal, the sensitivity is prioritized, and when there is interference, the interference resistance is prioritized, and it is possible to realize a receiving characteristic with excellent overall balance. It should be noted that the passband width once narrowed is restored by, for example, determining that the interference source has disappeared because a bit error has not been detected for a certain period of time, and restores the passband width. Also, FIG.
The change of the pass band of the band pass means 21A to 21B shown in 0 is realized by switching and selecting a plurality of band pass filters each having a different pass band width. In that case, when each of the band pass means 21A to 21B switches and selects a plurality of band pass filters, some or all of the band pass filters may be shared by some or all of the band pass means. Good. Also,
When the bandpass means 21A to 21B are equivalently realized as shown in FIG. 11, the passband of the bandpass filter 211 is realized by similarly switching and selecting a plurality of bandpass filters.

As described above, according to the above-described embodiment, since the signal components of a plurality of bands, which are partial within the band of the spread spectrum signal, are detected and selected at the same time, a strong interfering wave localized in the signal band is generated. In addition to having the ability to avoid, in a non-jamming environment, a plurality of detection signals are combined and decoded data is taken out, so that reduction in reception sensitivity can be reduced, transmission power can be reduced, or range can be increased. .

[0040]

As is clear from the above description, the present invention has the ability to avoid strong interfering waves localized in the signal band, and reduces the deterioration of the receiving sensitivity in an environment without interference. However, it has an advantage that the transmission power can be reduced or the reach can be increased.

[Brief description of drawings]

FIG. 1 is a block diagram of a data transmission / reception device according to a first embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration example of an automatic gain adjuster in the same embodiment.

FIG. 3 is a block diagram showing a configuration example of a wave detector in the data transmitting / receiving apparatus of the same embodiment.

FIG. 4 is a code configuration diagram of an example of a data packet in the embodiment.

FIG. 5 is an explanatory diagram of an example of a data packet observed in a determination data string in the data transmitting / receiving device of the same embodiment.

FIG. 6 is a schematic diagram of a spectrum of a conventional signal and a signal of the present invention.

FIG. 7 is a block diagram of a data transmission / reception device according to a second embodiment of the present invention.

FIG. 8 is a signal waveform diagram showing an example of a signal waveform of each part of the transmitter when the spread modulation signal is a chirp signal in the same embodiment and the conventional data transmitter and receiver.

FIG. 9 is a signal waveform diagram showing an example of the signal waveform of each part of the receiving device when the spread modulation signal is a chirp signal in the embodiment.

FIG. 10 is a block diagram of a data transmitter / receiver according to a third embodiment of the present invention.

FIG. 11 is a block diagram showing a configuration example of bandpass means in the first exemplary embodiment of the present invention.

FIG. 12 is a block diagram of a conventional data transceiver.

[Explanation of symbols]

10 transmitter 11 differential encoder 12 phase modulator 13 spreading modulation signal generator 14 spreading modulation multiplier 15 clock generator 16 packet assembler 17 primary modulator 20, 201, 202, 20 'receiving device 21, 21A -21B Band pass means 211 Band pass filter 212 Frequency mixer 213 Local oscillator 22A-22B Detector 22 Delay detector 221 Symbol delay device 222 Multiplier 223,303 Low pass filter 23A-23B, 23C Decoder 24 Judgment selector 25A to 25B, 25C Clock regenerator 26A to 26B, 26C Unique word detector 27A to 27B, 27C Packet extractor 29A to 29B Error detector 30A to 30B Automatic gain adjuster 301 Amplifier 302 Envelope detector 31A to 31B Inverse gain Adjuster 32 Synthesizer 33A-33B Delay compensator

Claims (10)

[Claims] 1. A transmission data is divided into a predetermined number of bits, at least a unique word and an error detection bit are added to form a data packet, and a primary modulation signal obtained by digitally modulating a carrier wave with the data packet is added to the primary modulation signal. In a data transmission / reception device, comprising: a transmission device that outputs a spread spectrum signal obtained by multiplying a spread modulation signal having a band wider than that of a modulation signal, and a reception device that demodulates the spread spectrum signal and outputs decoded data. The apparatus amplifies a plurality of band pass means for extracting only a signal component of a partial band within the band of the spread spectrum signal and a plurality of intermediate signals output from the band pass means to make the output level constant. A plurality of automatic gain adjusters for adjusting the gain and a plurality of automatic gain adjusters for detecting the output signals of the automatic gain adjusters, respectively. Number of detectors, a plurality of inverse gain adjusters for adjusting the gains of the plurality of detector signals output from the detectors, and the gain correction detector signal output from the inverse gain adjuster into one combined detector signal. A synthesizer for synthesizing, a plurality of clock regenerators that respectively generate a reproduction clock from the gain correction detection signal or the synthesis detection signal, and a determination data string from the gain correction detection signal or the synthesis detection signal and the reproduction clock, respectively. Outputting a plurality of decoders, a plurality of unique word detectors that respectively find the head of the decoded data packet by detecting the unique words from the plurality of determination data strings, and a frame that is an output of the unique word detectors A plurality of packet extractors for extracting the decoded data packets from the determination data string based on a signal, A plurality of error detectors for respectively detecting bit errors in the decoded data packet using error detection bits, and the decoded data from the decoded data packet determined to have no bit error by the error detector To get the
And the inverse gain adjuster adjusts the gain with a gain that is inversely proportional to the gain of the automatic gain adjuster in the corresponding band. 2. An automatic gain adjuster comprises an amplifier whose gain can be varied by a gain control signal, an envelope detector which obtains a gain control signal of the amplifier from an output signal of the amplifier, and a low pass filter. 2. The data transmitting / receiving apparatus according to claim 1, wherein the output signal of the amplifier is input to the detector, and the gain of the inverse gain adjuster is adjusted by using the gain control signal. 3. The digital modulation is differential phase modulation, the cycle of the spread modulation signal is an integral multiple or a fraction of the symbol cycle of the primary modulation signal, and the detector detects the intermediate signal and the intermediate signal. 2. The data transmitting / receiving apparatus according to claim 1, wherein the data transmitting / receiving apparatus is a delay detector that multiplies a delay signal delayed by an integer multiple of a symbol period of the primary modulation signal to obtain a detection signal. 4. The bandpass means comprises a frequency mixer, a local oscillator for supplying a local oscillation signal to the frequency mixer, and the frequency mixer converted into a frequency band of a difference between the frequencies of the local oscillation signal. A band pass filter for extracting only a signal component of a partial band of the output of the local oscillation signal, and changing the frequency of the local oscillation signal by an integer multiple of 1 of the symbol period, or the frequency of the local oscillation signal. 4. The data transmitting / receiving apparatus according to claim 3, wherein a plurality of the local oscillators are arranged at frequency intervals that are integer multiples of 1 of the symbol period. 5. The data transmitting / receiving apparatus according to claim 1, wherein the spread modulation signal is a chirp signal obtained by repeatedly sweeping the frequency of a sine wave in each cycle. 6. A receiver comprises a delay compensator for adjusting the time delay of each detection signal, and the peak position of the amplitude of each detection signal determined by the characteristics of said bandpass means and the characteristics of said spread modulation signal. The data transmitting / receiving apparatus according to claim 1, wherein the difference between the two is corrected. 7. The data transmitting / receiving apparatus according to claim 1, wherein the error detection bit is a CRC code. 8. The data transmitting / receiving apparatus according to claim 1, wherein the bandpass means, the automatic gain adjuster, the detector and the inverse gain adjuster are all two systems. 9. The data transmitting / receiving apparatus according to claim 1, wherein the bandpass means changes the passband of the bandpass means when the corresponding error detector detects a bit error. 10. The data transmitting / receiving apparatus according to claim 1, wherein the bandpass means narrows or narrows the passband width of the bandpass means when the corresponding error detector detects a bit error.