JPH0831378B2 - High frequency boost transformer - Google Patents

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a compact and thin high-frequency step-up transformer suitable for use in an inverter for lighting a cold cathode tube for illuminating the back surface of liquid crystal.

[0002]

2. Description of the Related Art A conventional high frequency boosting transformer for an inverter is constructed as shown in FIG. A low-voltage side primary winding L ₁ and a high-voltage side secondary winding L ₂ are separately wound by the collar 1 on a hollow bobbin 2 having a plurality of collars 1. The secondary winding L ₂ in which a high voltage is generated is further divided into a plurality of stages by the collar 1 and wound, so that the potential difference between the overlapping lines is reduced and dielectric breakdown is prevented. In the hollow portion 3 of the bobbin 2, the central legs 6, 7 of the two E-shaped cores 4, 5 are respectively provided.
Are inserted, and the core 4 and the core 5 are fixed to each other.

[0003]

A step-up transformer of this type needs to be thin so that it can be mounted in a narrow gap. For this reason, the bobbin 2 has a flat rectangular cross section, and even if the number of turns is the same, the winding length becomes long, the conductor resistance increases, and the efficiency decreases. The conventional rectangular transformer has a limit of about 5 mm in terms of thinning, and there is a problem that if the thickness is further compressed to widen the width, copper loss remarkably increases. In addition, since the secondary winding L ₂ has to be divided and wound, the winding work is complicated and the entire volume is increased.

[0004]

SUMMARY OF THE INVENTION It is an object of the present invention to provide a high-frequency step-up transformer having a small copper loss, a high efficiency and a low profile.

[0005]

SUMMARY OF THE INVENTION The present invention is a high frequency boosting transformer in which a primary winding and a secondary winding are wound around a cylindrical center leg of a core having an E-shaped cross section. Is wound in multiple layers on the outside of the primary winding around the center leg, and the number of turns per secondary winding is 1/10 or less of the total number of turns of the secondary winding, and the radius of the center leg is When Rm is the winding width of the primary winding and the secondary winding in the direction orthogonal to the central axis of the central leg, and W ₁ and W ₂ respectively, Rm / (W ₁ + W
> ₂ ) is in the range of 0.6 to 1.7.

[0006]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT An embodiment of the present invention will be described with reference to FIGS. The core 10 having an E-shaped cross section has a flat plate portion 11, outer legs 12 integrally formed at both ends thereof, and a cylindrical central leg 13 integrally formed at the center of the flat plate portion 11. 20
Is a plastic bobbin having two collars 21 and a hollow winding shaft 22 as shown in FIG. 4, and the central leg 13 of the core 10 is inserted into the hollow portion 23 of the winding shaft 22. A low-voltage side primary winding L ₁ is wound around the winding shaft 22, and a high-voltage side secondary winding L ₂ is provided outside the primary winding L ₁ via an insulating sheet 40.
It is wound in multiple layers in a direction orthogonal to the central axis of the central leg. On the core 10, a flat core 30 having an I-shaped cross section is butted and fixed. Although illustration is omitted, magnetic saturation can be prevented by inserting a plastic film or providing a gap between the core 30 and the central leg 13 or between the core 30 and the central leg 13 and the outer leg 12. It has a structure that makes it hard to get up. In addition, an insulating core 1 such as Ni-Zn system
When 0 and 30 are used, the bobbin 20 can be omitted.

As shown in FIG. 5, when the wire rod 8 is wound in multiple layers in the direction of the arrow, the winding start portion and the winding end portion of the next layer are adjacent to each other. However, even when the voltage applied across the secondary winding L ₂ is ₂ kV,
For example, if the secondary winding L ₂ is wound in 20 layers, the voltage applied between them is 2000/10, which is 200 V. Therefore, as in the conventional example in which the secondary winding L ₂ is divided into a plurality of stages with a collar, the overlapping wire materials are overlapped. The potential difference between the two becomes low and dielectric breakdown is prevented. Some high-frequency step-up transformers for inverters have a secondary winding L _{2 with} a small voltage of 1000 V or less, but one layer of the secondary winding L ₂ is required to surely prevent dielectric breakdown. It is preferable that the number of turns per hit is 1/10 or less of the total number of turns of the secondary winding L ₂ .

Next, let us consider conditions under which the required inductance L ₀ and saturation current I ₀ can be obtained in the inductance element. Now, assuming that the number of turns of the coil is N and the magnetic resistance is R, the inductance is given by N ² / R, so that R ≦ N ² / L ₀ ─────────── (1) If the cross-sectional area of the magnetic circuit having an area is S and the saturation magnetic flux density is Bm, the saturation current is given by BmSR / N, so R ≧ I ₀ N / BmS ─────────── (2) That is, The magnetic resistance R needs to be small in terms of the inductance L ₀ , and the magnetic resistance R needs to be large in terms of the saturation current I ₀ .

The number of turns N is plotted on the horizontal axis and the magnetic resistance R is plotted on the vertical axis.
FIG. 6 is a plot of the relationship between equations (1) and (2). In the figure, the shaded area indicates the range that satisfies the inequalities of the equations (1) and (2). As shown in FIG. 6, the minimum number of turns satisfying the required characteristics is N ₀ , and the minimum magnetic resistance is R ₀ . When the average magnetic path length is l and the magnetic permeability of the core is μ, the magnetic resistance R is generally given by R = 1 / μS. Therefore, if the cross-sectional area S is constant, the magnetic resistance R corresponds to the volume of the magnetic body. It is considered to be proportional. Therefore, N
_{It is considered that 0} and R ₀ are solutions that satisfy the required characteristics and minimize the volume of the inductance element. That is, N ₀ = L ₀ I ₀ / BmS ────── (3) R ₀ = L ₀ I ₀ ² / (BmS) ² ─── (4) is the optimum solution of the number of turns and the magnetic resistance.

Next, regarding the rectangular transformer as shown in FIG. 1 and the round transformer as shown in FIG. 3, the specifications of both transformers are defined as follows, and the respective characteristics are compared. Primary primary inductance 200μH or primary winding saturation current 1A above, the following withstand voltage 100 V _PP of the secondary winding ratio of 1:50 primary winding diameter 0.25mmΦ secondary winding diameter 0.07mmΦ winding

The relative permeability μ _r of the magnetic material is 3000, the saturation magnetic flux density Bm is 0.3 T, and the minimum cross-sectional area S of the magnetic circuit is 15
When N ₀ and R ₀ are calculated from the equations (3) and (4) as mm ² , N ₀ = L ₀ I ₀ / BmS = 200 × 10 ⁻⁶ × 1 / (0.3 × 15 × 10 ⁻⁶ ) = 45 (turns) R ₀ = L ₀ I ₀ ² / (BmS) ² = 200 × 10 ⁻⁶ × 1 ² /(0.3×15×10 ⁻⁶ ) ² = 9.88 (AT / Wb)

Therefore, since the primary winding has 45 turns and the winding ratio is 1:50, the secondary winding has 2250.
It will be a turn. When the rectangular and round transformers were designed under the above conditions, the following results were obtained.

[0013]

[Table 1] Comparison between square transformer and round transformer

As is clear from Table 1, when the same performance is realized with the same magnetic material, the round transformer shape is advantageous in terms of copper loss and volume. In particular, when used as an inverter transformer, a large amplitude resonance current of several tens of kHz flows in the primary winding, and therefore reduction of copper loss in the primary winding is a factor that can greatly contribute to improvement in conversion efficiency of the transformer.

FIG. 7 shows a core 10 and a core 30, and a primary winding L.
₁ is a diagram showing a side cross section of only the secondary winding L _2. FIG.
In such a transformer, the condition that the sum of the sectional areas of the central leg 13 and the primary winding L ₁ and the secondary winding L ₂ on the plane orthogonal to the central axis C of the central leg 13 is minimized, that is, in FIG. Next, the condition for minimizing the sum of the radius Rm of the center leg 13 and the winding width W ₁ of the primary winding and the winding width W ₂ of the secondary winding will be obtained. Now, let the radius of the central leg 13 be Rm, the turn ratio of the primary winding to the secondary winding be 1: n, the thickness of the flat plate portion 11 of the core 10 and the core 30 be t, and the depth of the bobbin be h, The wire diameter of the winding is d
₁ and the diameter of the wire of the secondary winding is d ₂ , the primary winding L
₁ turns per layer of the h / d _1, the number of turns per layer of the secondary winding L ₂ becomes h / d _2. Therefore, the number of layers to be wound on each side is N ₀ d ₁ / h on the primary side and nN ₀ d ₂ / h on the secondary side, and the respective winding widths W ₁ , W
₂ is W ₁ = N ₀ d ₁ ² / h ──────── (5) W ₂ = nN ₀ d ₂ ² / h ───────── (6)

The surface of the core extending from the side surface of the central leg 13 intersects with the flat plate portion 11, that is, a cylindrical portion having the same diameter as the central leg 13 and the same height as the thickness t of the flat plate portion 11, , I will call it a connecting part. In small and low profile transformers, the minimum cross-sectional area S of the magnetic circuit is generally limited by this joint of the core. In this case, since the area of the connecting portion of the core is equal to or smaller than the cross-sectional area of the central leg 13, 2πRmt ≦ πRm ² ∴Rm ≧ 2t ──────────────────────────────────── (7) Becomes S = 2πRmt ────────── (8).

Total cross-sectional area ΣS including the central leg 13 and the winding portion
Is ΣS = π (Rm + W ₁ + W ₂ ) ² ──── (9) if the distance between the primary winding and the secondary winding is ignored. Therefore, to minimize this, It is sufficient that P = Rm + W ₁ + W ₂ ────────────── (10) is minimized, with P in parentheses. Substituting W ₁ and W ₂ of the equations (5) and (6) into this equation, P = Rm + (d ₁ ² + nd ₂ ² ) / h × N ₀ Therefore, from the equation (3), P = Rm + (d ₁ ² + Nd ₂ ² ) / h × (L ₀ I ₀ / BmS) Substituting S in the equation (8), P = Rm + [(d ₁ ² + nd ₂ ² ) L ₀ I ₀ / 2πBmht] × 1 / Rm where K = (d ₁ ² + nd ₂ ² ) L ₀ I ₀ / 2πBmh
Letting t be P = Rm + K / Rm ───────── (11) The value Rm ₀ of the radius Rm of the central leg 13 that minimizes P is dP /
From dRm = 0, dP / dRm = 1-K / Rm ² (12) Therefore, Rm ₀ = K ^1/2 ──────────── (13). From equations (11) and (12), the minimum value of P is P = K ^1/2 + K / K ^1/2 = 2K ¹ / _2- (14) That is, Rm = K ^1/2 , W ₁ + W ₂ = K ^1/2 , that is, when Rm = W ₁ + W ₂ , P becomes minimum.

Now, assuming that W ₁ + W ₂ = K ^1/2 is fixed and Rm is X times this, Rm = X · K ^1/2 is substituted into the equation (11) to obtain P = X · K ^{1 / 2} + K / X · K ^1/2 = X · K ^1/2 + K ^1/2 / X ∴ P = K ^1/2 (X + 1 / X) ────── (15) Fig. 8 shows these P and X. It is the figure which plotted the relationship of. P
Becomes minimum when X = 1, and increases with a curve of X + 1 / X with respect to X. As a practical range of the value of P, from the minimum value to plus 15%, the value of X at this time, that is, the ratio of Rm and W ₁ + W ₂ is 0.6 to 1. The range is 7.

[0019]

According to the present invention, a high-frequency step-up transformer having a low profile and a small bottom area can be constructed, and the wire loss can be shortened to reduce copper loss. Further, since the secondary winding does not have to be divided into a plurality of stages, there is an effect that the winding process is simplified and the productivity is improved.

[Brief description of drawings]

FIG. 1 is an exploded perspective view of a conventional transformer

FIG. 2 is a front sectional view showing an embodiment of the transformer of the present invention.

FIG. 3 is a partially cutaway perspective view of the transformer main part .

FIG. 4 is a front sectional view of a bobbin in the transformer.

FIG. 5 is an explanatory view of a cross-section of a wire wound in multiple layers.

FIG. 6 is a diagram showing the relationship between the number of turns and the magnetic resistance.

FIG. 7 is a side sectional view showing the dimensional relationship between the core and the winding.

FIG. 8 is a diagram showing a relationship between X and P.

[Explanation of symbols]

10 core 13 central leg L ₁ primary winding L ₂ secondary winding

Claims (1)

[Claims] 1. A high-frequency step-up transformer in which a low-voltage side primary winding and a high-voltage side secondary winding are wound around a substantially cylindrical central leg of a core having an E-shaped cross section, and the secondary winding has a central leg. Centered around the outer side of the primary winding in multiple layers, the number of turns per secondary winding is less than 1/10 of the total number of turns of the secondary winding, and the radius of the center leg is Rm. The winding width of the primary winding and the secondary winding in the direction orthogonal to the center axis of the leg is W respectively.
_1, when the _{W 2, Rm / (W 1} + W 2) is 0.6
A high frequency boosting transformer characterized by being in the range of 1.7.