JPH08237958A - Power supply - Google Patents

Abstract

PURPOSE: To realize a power supply having simple circuitry using inexpensive components by reducing the voltage being applied to a switching element and a smoothing capacitor. CONSTITUTION: A series circuit of an inductor L1 and a switching element Q1 is connected with the output of a DC power supply E and a series circuit of a reverse blocking diode D1 and a smoothing capacitor C0 is connected in parallel with the switching element Q1 thus constituting a booster chopper circuit 1. A load circuit 2 including at least one set of inductor and capacitor is connected with at least one terminal of the switching element Q1 . A clip circuit 3 comprising the diode D. and the smoothing capacitor C0 limits the voltage across the switching element Q1 below the voltage of the smoothing capacitor C0 .

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power supply device for converting the output of a DC power supply into a high frequency voltage, for example, for use in lighting a discharge lamp by converting a DC power supply obtained by rectifying a commercial power supply into a high frequency. It is used.

[0002]

2. Description of the Related Art FIG. 30 is a circuit diagram of a conventional discharge lamp lighting device. The circuit configuration will be described below. The AC input terminal of the full-wave rectifier DB is connected to the AC power supply AC. The smoothing capacitor C ₀ is connected to the DC output terminal of the full-wave rectifier DB. This smoothing capacitor C ₀
A series circuit of an inductor L ₁ and a switching element Q ₁ is connected to the. At both ends of the switching element Q ₁ , a capacitor C ₂ is connected in parallel and a diode D ₁ is connected in antiparallel. The discharge lamp La is connected to both ends of the inductor L ₁ via the inductor L ₂ . A capacitor C ₃ is connected in parallel between the non-power supply side terminals of the filament of the discharge lamp La. The switching element Q ₁ is driven on / off at a high frequency by the control circuit 4. As a result, high frequency power is supplied to the discharge lamp La. In this conventional example, an AC power supply AC is rectified by a full-wave rectifier DB, smoothed by a smoothing capacitor C ₀ and converted into a DC voltage Vdc, which is converted into a high frequency by a one-stone voltage resonant inverter circuit, and then converted into a discharge lamp La. In this circuit configuration, since a smoothing capacitor C ₀ having a large capacity is used, a large inrush current flows when the power is turned on, and the reliability of the full-wave rectifier DB, the power switch, and the breaker is increased. There is a problem that it decreases, and there is a problem that the power factor is low.

Therefore, as shown in FIG. 31, an AC power source A
A rectifier DB for full-wave rectifying C, a closed circuit composed of an inductor L ₁ and a switching element Q ₁ connected to the output side of the rectifier DB, and a current limiting inductor L ₂ connected to both ends of the switching element Q ₁ . A series circuit including a discharge lamp La and a smoothing capacitor C ₀ , and a switching element Q
₁ and the control circuit 4 for controlling on and off the power supply device having the switching elements Q ₁ and diode D ₁ to flow a current in the opposite direction, the capacitor C ₂ constituting the oscillating circuit (JP-A-3-74091) Is proposed. In this conventional example, since the smoothing capacitor C ₀ is not connected to the rectifier DB, the inrush current when the power is turned on can be reduced, and the smoothing capacitor C ₀ is charged via the inverter circuit, so that the input The power factor can be improved.

Although it is not limited to the one-stone type inverter,
As shown in Japanese Patent Application Laid-Open No. 60-134776, there has been proposed a power supply device which also serves as a (step-up) chopper circuit and a part of an inverter circuit. That is, as shown in FIG.
The output terminal of the rectifier DB that rectifies the AC power supply AC is connected to both ends of the switching element Q ₁ via the inductor L ₁ , and the reverse current blocking diode D ₁ and the smoothing capacitor C ₀ are connected in series with the switching element Q _1. A circuit is connected, and both ends of the smoothing capacitor C ₀ are connected to the input ends of an inverter circuit that converts a DC input into an AC output by turning on / off the switching element Q ₁ . In this conventional example, the smoothing capacitor C _{0 serving} as the power supply of the inverter circuit is charged by the boost chopper circuit composed of the inductor L ₁ , the switching element Q ₁ , and the diode D _1, so that the input power factor can be improved.

[0005]

SUMMARY OF THE INVENTION A power supply device comprising a one-stone voltage resonance type inverter circuit of FIG.
A common problem of the power supply device of FIG. 31 proposed in No. 74091 is that the voltage applied to the switching element Q ₁ is higher than that of other inverter circuit systems such as a half bridge type inverter circuit. This will be described using the inverter circuit of FIG. In FIG. 30, the voltage Vce applied to the switching element Q ₁ is a damping vibration that vibrates sinusoidally with reference to the input voltage Vdc. Therefore, if the off section is maintained, as shown in FIG.
ce converges at the end. Therefore, for stable oscillation operation, the condition of Vp ₂ ≧ Vdc must be satisfied. When Vp ₂ <Vce, the element voltage V
Since ce does not return to 0 V, when an ON signal is input to the switching element Q ₁ , as shown in FIG. 34, a rush current flows at the moment when the element current Ic starts to flow and stress is applied to the switching element Q ₁ . . Therefore, when designing the inverter circuit of FIG. 30, the element voltage Vce is 0.
The component, operating frequency, and on-duty are determined so as to return to V. Here, the input voltage Vdc is Vdc = 100 × √2≈141 V when the commercial power supply AC is 100 V.
Becomes Also in the circuit of FIG. 31, the maximum value of the output voltage of the full-wave rectifier DB is about 141V, which is the same as the input voltage Vdc of FIG. In fact, the element voltage Vce of the power supply device of FIG. 30 commercialized using a fluorescent lamp as a load is 500 V or more even in the normal operation, and the switching element Q ₁ having a withstand voltage of 700 V or more is taken into consideration in consideration of the transient state when the power is turned on. It has been selected. Also in the circuit of FIG. 31, the maximum value of the input voltage Vdc is the same as that of the input voltage Vdc of FIG. 30, and therefore the switching element Q ₁ having a high breakdown voltage is required. However, the one-stone voltage-resonant inverter circuit has a simpler structure than other inverter circuits such as a half-bridge inverter circuit, and the switching element drive circuit floats (states away from the ground level). There is an advantage that it does not occur, and a circuit that does not require a high breakdown voltage switching element is desired.

On the other hand, in the power supply device of FIG. 32 shown in Japanese Patent Application Laid-Open No. 60-134776, a smoothing capacitor charged by the output of a boost chopper circuit composed of an inductor L ₁ , a switching element Q ₁ and a diode D _1. C ₀
Is used as the power source of the inverter circuit, the chopper voltage (smoothing capacitor C
There is a disadvantage in that the voltage (both ends of ₀ ) Vo changes depending on the load condition. In particular, in the load short-circuit mode of the inverter circuit, the chopper voltage Vo is abnormally boosted. Therefore,
In the case of a power supply device such as a discharge lamp to which a load having a non-linear characteristic is connected, a detection circuit for detecting the chopper voltage Vo is provided, and the switching element Q ₁ is turned on so that the chopper voltage Vo is lowered during abnormal boosting. Needs to be controlled off.

The present invention has been made in view of the above points, and it is an object of the present invention to reduce the voltage applied to the parts constituting the device, particularly the switching element and the smoothing capacitor. The purpose of the present invention is to realize a power supply device with a simple structure that can use inexpensive parts. Furthermore, when a rectified DC voltage is input to the AC power supply, the input current waveform is brought closer to the input voltage waveform to increase the input power factor with a simple configuration and improve the input harmonic distortion from the AC power supply. The purpose is to do.

[0008]

FIG. 1 is a basic circuit diagram of the present invention. The input power source E is, for example, a DC power source obtained by rectifying a commercial power source AC. The output of the DC power source E includes a first inductor L ₁ for current limiting and a switching element Q ₁
Are connected in series. Switching element Q
₁ is, for example, a bipolar transistor or MOSFET
Etc., and a series circuit of a backflow prevention diode D ₁ and a smoothing capacitor C ₀ is connected in parallel with the switching element Q ₁ . The load circuit 2 including at least one set of inductors and capacitors is connected to at least one terminal of the switching element Q ₁ . By controlling the switching element Q ₁ to be turned on / off by the control circuit 4, the inductor L ₁ , the switching element Q ₁ , and the diode D ₁ operate as the step-up chopper circuit 1 and the smoothing capacitor C ₀ is charged at a high frequency. It The clip circuit 3 composed of the diode D ₁ and the smoothing capacitor C ₀ limits the voltage across the switching element Q ₁ so that it does not become higher than the voltage of the smoothing capacitor C ₀ .

In addition, the inductor L ₁ and the switching element Q _{1 of} FIG. 1, the diode D ₁ and the smoothing capacitor C ₀ ,
The connection order of each component in the load circuit 2 is not limited to the illustrated order. Further, the load circuit 2 may have two or more inductors and capacitors, respectively. Further, in the figure, the load circuit 2
One end is to be connected to the switching element Q _1, the other end may not be connected to the switching element Q _1.

[0010]

Next, the operation of the present invention will be described with reference to the circuit diagram of FIG. The switching element Q ₁ performs on / off operation according to the on / off signal A from the control circuit 4. The flow of current when the switching element Q ₁ is turned on and off is shown in FIGS. 2 and 3. At the on-switching element Q ₁ is, the current Ia ₁ from the DC power source E via the inductor L ₁
Flows, and electromagnetic energy is stored in the inductor L ₁ . Meanwhile in the capacitor C ₁ discharging current Ia ₂ flows, B point of the capacitor C ₁ is negative, C point charges are accumulated such that the positive. As a result, inductor L
_The (oscillating) current Ia ₃ flows through the load Z through the load ₂ . During the off-switching element Q _2, the electromagnetic energy accumulated in inductor L _1, a charging current Ib ₁ flows through the capacitor C _1, the charge as point B of the capacitor C ₁ becomes positive, C point and negative accumulation To be done. as a result,
A (oscillating) current Ib ₃ flows through the inductor L ₂ to the load.

By the way, with respect to the inductor L ₁ and the switching element Q ₁ , the capacitor C ₁ becomes a voltage resonance element, and the switching element Q is charged by charging the capacitor C _1.
The voltage across ₁ is boosted. Switching element device voltage Vds of Q ₁ is the capacitor C ₀ voltage or more smoothing (to be precise, the relationship between the voltage Vo of the ON voltage Vf> smoothing capacitor C ₀ of the device voltage Vds + diode D ₁ of the switching element Q ₁ is satisfied ), The charging current I flows from the inductor L ₁ to the smoothing capacitor C ₀ via the diode D _1.
b ₂ flows.

By the way, the smoothing capacitor C₀The voltage of
Switching element Q₁On duty and input voltage
Is determined. For example, the power consumed by the load Z, smoothing
Capacitor C₀Voltage Vo is stable and input current
Operates in continuous mode in a typical chopper circuit
Assuming that₁Current I flowing through ₁
Is as shown in FIG. Input from DC power supply Eo
Voltage is Vi, smoothing capacitor C₀The voltage of Vo,
Touching element Q₁Inductor L when turning on₁of
Current I₁Value of Ib, switching element Q₁On On
One cycle of control is T, and the ON period is T₁, Off period
T₂, Switching element Q₁At the time of turning off
Kuta L₁Current I₁Is set to Ip. Switching element
Child Q₁Current I for turning on and off₁Switching
Element Q₁When is on, I₁= T × (Vi / L₁)
+ Ib, switching element Q₁When is off
Is I₁= Ip-tx (Vo-Vi) / L₁Becomes This
Where Ip = T₁× (Vi / L₁) + Ib, Ib = Ip
-T₂X (Vo-Vi) / L₁= T₁× (Vi / L₁)
+ Ib-T₂X (Vo-Vi) / L₁Is. Therefore,
T₁× (Vi / L₁) = T ₂X (Vo-Vi) / L₁When
Become. Therefore, Vi: (Vo-Vi) = T₂: T₁
The relationship is established. Where on-duty is 50%
In case of, T₁= T₂= T / 2, so Vi = V
From o-Vi, the relationship of Vo = 2Vi is established. DC power
Source Eo full-wave rectifies and smoothes commercial power (100V)
When DC voltage is used, Vo≈282V from Vi≈141V
Becomes

Since the voltage across the switching element Q ₁ is limited to the voltage across the smoothing capacitor C ₀ by the clip circuit composed of the diode D ₁ and the smoothing capacitor C ₀ , if the above assumption is satisfied, since the voltage applied to the switching element Q ₁ is about 282V, even taking into account the on-voltage and transient overvoltages diode D _1, clearly lower breakdown voltage than the switching element of Figure 30 shown in the prior art Switching elements can be used.

By the way, since the smoothing capacitor C ₀ only limits the element voltage Vds of the switching element Q ₁ and does not supply a DC voltage to the load Z, the smoothing capacitor C ₀ of the smoothing capacitor C ₀ changes depending on the state of the load Z. The voltage never fluctuates. Therefore, the abnormal boosting phenomenon at the time of load short circuit, which is a problem in a general boosting chopper circuit, does not occur.

[0015]

FIG. 5 shows a first embodiment of the present invention. In this embodiment, a discharge lamp La such as a fluorescent lamp is used as a load. As the DC power source, an output voltage obtained by full-wave rectifying the commercial power source AC by the full-wave rectifier DB is used. Current limiting inductor L ₁ , switching element Q ₁ (MOSFE
T), backflow prevention diode D ₁ , smoothing capacitor C
₀ is the same as the basic circuit of FIG. The load circuit is composed of a discharge lamp La, which is a load, a preheating capacitor C ₃ connected to the secondary side of the discharge lamp, and a series circuit of a set of inductance L ₂ and capacitor C ₁ . Control circuit IC
₃ is a general timer IC (for example, NEC Corporation)
ΜPC1555) of a monostable oscillator circuit. The trigger signal of the monostable oscillator circuit is a zero volt detection signal that detects when the drain voltage Vds of the switching element Q ₁ returns to 0V. The operation of the control circuit will be described with reference to the waveform chart of each part in FIG.

First, the switching element Q₁Drain of
When the voltage Vds returns to 0V, the resistance R₁And R₂Consists of
The voltage Va at the middle point a of the voltage dividing circuit is as shown in FIG.
And the resistance R₂And capacitor C_FourA little delayed due to the time constant of
Becomes 0V. Next, the negative logic circuit IC₁(The power terminal is
An input voltage Va (not shown) is a reference voltage (usually IC ₁
6) when the voltage drops below half of the power supply voltage of
As described above, the logic of the output voltage Vb at the point b is inverted and becomes High.
Level (abbreviated IC₁Power supply voltage). Then conden
SA C_FiveAnd resistance R_FourOutput of the differentiation circuit composed of
Voltage Vd rises suddenly as shown in FIG. 6 (d).
Then capacitor C_FiveAnd resistance R_FourTime constant set by
Return to 0V with. Negative logic circuit IC₂(Power terminals are shown
(Omitted) input voltage Vd is negative logic circuit IC₂Reference voltage of
Negative logic circuit IC only for the above period₂Output (point e)
The voltage Ve has a low level as shown in FIG.
It goes down to (0V). Timer circuit IC₃Trigger terminal
When the voltage Ve at point (e) falls, the resistance R_FiveAnd Conde
Sensor C₇The integration circuit composed of₇Electric power
The pressure (voltage Vf at point f) rises. As shown in Figure 6 (f)
As described above, the voltage Vf at point f is the timer circuit IC₃Power supply voltage
When the voltage rises to 2/3 of Vcc, the capacitor C₇
Discharges and falls sharply. During the charging period of this f point,
Timer circuit IC₃Output terminal (3rd pin) is High
Level (IC₃Power supply voltage), as shown in FIG.
So that the resistance R₃The potential at point g is increased via
Itching element Q₁Turn on. In addition, switching
Element Q₁If the drive capacity of the
₃Output and switching element Q₁Driver circuit between
Are required, but are omitted in FIG. Also the timer
-Circuit IC₃Control terminal (pin 5) is a timer
-Circuit IC₃Internal resistance and capacitor C₆To the integration circuit of
The timer circuit IC rises with a predetermined time constant₃Is
Oscillation is started by High level logic, but here, c
As shown in Fig. 6 (c), the potential at the point is already high.
I explained that it is Bell logic. Then thailand
Mar Circuit IC ₃When the output of drops to the Low level,
Touching element Q₁Turns off and the element voltage Vds rises.
Go up. Divided voltage of element voltage Vds (voltage Va at point a)
Is a resistance R as shown in FIG.₁And capacitor C_Four
Rises with a time constant determined by₁Out of
The voltage Vb at the point b of force is inverted as shown in FIG.
Constant logic IC₂Input (point d) potential Vd is shown in FIG.
It becomes a negative potential as shown in (d), but it is a negative logic
Road IC₂Since it is less than the reference voltage of, the timer circuit IC
₃The trigger voltage (potential Ve at point e) of is shown in Fig. 6 (e).
Does not change. Therefore, the timer circuit IC₃
Output voltage does not change. That is, the switching element Q
₁Input (g point) potential changes as shown in Fig. 6 (g)
do not do. That is, the switching element Q₁On period
Is the resistance R_FiveAnd capacitor C₇Freely set by the integration circuit of
However, the off period is determined by the main circuit (other than the control circuit).
It

In this embodiment, since the switching element Q ₁ does not turn on unless the element voltage Vds of the switching element Q ₁ returns to 0 V as described above, the element voltage V
A capacitor C ₂ is added as a voltage resonance element for the inductor L ₁ and the switching element Q ₁ so that ds returns to 0V. As described in the explanation of the operation, the capacitor C ₁ also operates as a voltage resonance element, but the inductor L ₂
Element voltage Vds due to the influence of the discharge lamp FL and the capacitor C _3.
Does not always return to 0V. Further, LC resonance of the load circuit is performed by the inductor L ₂ and the capacitor C ₃ , and the capacitor C ₁ shapes the lamp current waveform, so that the capacitance is made sufficiently large so as not to be related to resonance.

Switching elements Q are shown in FIGS.
_The waveforms of the drain voltage Vds of ₁ , the drain current Id, the gate voltage Vgs, and the lamp current I are shown. Gate voltage V
When gs becomes High level and the switching element Q ₁ is turned on, the inductor L ₁ is output from the output of the full-wave rectifier DB.
A drain current flows through the switching element Q ₁ via the. This corresponds to the current Ia ₁ in FIG. At this time, at the same time, the discharge current of the capacitor C ₁ is changed to the switching element Q ₁
Flows to This corresponds to the current Ia ₂ in FIG. The combined current of these becomes the drain current Id. At this time, a charging current flows through the capacitor C ₁ via the inductor L ₂ . This corresponds to the current Ia ₃ in FIG. At this time, the lamp current I becomes an oscillating current due to the resonance circuit of the inductor L ₂ and the capacitor C ₃ . The gate voltage Vgs becomes Low level, the switching element Q ₁ is turned off, the charging current flows into the capacitor C ₁ through the inductor L ₁ from the full-wave rectifier DB by electromagnetic energy accumulated in inductor L _1. This corresponds to the current Ib ₁ in FIG. At this time, the charging current also flows through the capacitor C ₂ at the same time, and the element voltage Vds of the switching element Q ₁ rises. When the element voltage Vds becomes equal to or higher than the voltage of the smoothing capacitor C ₀ , the smoothing capacitor C is passed through the diode D _1.
Charge current flows to ₀ . This corresponds to the current Ib ₂ in FIG. As a result, the element voltage Vds is equal to the smoothing capacitor C.
Clipped at ₀ voltage. At this time, the capacitor C ₁
Discharge current flows through the inductor L ₂ . This corresponds to the current Ib ₃ in FIG. The lamp current Iz becomes an oscillating current due to the resonance circuit of the inductor L ₂ and the capacitor C ₃ .

In this embodiment, the switching element Q₁of
If the element voltage Vds does not return to 0V, the switching element
Q₁Does not turn on, the switching element
Child Q ₁There is no rush current flowing in. Accordingly
Switching element Q₁Improves reliability and rush
Unnecessary radiation noise generated from the current can be reduced
It Furthermore, the waveforms of the element voltage Vds and the drain current Id
The overlapping part of the
It can be reduced. Also, inductor L₁And switching element
Q₁, Capacitor C₂Resonance of voltage resonant circuit composed of
The peak potential Vp of FIG.
Actually switching element Q₁The voltage applied to is smooth
Capacitor C₀Element voltage Vd
The resonant circuit can be designed to ensure that s returns to 0V. You
That is, the degree of freedom in design can be increased.

Furthermore, by setting the drive frequency of the switching element Q ₁ sufficiently higher than the commercial power supply frequency, the peak of the input current is generated by the boost chopper circuit composed of the inductor L ₁ , the switching element Q ₁ and the diode D _1. The value waveform can be similar to the input voltage waveform. Therefore, if a low-pass filter is added to the input of the full-wave rectifier DB, the input current waveform itself can be approximated to the input voltage waveform by cutting the high frequency component of the input current. As a result, the input power factor can be increased and the harmonic distortion characteristics can be improved.

A circuit diagram of the second embodiment of the present invention is shown in FIG. The difference from the first embodiment is that the capacitor C ₂ which is a voltage resonance element is connected in parallel to the output of the full-wave rectifier DB so as to form a closed circuit together with the inductor L ₁ and the switching element Q ₁ . Operation and configuration of other parts is first
The description is omitted because it is the same as the embodiment. Further, the control circuit 4 only needs to be able to control ON / OFF of the switching element Q ₁ at a constant cycle, and therefore the illustration of the internal configuration is omitted. Of course, it may have a zero-volt detection function as in the first embodiment. Also in the following embodiments, like the second embodiment, only the points different from the first embodiment will be described. Further, the detailed description of the internal configuration of the control circuit 4 is omitted.

A circuit diagram of the third embodiment of the present invention is shown in FIG.
You In this embodiment, the capacitor C₂Diode D₁When
In parallel, that is, one end of inductor L₁And switching
Element Q ₁Is connected to the connection point and the other end is a smoothing capacitor
C₀It is connected to the. Smoothing capacitor C₀Is relatively
Smoothing capacitor C due to its large capacity₀Voltage is almost
stable. Therefore, the capacitor C₂Is voltage resonance
It becomes an element.

FIG. 10 is a circuit diagram of the fourth embodiment of the present invention. In this embodiment, the capacitor C ₂ is connected in parallel with the inductor L ₁ .

FIG. 11 is a circuit diagram of the fifth embodiment of the present invention. In this embodiment, the capacitor C ₂ is connected between the connection point of the capacitor C ₁ and the inductor L ₂ and the ground, and the combined capacitance of the capacitors C ₁ and C ₂ is the inductor L _1.
And becomes a voltage resonance element of the switching element Q ₁ .

FIG. 12 is a circuit diagram of the sixth embodiment of the present invention. In this embodiment, similarly to the fifth embodiment, the capacitor C
₂ is connected between the connection point of the capacitor C ₁ and the inductor L ₂ and the ground, and the arrangements of the capacitor C ₁ and the inductor L ₂ are exchanged.

FIG. 13 is a circuit diagram of the seventh embodiment of the present invention. In the present embodiment, the current limiting inductor L ₁ is connected to the low voltage side of the full wave rectifier DB. If the direction of the current flowing through the inductor L ₁ is considered to be opposite, it is obvious that the same operation as in the first embodiment is performed.

FIG. 14 is a circuit diagram of an eighth embodiment of the present invention. In this embodiment, the backflow prevention diode D ₁ is connected between the smoothing capacitor C ₀ and the ground. Switching element Q ₁ , smoothing capacitor C ₀ , diode D
_{Since 1} constitutes a closed circuit as in the first embodiment, the operation is the same as in the first embodiment.

FIG. 15 is a circuit diagram of the ninth embodiment of the present invention. This embodiment is a combination of the seventh embodiment and the eighth embodiment, and the arrangement of the diode D ₁ and the inductor L ₁ is different. That is, the current limiting inductor L ₁ is connected to the low-voltage side of the full-wave rectifier DB, and the backflow prevention diode D ₁ is connected between the smoothing capacitor C ₀ and the ground.

FIG. 16 is a circuit diagram of the ninth embodiment of the present invention. In this embodiment, one terminal is connected to the connection point of the switching element Q ₁ and the inductor L ₁ , and the other terminal of the load circuit is connected to the connection point of the diode D ₁ and the smoothing capacitor C ₀ .

FIG. 17 is a circuit diagram of the eleventh embodiment of the present invention. In this embodiment, the arrangement of the inductor L ₁ is changed from that of the tenth embodiment, and the inductor L ₁ is connected between the switching element Q ₁ and the low voltage side output of the full-wave rectifier DB. Further, the other terminal of the load circuit having one terminal connected to the connection point of the switching element Q ₁ and the diode D ₁ has the diode D ₁ and the smoothing capacitor C _0.
Is connected to the connection point of.

FIG. 18 is a circuit diagram of the twelfth embodiment of the present invention.
is there. In this embodiment, the inductor L ₂And capacitor C₁
The arrangement of is replaced. In addition, inductor L₂And con
Densa C₁Not only the discharge lamp La as a load is included
The current that flows even if the order of connecting the series circuits is changed
The behavior is the same because it does not change.

FIG. 19 is a circuit diagram of the thirteenth embodiment of the present invention.
is there. In this embodiment, the inductor L ₁And switching element
Child Q₁Other load circuits with one terminal connected to the connection point
One terminal is full-wave rectifier DB and inductor L₁At the connection point
Connected. Switching element Q₁The operating frequency of
If the frequency is sufficiently higher than the power supply frequency for the
It is considered that the output voltage of the full-wave rectifier DB is almost stable.
Therefore, the difference in operation from the first embodiment can be ignored.
It In addition, inductor L₂And capacitor C₁Reverse the placement of
I have to.

FIG. 20 is a circuit diagram of the fourteenth embodiment of the present invention.
is there. In this embodiment, the inductor L ₁And switching element
Child Q₁Other load circuits with one terminal connected to the connection point
One terminal is a smoothing capacitor C₀Connected in parallel to
Two capacitors C₁And C _FourConnected to the connection point of the series circuit of
It continues. The capacitor C₁And C_FourIndak
L₂And capacitor C₃Big enough not to affect the resonance of
Use a capacitor of capacity.

FIG. 21 is a circuit diagram of the fifteenth embodiment of the present invention. The present embodiment is an example in which the second smoothing capacitor C ₄ is added to the output of the full-wave rectifier DB.
C is full-wave rectified by the full-wave rectifier DB and smoothed by the capacitor C ₄ . This circuit can be considered as a circuit in which a capacitor C ₂ is added to both ends of the switching element Q ₁ in the basic circuit of FIG.

FIG. 22 is a circuit diagram of the 16th embodiment of the present invention. In this embodiment, a series circuit of diodes D ₂ and D ₃ is connected to the output of the full-wave rectifier DB, and a relatively high voltage is provided between the connection point of the diodes D ₂ and D ₃ and one end of the commercial power supply AC. The capacitor C ₄ having a large capacity is added. Since the output of the full-wave rectifier DB is a pulsating voltage, the input voltage of the voltage resonant circuit of the inductor L ₁ , the switching element Q ₁ , and the capacitor C ₂ is too low during the period of the valley portion, and the drain of the switching element Q ₁ is too low. The voltage does not reach the voltage of the smoothing capacitor C ₀ . Also, the capacitor C
It is conceivable that the lamp current is temporarily stopped due to insufficient charging current of ₁ and so on. In order to eliminate the rest period, the capacitor C ₄ is charged in the peak portion of the voltage of the commercial power source AC, that is, in the period when the output voltage of the full-wave rectifier DB is high, and the valley portion of the voltage of the commercial power source AC, that is, the full-wave rectifier D.
During the period when the output voltage of B is low, the discharge current of the capacitor C ₄ is supplied as the input current of the circuit. In addition, full-wave rectifier DB
The combined impedance when viewing the load side when the Z from, since the use of the output voltage and the phase of only the full-wave rectifier DB time constant determined by the capacitor C ₄ and the impedance Z is shifted, the capacitor C ₄ is comparative Unless the capacitance is relatively large, the valley of the output voltage of the full-wave rectifier DB cannot be filled.

FIG. 23 is a circuit diagram of the seventeenth embodiment of the present invention. This embodiment also improves the quiescent period of the lamp current. A series circuit of resistors in the smoothing capacitor C ₀ R ₁ and smoothing capacitor C ₄ connected in parallel, the current from the connecting point of the resistors R ₁ and smoothing capacitor C ₄ via the diode D ₂ to the full-wave rectifier DB To supply. The capacitor C ₄ becomes a constant voltage by being charged from the smoothing capacitor C ₀ via the resistor R ₁ . When the output voltage of the full-wave rectifier DB is higher than the voltage of the capacitor C ₄ , current is supplied from the full-wave rectifier DB to the inductor L ₁ and the capacitor C ₄ is supplied.
_When the voltage of ₄ is higher than the output of the full-wave rectifier DB, the current is supplied from the capacitor C ₄ to the inductor L ₁ .
Therefore, the valley portion of the output voltage of the full-wave rectifier DB can be filled, and the idle period of the lamp current can be eliminated.

[0037] In order to stabilize the voltage of the capacitor C _4, may be connected in parallel to the resistor and the constant voltage diode and the capacitor C _4. Further, a choke coil may be connected instead of the resistor R ₁ , and in short, it is sufficient if a direct current can be passed. A control power supply circuit for supplying power to the control circuit may be configured with the same configuration as the series circuit of the resistor R ₁ and the capacitor C ₄ .

FIG. 24 is a circuit diagram of the eighteenth embodiment of the present invention. This embodiment also improves the idle period of the discharge lamp La. Capacitors C ₄ and C at the output of full-wave rectifier DB
₅ and a general partial smoothing circuit consisting of diodes D ₂ , D ₃ , and D ₄ are added to smooth the valley of the output voltage of the full-wave rectifier DB at half the maximum voltage, thereby reducing the lamp current It eliminates the rest period.

FIG. 25 is a circuit diagram of the nineteenth embodiment of the present invention. In this embodiment, resistances R ₁ and R ₂ are added to the partial smoothing circuit in the 18th embodiment to improve the stability of the circuit.

FIG. 26 is a circuit diagram of the twentieth embodiment of the present invention.
is there. In this embodiment, the inductor L ₃For smoothing
Capacitor C₀Inductor L₃And switch
Element Q₁, And capacitor C₂Voltage resonance times consisting of
By configuring the path, the output voltage of the full-wave rectifier DB
Switching element Q in the valley₁Element voltage Vds of
Boosting the lamp current to improve the rest period of the lamp current.
is there.

FIG. 27 is a circuit diagram of the twenty-first embodiment of the present invention. In the present embodiment, the inductor L ₂ of the first embodiment is replaced with a transformer T, and the discharge lamp La as a load is connected to the secondary side of the transformer T, whereby the degree of freedom of the load voltage is increased.

FIG. 28 is a circuit diagram of the 22nd embodiment of the present invention. In this embodiment, the smoothing capacitor C ₀ is inserted between the switching element Q ₁ and the closed circuit of the load circuit. The position of the diode D ₁ is connected to the ground side of the full-wave rectifier DB.

FIG. 29 is a circuit diagram of the 23rd embodiment of the present invention.
is there. In this embodiment, the inductor L ₁And L₂Combined with
Things.

Although the load in each of the first to twenty-third embodiments is the discharge lamp La for convenience, the load of the present invention is not limited to the discharge lamp.

[0045]

According to the present invention, the stress applied to the switching element can be reduced by clipping the voltage applied to the switching element with the voltage across the smoothing capacitor. Further, even when the load is short-circuited, the smoothing capacitor does not have abnormal boosting. Therefore, the reliability of the component can be improved. In other words, an inexpensive and small-sized device can be realized by using a component having a low breakdown voltage. Further, in the case where the DC power source has a configuration in which the commercial power source is rectified, the input current waveform can be approximated to the input voltage waveform, the input power factor can be increased, and the harmonic distortion characteristic can be improved.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a basic configuration of the present invention.

FIG. 2 is a circuit diagram showing an operation when the switching element of the present invention is on.

FIG. 3 is a circuit diagram showing an operation of the switching element of the present invention when it is off.

FIG. 4 is a current waveform diagram according to the on / off operation of the switching element of the present invention.

FIG. 5 is a circuit diagram of a first embodiment of the present invention.

FIG. 6 is a waveform diagram showing the operation of the control circuit according to the first embodiment of the present invention.

FIG. 7 is a waveform chart showing the operation of the main circuit of the first embodiment of the present invention.

FIG. 8 is a circuit diagram of a second embodiment of the present invention.

FIG. 9 is a circuit diagram of a third embodiment of the present invention.

FIG. 10 is a circuit diagram of a fourth embodiment of the present invention.

FIG. 11 is a circuit diagram of a fifth embodiment of the present invention.

FIG. 12 is a circuit diagram of a sixth embodiment of the present invention.

FIG. 13 is a circuit diagram of a seventh embodiment of the present invention.

FIG. 14 is a circuit diagram of an eighth embodiment of the present invention.

FIG. 15 is a circuit diagram of a ninth embodiment of the present invention.

FIG. 16 is a circuit diagram of a tenth embodiment of the present invention.

FIG. 17 is a circuit diagram of an eleventh embodiment of the present invention.

FIG. 18 is a circuit diagram of a twelfth embodiment of the present invention.

FIG. 19 is a circuit diagram of a thirteenth embodiment of the present invention.

FIG. 20 is a circuit diagram of a fourteenth embodiment of the present invention.

FIG. 21 is a circuit diagram of a fifteenth embodiment of the present invention.

FIG. 22 is a circuit diagram of a sixteenth embodiment of the present invention.

FIG. 23 is a circuit diagram of a seventeenth embodiment of the present invention.

FIG. 24 is a circuit diagram of an eighteenth embodiment of the present invention.

FIG. 25 is a circuit diagram of a nineteenth embodiment of the present invention.

FIG. 26 is a circuit diagram of a twentieth embodiment of the present invention.

FIG. 27 is a circuit diagram of a twenty-first embodiment of the present invention.

FIG. 28 is a circuit diagram of a twenty-second embodiment of the present invention.

FIG. 29 is a circuit diagram of a twenty-third embodiment of the present invention.

FIG. 30 is a circuit diagram of a first conventional example.

FIG. 31 is a circuit diagram of a second conventional example.

FIG. 32 is a circuit diagram of a third conventional example.

FIG. 33 is a waveform chart of the element voltage of the first conventional example.

FIG. 34 is a waveform diagram showing an inrush current at the time of switching in the first conventional example.

[Explanation of symbols]

E DC power supply C ₀ Smoothing capacitor D ₁ Reverse current blocking diode Q ₁ Switching element L ₁ Inductor Z Load 1 Boost chopper circuit 2 Load circuit 3 Clip circuit 4 Control circuit

Claims (10)

[Claims] 1. A power supply device having a DC power supply and an inverter circuit which includes at least one switching element and which converts a DC input into an AC output by an ON / OFF operation of the switching element, and is connected to an output of the DC power supply. A series circuit of a current limiting inductance and a switching element, a series circuit of a backflow prevention diode and a smoothing capacitor connected in parallel with the switching element,
The load circuit is connected to at least one terminal of the switching element, and includes a load circuit including at least one set of inductance and a capacitor, and a control circuit that controls on / off of the switching element. The current limiting inductance and the switching element, the backflow prevention diode operate as a step-up chopper circuit to charge the smoothing capacitor at high frequency, and the switching circuit is formed by the clip circuit composed of the backflow prevention diode and the smoothing capacitor. The power supply device is configured so as to prevent the voltage across both terminals from becoming higher than the charging voltage of the smoothing capacitor. 2. The power supply device according to claim 1, wherein a current limiting inductance and a resonance capacitor for performing voltage resonance operation are added. 3. The power supply device according to claim 1, wherein the control circuit is configured to detect that the element voltage of the switching element has dropped to 0V and turn on the switching element. 4. The power supply device according to claim 1, wherein the smoothing capacitor forms a closed circuit with the load circuit and the switching element. 5. The power supply device according to claim 4, wherein the load circuit and the switching element form a closed circuit. 6. The power supply device according to claim 2, wherein the resonance capacitor is connected in parallel with the switching element. 7. The power supply device according to claim 1, further comprising means for supplying electric power from the smoothing capacitor to the control power supply of the control circuit. 8. The DC power supply is a pulsating current power supply obtained by rectifying AC with a rectifier, a low-pass filter is formed on the input side of the rectifier, and the ON / OFF control frequency of the switching element is higher than the AC frequency. The power supply device according to claim 1, wherein the power supply device is a power supply device. 9. The power supply device according to claim 8, further comprising means for partially smoothing the output of the rectifier. 10. The power supply device according to claim 1, wherein the load circuit is connected in parallel with the current limiting inductance.