JPH0744843B2 - Inverter pulse width modulation controller - Google Patents

Description

Description: TECHNICAL FIELD The present invention relates to a pulse width modulation control device for an inverter, and more particularly to an improvement of a device for increasing a carrier frequency to perform precise waveform control.

(Prior Art) In recent years, elements such as MOSFETs (metal oxide gate field effect transistors) have appeared as high-speed switching devices. If these are used for pulse width modulation control of inverters, precise waveform control becomes possible. As a result, it has become possible to obtain effects such as reduction of electromagnetic noise and increase of motor efficiency.

Therefore, conventionally, by providing an analog control circuit, or using a dedicated hardware for a digital circuit or a high-speed arithmetic unit such as DSP, it is possible to perform pulse width modulation control at a high carrier frequency (for example, 20 KHz), and the electromagnetic noise described above is generated. It is known to secure the effect of reducing the above. (For example, see "Application of high-frequency switching to general-purpose inverters", Proceedings of the National Conference of the Institute of Electrical Engineers of Japan, 1987, Presenter, Chihiro Okado, etc.).

(Problems to be Solved by the Invention) However, the above-mentioned conventional ones have drawbacks that the circuit is complicated, various adjustments are complicated, and the cost is high.

Therefore, it is conceivable to adopt a one-chip microcomputer (hereinafter abbreviated as a microcomputer) that is inexpensive and has a simple circuit configuration. In this idea, the microcomputer is used for a series of processes necessary for generating the PWM control pattern. Takes a long time, for example, about 200 μS, and the carrier frequency remains at about 5 KHz at the maximum. Therefore, it is generally difficult to control the pulse width modulation with a high frequency (20 KHz or more) carrier frequency.

The present invention has been made in view of the above points, and an object thereof is to make a situation equivalent to a higher carrier frequency by using a one-chip microcomputer.
The purpose is to enable pulse width modulation control at an equivalently high carrier frequency with a low-priced and simple circuit configuration, and obtain effects such as reduction of electromagnetic noise and increase of motor efficiency by precise waveform control.

(Means for Solving the Problems) In order to achieve the above object, in the present invention, the algorithm for generating the PWM control pattern (that is, the ON time of a plurality of switching elements provided in the inverter) is changed to calculate the PWM control pattern. Even if the time (calculation cycle) is long, the calculated ON time of each switching element is divided into a plurality of pulses to raise the carrier frequency equivalently.

In that case, when equally dividing the ON time of each switching element into pulses of equal width, as shown in FIG.
When the ON time greatly changes, it does not respond well to this, and the good reproducibility of the signal wave is slightly reduced. On the other hand, when dividing the ON time of each switching element into multiple pulses of unequal width, as shown in (b) in the figure, even if the ON time is small, even division can respond well and the signal wave can be reproduced well. However, the calculation and processing time is lengthened by the division into unequal widths, and as a result, the effect of increasing the carrier frequency is reduced accordingly.

Therefore, in the present invention, the mode of division of the ON time is not fixed, and it is possible to appropriately switch between division into equal-width pulses and division into non-uniform-width pulses, thereby ensuring good reproducibility of the signal wave. However, the calculation and processing time required to divide the ON time of the switching element is shortened as much as possible to perform pulse width modulation control with a sufficiently high carrier frequency.

As a concrete solution, a bridge circuit (4) connected to a three-phase winding (2) and having a plurality of switching elements (Tra) to (Trc ') is provided as shown in FIGS. The bridge circuit (4) is provided with each switching element (Tr
a) It is premised on a pulse width modulation control device for an inverter in which a DC pulse width modulation is performed by ON / OFF operations of (Trc ') and a three phase AC voltage is applied to the three phase winding (2). . Then, as shown in FIG. 6, FIG. 7, FIG. 11 and FIG. 15, a calculation means (for calculating the ON time of each of the switching elements (Tra) to (Trc ′) at a calculation cycle corresponding to the carrier frequency ( 10) and the ON time of each of the switching elements (Tra) to (Trc ') calculated by the calculation means (10), according to the rate of change of the ON time, the total pulse width during the ON time and at the same time. A plurality of equal-width pulses, or the pulse width is gradually changed based on the previous ON time and the current ON time of the switching elements (Tra) to (Trc ') calculated by the calculation means (10). Divided into one of a plurality of non-uniform width pulses,
The period of the divided plural equal-width pulses or non-uniform-width pulses is set to ON of the switching elements (Tra) to (Trc ′).
Each of the switching elements (Tra) to (Trc ') is composed of a dividing means (11) for making the number of times divided by the dividing number of time and a plurality of pulses divided by the dividing means (11).
And a control means (12) for controlling ON.

(Operation) With the above configuration, in the present invention, even when the carrier frequency is a normal value (for example, about 5 KHz), the ON time (PWM control pattern) of each switching element (Tra) to (Trc ′) is
The calculation means (10) repeatedly calculates each of the switching elements (Tr
Since the ON time of a) to (Trc ') is divided into a plurality of (for example, four) pulses by the dividing means (11), the carrier frequency is multiplied by this number of divisions and the carrier frequency is equivalently high. The situation is the same as when the pulse width modulation control is performed (for example, about 20 KHz). As a result, when each of the switching elements (Tra) to (Trc ') is ON-controlled by the control means (12) with each of these divided pulses, an accurate output waveform close to a sine wave is obtained, and electromagnetic noise is generated. Is effectively reduced and the motor efficiency is effectively increased.

Here, the carrier frequency of pulse width modulation control is a normal value (about 5 KHz), and even a 1-chip microcomputer with a long calculation time can sufficiently calculate the PWM control pattern, so pulse width modulation control with a high carrier frequency is low cost. Therefore, it can be performed with a simple circuit configuration.

Furthermore, the switching element (Tr
The ON time division of a) to (Trc ′) can be appropriately selected into division into equal-width pulses and division into non-uniform-width pulses according to the change rate of the ON time, so that each switching element ( Tr
a) ~ (Trc ') When the rate of change of ON time is large, division into non-uniform width pulses can be selected to ensure good reproducibility of the signal wave, and when the rate of change of ON time is small, uniform width By selecting division into pulses, the calculation and processing time can be shortened, and the calculation cycle can be set to a short time. As a result, it is possible to equivalently perform pulse width modulation control with a sufficiently high carrier frequency while ensuring good reproducibility of the waveform of the signal wave.

(Example) Hereinafter, the Example of this invention is described based on drawing.

1 and 2 show a pulse width modulation (hereinafter abbreviated as PWM) control device for an inverter according to the present invention. In each figure, (1) is three windings (2a), (2b), (2c) Y
An induction motor having a three-phase winding (2) connected thereto, (3) is a voltage type inverter connected to the induction motor (1), and the inverter (3) includes the induction motor (1). A three-phase winding (2) of the transistor bridge circuit (4) is provided, the bridge circuit (4)
Are transistors (switching elements) (Tra), (Tra '), (Trb), (Trb'), (Tr) such as a plurality (six) of MOSFETs each having free wheeling diodes (Da) to (Dc ').
c) and (Trc ′). Then, the inverter (3)
A DC voltage is applied to the rectifier (6) for rectifying the three-phase AC of the three-phase power supply (5).

Further, (8) is a one-chip microcomputer that forms the ON time of the six transistors (Tra) to (Trc ') of the bridge circuit (4), that is, a PWM control pattern, and Are the transistors (Tra) to (Tr
A base driver (8a) for turning on / off c ') is provided, and a transistor (Tr
DC pulse width modulation is performed by ON / OFF control of a) to (Trc ').

Next, the formation of the PWM control pattern by the microcomputer (8) will be described.

The outline of the formation of this PWM control pattern is to determine the PWM control pattern so that the locus of time integration of the output voltage approaches a circular locus. To explain this in detail, first, let va, vb, vc be the output terminal potential of the inverter (3), vn be the neutral point potential of the three-phase winding (2), and the output defined by the following equation. Voltage vector Time integral of think of.

Now, the balanced three-phase voltage of angular frequency ω is applied to the three-phase winding (2) of the induction motor (1). Voltage vector when voltage is applied And its time integration Draws a circular locus on the complex plane.

On the other hand, in the voltage source inverter (3), one of the transistors in each phase arm is always in the ON state, so for convenience, the + side ON state is set to “1” and the − side ON state is set to “0”. If the expressions a, b, and c are written in this order as "101", "011", etc., there are eight states of the inverter (3).
Voltage vector for each state Is the size (Vd is the DC voltage of the rectifier (6)) and its direction is
The direction is as shown in FIG. here, And is a zero vector. Time integration of the above voltage vector Therefore, time integration when driving the inverter (3) It moves at the speed of (However, it stops when the vector is zero).

From the above, the PWM control pattern of the voltage source inverter (3) is the time integration of the voltage vector. Voltage vector so that the vector locus on the complex plane of moves along the circumference of specified radius R at angular velocity ω Is appropriately selected and determined. (The designated radius R is R = V _{1 where} V _{1 is} the effective value of the line voltage of the fundamental wave voltage and ω is the angular frequency.
/ ω).

That is, for example, as shown in FIG. 4, the angle φ is 0 ≦ φ ≦ π
In the range of / 3, the voltage vector And zero vector Used, time tau ₀ only stays at the point P ₀ (this state is shown by symbol °), then For time τ ₄ to reach point q ₁ Let us consider the case of taking time τ ₆ to reach point P ₁ . In this case, at ΔP ₀ q ₁ P ₁ , And τ ₀ + τ ₄ + τ ₆ = T ₀ , the above equation is solved to obtain the voltage vector within the period T ₀ . The times τ ₄ , τ ₆ and τ ₀ are obtained.

τ ₄ / T ₀ = k _S · Sin (π / 3−φ ₀ ) τ ₆ / T ₀ = k _S · Sin φ ₀ τ ₀ / T ₀ = 1−k _S · Sin (φ ₀ + π / 3) ... (3) where k _S is the voltage control rate, The above expression (3) is a relational expression when the angle φ is in the range of 0 ≦ φ ≦ π / 3, but since the inverter (3) performs symmetrical three-phase operation in other sections, the following first expression By replacing each symbol as shown in the table, a relational expression in the range of 0 ≦ φ ≦ 2π can be obtained.

Next, the ON / OFF pattern (PW) of each of the transistors (Tra) to (Trc ′) is calculated based on the time τ of the voltage vector in the above equation (3).
M control pattern). In this case, the relationship between time τ and the PWM control pattern of the voltage vector, because changes in response to the order to take the voltage vector, now, for simplicity, the repeated same pattern in each period T _0, each period T ₀ If the constraint condition that the ON / OFF switching of the transistor in the inside is only once is added, the PWM control pattern will be as shown in FIG.
It is represented by the four patterns shown in (d) (in the figure, τ ⁺ is +
The ON time of the transistor on the negative side and τ ⁻ indicate the ON time of the transistor on the negative side).

In this embodiment, the PWM control pattern shown in FIG. In the voltage source inverter (3), the PWM control pattern is uniquely determined by knowing the name of the transistor that is turned on at the beginning of the period T ₀ and the time when it turns off. Referring to Fig. 5 (a), PW
The M control pattern is determined by the following equation when the angle φ is in the range of 0 ≦ φ ≦ π / 3.

The relational expression (4) of the PWM control pattern in the range of 0 ≦ φ ≦ π / 3 is 0 ≦ φ if each symbol is replaced in the same manner as above.
The relational expression is in the range of ≦ 2π.

Next, the operation of the one-chip microcomputer (8) is shown in FIGS.
It will be described with reference to FIG. 8 based on the control flow of the figure.
For convenience of explanation, each transistor (Tra) to (Trc ')
First, the case where the ON time of is divided into a plurality of constant-width pulses will be described.

The control flow of FIG. 6 is that each transistor (Tra)-(Tr
c ′) ON time (PWM control pattern) calculation flow. The control flow in Fig. 7 is the actual flow of each transistor (Tr
This is a flow for turning on a) to (Trc ′). First, the control flow of FIG. 6 will be explained. The control flow is based on the calculation cycle T ₀ (for example, 200 K) depending on the carrier frequency (for example, 5 KHz).
μS), and after inputting the output voltage phase ωt (= φ ₀ ) and the output voltage amplitude V ₁ in step S _A1 ,
In step S _A2 , the ON time τ (n) of each of the transistors (Tra) to (Trc ′) is based on the relational expression (4) of the PWM control pattern.
+1) is calculated.

Then, subsequently, the ON time τ (n + 1) of the transistors (Tra) to (Trc ′) calculated above in step S _A3 is divided by a preset numerical value N (for example, 4) to obtain each ON time τ. A plurality of (n + 1) N (4) pulses τ ′ (n +
1) Divide into (τ '(n + 1) = τ (n + 1) / 4).
Then, in step S _A4 , this divided pulse τ ′ (n +
Store 1) in the switching time register for each phase as shown in FIG. 9 (in the voltage source inverter, one of the transistors in each phase arm is always in the ON state, so only one phase is required). And return.

Further, in the control flow of FIG. 7, the repetition cycle T ₀ ′ is earlier than the calculation cycle T _{0 of} FIG. 6 and the ON time τ (n +
According to the number of divisions N (4) in 1), T ₀ ′ = T ₀ / N is set (the number of divisions N is N = 2 ^m (m = 1 where division can be performed only by shifting). , 2 ...) is preferable).
Then, in the control flow of FIG. 6, the divided pulse τ ′
After (n + 1) is stored in the switching time register of each phase, the contents of the switching time register are input in step S _B1 in the next calculation cycle T ₀ as shown in FIG. 8, and in step S _B2 . With the divided pulse τ '(n + 1), the corresponding transistors (Tra) to (Trc') are ON-controlled and the process returns.

Therefore, in the calculation flow of the PWM control pattern of FIG. 6, the carrier frequency (5 KHz) is determined by steps S _A1 and S _A2.
Each transistor (switching element) based on the relational expression (4) of the PWM control pattern with the calculation cycle according to
The calculation means (10) is configured to calculate the ON time τ (n + 1) of (Tra) to (Trc ′).

Next, FIG. 10 to FIG. 14 show a case where the ON time of each transistor (Tra) to (Trc ′) is divided into a plurality of non-uniform width pulses.
It will be described with reference to the drawings.

That is, the division into unequal width pulses, the ON time of the transistor tau (n) and in the period T _0, a linear interpolation (linear interpolation between the next ON time of the period T ₀ of tau (n + 1) ) Is done.

This will be described in detail. Control flow of FIG. 10 is processing in a period T ₀ period, and inputs the phase ωt and amplitude V ₁ of the output voltage in step S _C1, the transistors of the last time step _{S C2 (Tra) ~ (Trc} ' ) ON time τ (n-1) (4
Input the operation result of the divided pulse).

Then, in step S _C3 , based on the relational expression (4) of the PWM control pattern, each transistor (Tra) to (Tr
c ') ON time τ (n) is calculated, and this ON time τ (n)
Divided pulse τ ′ into a plurality of N (4)
(N) is calculated, and then, in step S _C4 , the previous divided pulse τ ′ (n−1) is calculated according to the difference between the divided pulse τ ′ of the transistors (Tra) to (Trc ′) and the divided pulse τ ′ of the previous time. The interpolated value Δτ _{n-1 of} is calculated based on the following formula.

Δτ _n-1 = {τ ′ (n) −τ ′ (n−1)} / NN; the number of divisions is N = 4, and the previous four divided pulses τ ′ (n−1) are interpolated values Δτ. _In step S _C5 , each divided pulse τ ′ (n−1) is sequentially interpolated from the second one ΔΔ so as to interpolate gradually with n−1.
_n−1 , 2 · Δτ _n-1 , 3 · Δτ _n-1 are added, and step S _C6
Then, each of these divided pulses is stored in a plurality of N (N = 4) switching time registers for each phase, and step S _C7
Then, each divided pulse τ '(n-1) is stored and the process returns.

Further, the control flow of FIG. 11 is such that the divided pulse τ ′ is calculated and stored as shown in FIG. 12 in the period T ₀ to the second period T ₀ in which the divided pulse τ ′ is used for each of the transistors (Tra) to (Trc). ′)
Is ON-controlled, and its control cycle T ₀ ′ is 1 / N (N is the number of divisions) of the calculation cycle T ₀ of the control flow in FIG.

The control flow then is loaded Step S _D 1 at the 13th second first divided pulse for each phase stored in the switching time register as shown in FIG tau ', shifting the switching time register in step S _D2 Then, in step S _D3 , each transistor (Tr
a) to (Trc ′) are ON-controlled, and then the process is returned. Similarly, for each control cycle T ₀ ′, the second, third, and fourth switching time registers are sequentially stored for each phase. Read the divided pulse, each transistor (Tra) ~ (Trc ')
Repeat ON control.

Thus, FIG. 15 shows the transistors (Tra) to (Trc ').
Whether the ON time is divided into equal-width pulses or non-uniform-width pulses is appropriately selected according to the rate of change of the ON time.

In this embodiment, each transistor (Tra) to (Trc ') is simply used.
Dividing into equal-width pulses according to the rate of change of ON time, that is, the phase of the signal wave (control flow in FIGS. 6 and 7)
, And the division into non-uniform width pulses (control flow in FIG. 10 and FIG. 11), the symmetry of the voltage vector is used to select each transistor (Tra) to (Tr).
The calculation of the ON time of c ′) and the division into a plurality of N (N = 4) are performed in the same manner for all angles 0 ≦ φ ≦ 2π so that the calculation time can be further shortened. .

That is, as can be seen from the relational expression (4) of the PWM control pattern, when the angle φ ₀ is in the range of 0 ≦ φ ₀ ≦ π / 3, the ON time τa of the transistor is Sin (φ ₀ + π as shown in FIG. /
Since the change rate of the ON time is small in the range of 3), good waveform reproducibility is ensured while dividing into equal-width pulses with a short calculation time. Also, turn on the transistor
Since the time τb is in the range of Sin φ ₀ and the change rate of the ON time is large, division into non-uniform width pulses is adopted in order to ensure good waveform reproducibility. Then, in consideration of the above, in order to expand the angle φ ₀ to 0 ≦ φ ₀ ≦ 2π, the entire section is divided into 6 sections for each π / 3, and is shown in the following Table 2.

FIG. 15 shows the above-mentioned one-chip microcomputer (8). In the figure, (15) is a section information calculation circuit for calculating and discriminating the section N of the above Table 2 from the phase ωt, and (16) is a section signal N from the section information calculation circuit (15) and the phase ωt. From the symmetry of the voltage vector, the angle φ ₀ is 0 ≦ φ ₀ ≦ π over the entire interval N (N = 0 to 5).
The angle calculation circuit is calculated by the following formula φ ₀ = ωt− (N · π / 3) to unify the range of / 3, and (17) is the angle calculation circuit (1
Input the angle φ ₀ calculated in 6) and the effective value V ₁ of the fundamental wave voltage, and calculate the ON time of each transistor (Tra) to (Trc ′) based on the relational expression (4) of the PWM control pattern. Together with this, it is a pulse division circuit that divides this ON time into a plurality of constant-width pulses. Further, (18) is a non-uniform width pulse arithmetic circuit, and the non-uniform width pulse arithmetic circuit (18) is a transistor (Tra) calculated by the pulse division circuit (17).
To (Trc ') ON time, the divided equal-width pulses, and the section signal N from the section information calculation circuit (15), and according to section N, based on the above table 2, non-uniform width Grasping the ON time of the transistors (Tra) to (Trc ') that should be divided into pulses (that is, the function of Sin0 to Sin π / 3 in Table 2), only the ON time that should be divided into these non-uniform width pulses The above 10
By the same operation as the control flow of FIG. 11 and FIG. 11, the pulse is divided into a plurality of non-uniform width pulses, and the divided non-uniform pulse and the equal-width pulse equally divided by the pulse division circuit (17) are added to the base driver ( 18a) output function.

Therefore, the ON time τ of each of the transistors (Tra) to (Trc ′) calculated by the calculating means (10) of FIG. 6 by the unequal width pulse calculating circuit (18) is calculated as the rate of change of the ON time. , When the rate of change is small (in Fig. 14, the angle φ ₀ is
Sin π / 3 to Sin (φ ₀ + π / 3)) has a total pulse width of ON time τ of each of the transistors (Tra) to (Trc ′) and the same time (= τ), When it is divided into a plurality of (= N) equal-width pulses of pulse width τ / N (N is the number of divisions) and the change rate of the ON time is large (in Fig. 14, the angle φ ₀ is Sin0 to Si).
In the range of n π / 3), the previous values of the transistors (Tra) to (Trc ′) calculated by the calculation means (10) are
Based on the ON time τ (n-1) and the current ON time τ (n), a plurality of previous ON times τ (n-1) N (N =
Split pulse τ '(n-1) divided into 4) and this time ON
A similar divided pulse τ ′ (n) of time τ (n) is calculated, and the difference τ ′ (n) −τ ′ (n
-1), an interpolation value Δτ _n-1 is calculated, and the pulse width gradually changes by the interpolation value Δτ _n-1 between the divided pulses τ '(n-1) and τ (n-1). N (N = 4) non-uniform width pulses τ ′, τ ′ + Δτ, τ ′ + 2 · Δτ, τ ′ +
3 · .DELTA..tau is divided into 'a, the transistors (Tra) ~ (Trc' period T ₀ of the divided plurality of equal-width pulse or unequal width pulse) of ON time of the calculation period T ₀ of the The dividing means (11) is configured to multiply the number of divisions (N) by one (T ₀ / N). Further, according to the control flow of FIG. 7 and the control flow of FIG. 11, each transistor () is divided into a plurality of N (N = 4) equal-width pulses and unequal-width pulses divided by the dividing means (11). Tra) to (Trc ') are turned on to constitute a control means (12).

Therefore, in the above embodiment, each of the transistors (Tra) to (Trc ′) is turned on based on the relational expression (4) of the PWM control pattern in the PWM control pattern calculation flow (FIG. 6).
After the time τ is calculated by the calculation means (10), each of these ON
After the time τ is calculated by the calculation means (10), each of these ON
The time τ is divided by the dividing means (11) into a plurality of N (4) pulses τ ′, and the divided pulses τ ′ are stored in the switching time registers for the respective phases a, b, c.

Then, in the subsequent cycle T ₀ , as shown in FIGS. 8 and 12, in each period T ₀ , each transistor (Tr
a) to (Trc ′) ON time τ is calculated and divided, and at this time period T _0, at the period T ₀ / N (= T ₀ ′), the period T Each of the transistors (Tra) to (Trc ') corresponding to the divided pulse τ'in the switching time register of each phase of a, b, c obtained by ₀ is controlled by the control means (1
Since the ON is controlled by 2), the frequency of the high frequency component can be increased compared to the conventional one as shown in Fig. 18 (where the ON time is not divided into multiple pulses), and the carrier frequency is turned ON equivalently. The time division number N (N = 4) can be multiplied, and the original carrier frequency (5 KHz) can be increased to a high carrier frequency (20 KHz). Incidentally, FIG. 8 and FIG.
The figure shows the case where the ON time of the + side transistor of each phase is calculated.

Here, the original carrier frequency (5 KHz), that is, the ON time calculation cycle T ₀ (200 μS) is the 1-chip microcomputer (8)
However, since the period is long enough to calculate the PWM control pattern, it is possible to perform PWM control at a high carrier frequency (about 20 KHz) while using the 1-chip microcomputer (8), and at a low cost and circuit. MOSF while simplifying the configuration
The three-phase AC waveform to the induction motor (1) can be precisely controlled by taking advantage of the capability of high-speed switching elements such as ET, and electromagnetic noise can be reduced and motor efficiency can be increased.

If a carrier frequency (5 KHz) similar to the conventional one is sufficient, the calculation time of the one-chip microcomputer (8) can be shortened and the processing capacity other than the PWM control can be enhanced.

Moreover, the division of ON time by the dividing means (11), as shown in FIG. 14, in the range of ON time rate of change is larger place in unequal width pulse, divide the pulse tau 'the first period T' ₀ in the output, so the next cycle T 'be greater divided pulses in ₀ than the divided pulse by interpolation value Δτ is output control period T' is repeated at ₀ (see FIG. 12), 17
As compared with the case of an equal width pulses shown in FIG. (B), with respect to the average value of the output voltage at the control cycle T _'0 corresponding to equivalent carrier frequency as shown in FIG. 14, the reproducibility of the waveform Can be secured satisfactorily.

Further, in the range where the change rate of the ON time is small, the change is small. Therefore, even if division is performed with a constant width pulse, the reproducibility of the waveform with respect to the average value of the output voltage is not as shown in Fig. 17 (b). It can be secured almost as well as when using a constant-width pulse, and in the range where this ON time change rate is small.
Since the ON time is divided into equal-width pulses, the calculation time required to calculate the interpolation value Δτ becomes unnecessary. Therefore, while maintaining good waveform reproducibility, there is an effect that the calculation and processing time can be saved, and the PWM control with a higher carrier frequency can be performed correspondingly.

Moreover, the calculation of the PWM control pattern can be performed in the same manner in the entire range of the angle ωt of 0 ≦ ωt ≦ 2π by utilizing the symmetry of the voltage vector, and thereafter, it is divided into equal-width pulses based on Table 2. Since it is possible to easily understand whether the ON time should be divided or the ON time should be divided into equal-width pulses, the calculation suitable for the microcomputer,
In addition to processing, the calculation and processing can be further simplified.

Further, FIG. 16 shows a modification, in which the number of divisions N of each of the transistors (Tra) to (Trc ') is set to a set value (N =
Instead of fixing to 4), each transistor (Tra) ~ (T
It is changed appropriately according to the change rate of the ON time of rc ').

That is, in the angle range where the change rate of the ON time is large, the period T ₀
The number N of divisions in the middle is set to a large N = 8, and the number of divisions N in the period T ₀ is set to the normal N in the angular range in which the change rate of the ON time is small.
= 4 is set. Therefore, as can be seen from the figure, while reducing the calculation and processing time of the microcomputer (8) in the angle range where the ON time change rate is small, the waveform reproducibility is improved in the angle range where the ON time change rate is large. It can be further improved.

The portion for converting the content of the switching time register of each phase into a pulse width may be processed by hardware such as an external pulse width modulation IC. Furthermore, FIGS. 9 and 13
With the configuration shown in the figure, even when the carrier frequency changes due to the change of the switching element, the dividing means (11)
It is sufficient to change only the control means (12). In addition, the switching time register is connected to the pulse width control unit (step S
If shared with the register of _B2 ), the processing of step S _B1 in FIG. 7 can be omitted.

Furthermore, the calculation cycle T ₀ in the calculation flow of the PWM control pattern
Is uniquely determined by the time required to actually calculate the PWM control pattern (including division of ON time).
The period T ₀ ′ of ON control of the transistor in the control flow of FIG. 11 and FIG. 11 is determined according to the desired carrier frequency,
Therefore, the number of divisions of the ON time of each transistor N (T ₀ /
The value of T ₀ ′) may be set to an appropriate value.

Further, in the above embodiment, the PWM control pattern is obtained based on the relational expression (4) in the case of voltage vector control.
Of course, it may be obtained based on a relational expression in the case of another PWM control method such as a triangular wave comparison method.

(Effects of the Invention) As described above, according to the pulse width modulation control device for the inverter of the present invention, the ON time of the switching element that is repeatedly calculated at the calculation cycle corresponding to the carrier frequency
It is divided into a plurality of equal-width pulses or non-uniform-width pulses according to the rate of change of time, and each switching element is ON controlled by this divided pulse, so it takes a relatively long time to calculate the ON time of the switching element. In this case, the carrier frequency can be equivalently increased. For example, even when a low-priced, simple-circuit 1-chip microcomputer is used, the three-phase AC waveform can be precisely controlled to reduce the electromagnetic noise and the motor. The efficiency can be increased. Moreover, when the ON time change rate is large, the division is performed by the non-uniform width pulse, and when the ON time change rate is small, the division is performed by the uniform width pulse, while ensuring good reproducibility of the waveform of the signal wave. The PWM control pattern calculation and processing time by the microcomputer can be effectively shortened, and pulse width modulation control at a higher carrier frequency can be achieved.

[Brief description of drawings]

1 to 16 show an embodiment of the present invention, FIG. 1 is an overall schematic configuration diagram, FIG. 2 is an electric circuit diagram, and FIG. 3 shows various states of a voltage source inverter by eight kinds of voltage vectors. The displayed explanatory diagram, FIG. 4 is an explanatory diagram of the voltage vector control for making the locus on the complex plane of the time integration of the voltage vector close to the circular locus, and FIGS. 0 ≦ φ ≦
FIGS. 6 and 7 are explanatory views of types of PWM control patterns that can be taken within the range of π / 3, and FIGS. 6 and 7 are flow charts showing ON / OFF control with a constant-width pulse of each transistor by a one-chip microcomputer, respectively. The figure shows the carrier frequency of the transistor
Fig. 9 is an explanatory diagram equivalently increased due to the equal division of ON time. Fig. 9 is an explanatory diagram of the operation when dividing into ON-width pulses, Fig. 10 and Fig.
Fig. 11 is a flow chart showing ON / OFF control of non-uniform width pulses of each transistor, Fig. 12 is an explanatory diagram of interpolation of each divided pulse when dividing into non-uniform pulses, and Fig. 13 is not FIG. 14 is an explanatory diagram showing the operation in the case of dividing into equal-width pulses, FIG. 14 is an explanatory view showing an angular range of a signal wave for selecting division with equal-width pulses and division with unequal widths, and FIG. Fig. 16 is a block diagram of the microcomputer in the case of dividing into equal width pulse and non-uniform width pulse as appropriate according to the rate of change of time, Fig. 16 shows the case where the number of ON time divisions is changed according to the change of transistor ON time. FIG. In addition, FIGS. 17A and 17B are explanatory diagrams showing the reproducibility of the waveform when divided into equal-width pulses and when divided into non-uniform-width pulses, respectively. Further, FIG. 18 is an explanatory view showing a conventional example. (2) …… Three-phase winding, (3) …… Voltage type inverter,
(4) …… Bridge circuit, (Tra) to (Trc ′) …… Transistor, (8) …… One-chip microcomputer, (10) …… Computing means, (11) …… Division means, (12) …… Control means.

Claims (2)

[Claims] 1. A bridge circuit (4) which is connected to a three-phase winding (2) and has a plurality of switching elements (Tra) to (Trc '), and each switching element () of the bridge circuit (4). Tra) to (Trc ′) ON / OFF operation for pulse width modulation of DC to apply a three-phase AC voltage to the three-phase winding (2). A calculating means (10) for calculating the ON time of each of the switching elements (Tra) to (Trc ') at an operation cycle corresponding to the carrier frequency, and each switching element (Tra) calculated by the calculating means (10) The ON time of (Trc ') is changed according to the rate of change of the ON time by a plurality of constant-width pulses having a total pulse width simultaneously with the ON time or switching calculated by the calculating means (10). Based on the previous ON time of elements (Tra) to (Trc ′) and this time The pulse width is divided into one of a plurality of non-uniform width pulses whose pulse width is gradually changed, and the periods of the plurality of divided uniform-width pulses or non-uniform-width pulses are divided into the switching elements (Tra) to (Trc '). ) ON time calculation cycle divided by the above division number (1)
1) and a control means (12) for ON-controlling each of the switching elements (Tra) to (Trc ') with a plurality of pulses divided by the division means (11). Pulse width modulation controller. 2. The dividing means (11) divides the switching elements (Tra) to (Trc ') into unequal width pulses when the rate of change of the ON time is large, and equal width when the rate of change of the ON time is small. The pulse width modulation control device for an inverter according to claim 1, wherein the pulse width modulation control device divides into pulses.