JPH07235839A - Variable conductance amplifier - Google Patents

Abstract

PURPOSE:To reduce the power consumption and transmission distortion by providing a 1st circuit employing an ohmic resistive element to implement voltage/current conversion with excellent linearity and a 2nd circuit implementing variable control of a mutual conductance to the amplifier. CONSTITUTION:An input voltage Vin applied between inputs inm, imp in a 1st circuit 1 is applied between terminals of an ohmic resistive element R through the source follower operation of transistors(TRs) M1, M2. A gate voltage of TRs M3, M4 is given to gates of TRs M5, M6 of a 2nd circuit 2. Thus, the TRs M5, M6 generate a drain current in response to a drain current of the TRs M3, M4, that is, the current Tin flowing to the resistive element R. A rate of change in an output current IGmc with respect to the input voltage Vin, that is, an output mutual conductance is controlled by controlling variably a current generated from a variable current source with a control voltage Vc.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a technique effective when applied to a variable conductance amplifier, and further to a variable conductance amplifier for converting a voltage / current of an analog input voltage, for example, an RC time constant type analog filter. The present invention relates to a technique effectively used for the time constant element of.

[0002]

2. Description of the Related Art For example, when an RC time constant type analog filter is to be realized in a semiconductor integrated circuit device, variations in the cutoff characteristic or frequency characteristic of the filter occur due to variations in manufacturing of the semiconductor integrated circuit device. It is generally difficult to obtain the characteristics described below with high accuracy and reproducibility.

Conventionally, in order to avoid the above-mentioned characteristic variation, a switched capacitor type filter has been used or a capacitance forming a time constant of the filter has been trimmed and adjusted.

However, in the switched capacitor system,
Since the charge and discharge of the capacity are repeated at high speed, there is a problem that the current consumption increases. Further, the trimming adjustment of the capacitance requires a larger element area on the semiconductor integrated circuit device as the trimming adjustment is performed with higher accuracy, which causes a problem that the required area of the semiconductor chip increases. Moreover, the number of trimming points is not limited to one, and it is often necessary to trim and correct a plurality of element constants at the same time.

Therefore, the present inventors have examined the use of a so-called variable conductance amplifier that equivalently realizes a time constant element such as an analog filter, especially a resistor, by an active circuit, as shown in FIGS.

In the circuit shown in FIG. 7, the drain currents flowing through the pair of MOS transistors M5 and M6 are controlled by the constant current sources I1 and I2.
The common source of the OS transistors M5 and M6 is a constant current source I
5, a source-coupled differential amplifier is configured, and the forming current of the constant current source I5 is variably manipulated by the control voltage Vc, whereby the rate of change of the output current Iout depending on the voltage Vin between the inputs inm and inp. This is intended to change the so-called mutual conductance Gm.

The circuit shown in FIG. 8 has a MOS transistor M.
1 to M6 and constant current sources I1 to I4, the voltage Vin between the inputs inm and inp is applied to the MOS transistor M.
1 and M2 are applied between the source and drain of the MOS transistor Mx, and the MOS is applied by the applied voltage Vin.
The source-drain current Ix of the transistor Mx is
The output current Iout formed by the MOS transistors M5 and M6 is reflected via the OS transistors M3 and M4. The gate applied voltage Vc of the MOS transistor Mx is variably manipulated to change the input voltage Vout.
The rate of change of the output current Iout due to in, so-called mutual conductance Gm is changed.

Documents relating to the variable conductance amplifier shown in FIG. 8 include, for example, the following.

(1) IEE Journal of Solid State Circuits, Volume 23 (June 1988) pp. 750-758 (IE
EE, JOURNAL OF SOLID-STATE
CIRCUIT, VOL23, NO. 3, JUNE
1988 pp. 750-758).

(2) Volume 26 (6th 1991)
Mon) pp. 859-867 (IEEE, JOURNAL
OF SOLID-STATE CIRCUIT, V
OL26, NO. 6, JUNE 1991 PP. 85
9-867).

[0011]

However, the present inventors have clarified that the above-mentioned technique has the following problems.

That is, the structure shown in FIG. 7 causes a problem that the transmission distortion due to the non-linearity in the common source becomes large. This non-linear distortion becomes more remarkable as the amplitude of the input voltage Vin increases.

Further, in the structure shown in FIG. 8, due to the non-linearity of the MOS transistor Mx due to the substrate effect or the like, the drain voltage between the drain-source applied voltage Vin and the drain-source current Ix of the MOS transistor Mx is good. However, the linear distortion cannot be ensured, and in this case, too, the transmission distortion becomes large.

An object of the present invention is to provide a technique which is suitable for low power consumption and which enables smooth change of mutual conductance while keeping transmission distortion low.

The above and other objects and characteristics of the present invention will be apparent from the description of this specification and the accompanying drawings.

[0016]

Of the inventions disclosed in the present application, a representative one will be briefly described below.
It is as follows.

That is, an input voltage applied between the two inputs, inverting and non-inverting, is applied to both ends of the ohmic resistance element via the first and second MOS transistors, and the ohmic resistance element by this applied voltage is applied. Forming current is 3rd and 4th M
A first circuit for feedback-controlling the gate voltages of the third and fourth MOS transistors so that the drain current of the OS transistor is obtained, and a gate circuit to which the gate voltages of the third and fourth MOS transistors are applied. 5th and 6th M
The common source of the OS transistor is controlled by the variable constant current source, and the currents flowing through the fifth and sixth MOS transistors are controlled by the current mirror to have a predetermined mirror ratio. A second circuit for obtaining an output current converted from the input voltage, and varying the forming current of the variable current source to change the mutual conductance.

[0018]

According to the above-mentioned means, in the first circuit, the voltage / current conversion having excellent linearity can be performed by the ohmic resistance element having no voltage dependency, and the conversion current is the gate voltage of the MOS transistor. By being converted to and transmitted to the gate of the MOS transistor of the second circuit, the second circuit can perform the variable control of the mutual conductance while maintaining the low distortion by the voltage input having the very small amplitude. .

With this, it is possible to achieve the object that the power consumption is suitable and the mutual conductance can be smoothly varied while keeping the transmission distortion low.

[0020]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT A preferred embodiment of the present invention will be described below with reference to the drawings.

In the figures, the same reference numerals indicate the same or corresponding parts.

FIG. 1 shows an embodiment of a variable conductance amplifier to which the technique of the present invention is applied.
Is a first circuit for converting the input voltage Vin into a current Iin,
Reference numeral 2 is a second circuit that controls the current Iin converted by the first circuit 1 so as to be output with a predetermined transconductance. inm is an inverting input, inp is a non-inverting input, V
in is an input voltage, out is an output, IGmc is an output current,
Vc is a control voltage for variably operating the rate of change of the output current IGmc with respect to the input voltage Vin, so-called mutual conductance (Gm), and Vdd is a power supply potential.

The variable conductance amplifier according to the present invention comprises a first circuit 1 forming a voltage / current conversion stage and a second circuit 1 forming a mutual conductance (Gm) control stage.

The first circuit 1 includes first to fourth MOS transistors M1 to M4, first to fourth constant current sources I1 to I4,
And an ohmic resistance element R. In this case, the p-channel type is used for the first and second MOS transistors M1 and M2, and the p-channel type is used for the third and fourth MOS transistors M1 and M2.
An n-channel type is used for each of M3 and M4.

In the first MOS transistor M1, the gate is connected to the inverting input inm, the source is connected to the first constant current source, and the drain is connected to the third constant current source I3.
The second MOS transistor M2 has a gate connected to the non-inverting input inm, a source connected to the second constant current source I2, and a drain connected to the fourth constant current source I4. The constant current sources I1 and I2 have the same current, and I3 and I4 have I
1, I2 is formed by subtracting the bias currents of M3 and M4.

In the third MOS transistor M3, the gate is the drain of the first MOS transistor, the drain is the source of the first MOS transistor M1 and one end of the ohmic resistance element R, and the source is the reference potential (ground potential). Each is connected. Fourth MOS transistor M4
Has a gate at the drain of the second MOS transistor,
The drain is connected to the source of the second MOS transistor M2 and the other end of the ohmic resistance element R, and the source is connected to the reference potential (ground potential).

The ohmic resistance element R is a pure resistance element having a linear voltage / current characteristic (ohmic characteristic).
Is connected between the source of the MOS transistor M1 and the source of the second MOS transistor M2.

The second circuit 2 is constituted by fifth and sixth MOS transistors M5 and M6 whose sources are commonly connected, a current mirror circuit 21, and a variable constant current source I5 whose forming current is controlled by a control voltage Vc. Has been formed. In this case, the fifth and sixth MOS transistors M5 and M6 have n
Channel type is used.

The gate of the fifth MOS transistor M5 is connected to the gate of the third MOS transistor M3, and the drain thereof is connected to one current forming path of the current mirror circuit 21. Sixth MOS transistor M6
Has a gate connected to the gate of the fourth MOS transistor M4 and a drain connected to the other current forming path of the current mirror circuit 21. The common source of both MOS transistors M5 and M6 is a variable constant current circuit I5.
It is connected to the.

Next, the operation will be described.

In the first circuit 1, the input voltage Vin applied between two inversion and non-inversion inputs inm and inp is
The source follower operation of the first and second MOS transistors M1 and M2 applies the voltage across the ohmic resistance element R. As a result, the applied voltage Vi is applied to the resistance element R.
n according to the resistance value of the resistance element R and Iin (Iin =
Vin / R) flows. This current Iin flows into the drains of the third and fourth MOS transistors M3 and M4.

At this time, the gate voltages Vg3 and Vg4 of the third and fourth MOS transistors M3 and M4 are I1-I3 + I, where Iin is the forming current in the resistance element R.
Feedback control is performed to a voltage such that in and I2-I4-Iin become drain currents. That is, the gate voltages Vg3 and Vg4 of M3 and M4 are equal to the forming current Iin of the resistance element R.
Change. In this way, the current Iin converted from the input voltage Vin by the resistance element R is taken out in the form of being converted into the gate voltages Vg3 and Vg4 and transmitted to the second circuit 2.

In the second circuit 2, the gate voltages Vg3 and Vg4 of the third and fourth MOS transistors M3 and M4, which flow the formation current Iin as a drain current, are the same as those of the fifth and sixth MOS transistors M5 and M6. Given to the gate. As a result, the fifth and sixth MOS transistors M
5 and M6 are a third current and a fourth current due to a kind of current mirror.
The drain current of the transistors M3 and M4, that is, the drain current corresponding to the current Iin flowing through the resistor R is formed.

At this time, the sixth MOS transistor M6
On the drain side of the current mirror circuit 21,
The same current as that flowing to the drain side of the fifth MOS transistor M5 is supplied. On the other hand, the total source current of both MOS transistors M5 and M6 is limited to the current formed by the variable constant current source I5.

As a result, the sixth MOS transistor M6
When the output out is drawn from the drain of the output current IGmc (= Iou), the output out has a constant proportional relationship with the current Iin formed by the resistance element R.
t) is obtained, and the rate of change of the output current IGmc with respect to the input voltage Vin, so-called mutual conductance, depends on the forming current of the variable constant current source I5.

As described above, in the first circuit 1, the voltage / current conversion excellent in linearity can be performed by the ohmic resistance element R having no voltage dependence. At the same time, the converted current Iin is transferred to the MOS transistors M3, M
It is converted into a gate voltage of 4 and transmitted to the gates of the MOS transistors M5 and M6 of the second circuit 2. Where M
When the change in the drain current of the OS transistors M3 and M4 is converted into the change in the gate voltage, the MOS transistor M
Due to the amplification effect (transconductance) of M3 and M4, the change in the gate voltage becomes very small with respect to the change in the drain current.

Therefore, in the second circuit 2, the transconductance can be variably controlled while maintaining the low distortion rate by the voltage input having a very small amplitude. This is suitable for low power consumption, and the mutual conductance can be smoothly varied while keeping the transmission distortion low.

The operation of the circuit shown in FIG. 1 will be described below using a mathematical model.

As described above, the current Iin flowing through the resistance element R due to the input voltage Vin is Iin = Vin / R, and this current Iin is the MOS transistors M3, M.
Pour into 4.

Gate voltages Vg3 and Vg4 of M3 and M4
Is the transconductance of M3, M4 gm3, gm4
In this case, ΔVg3 = ΔIin / gm3 ΔVg4 = ΔIin / gm4 changes with respect to the change ΔIin of the current Iin. The gate voltage Vg3 that changes in this way
Vg4 is input to the second circuit 2 forming the Gm control stage.

By inputting the gate voltages Vg3 and Vg4 to the gates, the variations ΔIm5 and ΔIm6 of the forming current in M5 and M6 are ΔIm5 = gm5 if the mutual conductances of M5 and M6 are gm5 and gm6. -ΔVg3 ΔIm6 = gm6 · ΔVg4

Here, the change amount ΔIGmc of the output current IGmc extracted from the drain of M6 is controlled by the current mirror circuit 21 so that the drain side current of M6 becomes the same as the drain side current of M5. Therefore, the forming currents of the constant current sources I1 to I5 are I1 to I5,
Set the channel width / length of MOS transistors M3 to M6 to W
3 / L3 to W6 / L6, ΔIGmc = ΔIm5-ΔIm6 = gm5 · ΔVg3-gm6 · ΔVg4 = 2ΔIin · gm5 / gm3 = 2ΔIin · √ [{β5 · I5 · W5 / L5} / {β
3 (I1-I3) W3 / L3}].

Here, gm5 = gm6 = √ {2 (β5 · I5 · W5 / L5) gm3 = gm4 = √ {2β3 (I1-I3) W3 / L
3}, the rate of change of the output current IGmc with respect to the input voltage Vin, that is, the output transconductance Gm is: Gm = ΔIGmc / ΔVin = 2√ [(β5 · I5 · W5 / L5) / {β3 (I1-
I3) · W3 / L3}] / R.

Therefore, the rate of change of the output current IGmc with respect to the input voltage Vin, that is, the output transconductance Gm can be controlled by variably controlling the forming current I5 of the variable constant current source I5 with the control voltage Vc.

Further, it should be noted that the input voltages Vg3 and Vg4 of the second circuit 2 forming the Gm control stage are reduced to 1 / (gm3 · R) of the input voltage Vin of the first circuit 1. Is Rukoto. Therefore, the voltage amplitude thereof is very small, whereby the transconductance Gm in the second circuit 2 can be varied while keeping the transmission distortion low. Also, for the same reason, the dynamic range of the input voltage Vin of the first circuit 1 can be increased.

FIG. 2 shows a variable conductance amplifier according to the second embodiment of the present invention.

In the second embodiment, in the variable conductance amplifier described above, instead of the variable current source I5,
By switching the current mirror ratio of one current forming path and the other current forming path of the current mirror circuit 21,
The rate of change of the output current IGmc depending on the input voltage Vin, that is, the mutual conductance Gm is variably set.

In FIG. 2, M7 is a p-channel MOS transistor forming one current forming path of the current mirror circuit 21, and M8 and M81 to M84 are p-channel MOS transistors forming the other current forming path. Further, M6, M61 to M64 are the current mirror circuit 2 described above.
A MOS transistor whose drain is connected to the other current forming path of No. 1 is provided corresponding to the MOS transistors M8, M81 to M84 of the other current forming path. 211 is a CMOS switch circuit,
OS transistors M6, M61 to M64 and M8, M
The parallel connection states of 81 to M84 are switched in association with each other. The current mirror ratio is variably set stepwise by the combination of the MOS transistors connected in parallel by this switching, and the change rate of the output current IGmc with respect to the input voltage Vin, that is, the output mutual change, is set by variably controlling the current mirror ratio. Conductance Gm
Can be controlled. C1 to C4 are switch circuits 2
11 is a control signal.

The MOS transistors M6 and M61-
By weighting the element sizes (channel width / length) of M64 and M8, M81 to M84, the variable control of the current mirror ratio can be performed in a wide range.

FIG. 3 shows a variable conductance amplifier according to the third embodiment of the present invention.

In the third embodiment, in addition to the embodiment shown in FIG. 1 or 2, a second current mirror circuit 22 for further current mirror transmitting the output current IGmc from the first current mirror circuit 21 is provided. It is provided. This second
Of the current mirror circuit 22 of the constant current sources I16 and I17 connected to the current forming path on one side and the current forming path on the other side.
Together with this, it forms an output stage for buffer-transmitting (buffer-transmitting) the output current IGmc taken out from the first current mirror circuit 21.

As a result, the first current mirror circuit 2
The drain voltage of the MOS transistor M6 in 1 is output o
The fluctuation due to the influence from the ut side is prevented, and a highly accurate variable conductance amplifier can be realized.

FIG. 4 shows a fourth embodiment of the variable conductance amplifier of the present invention, and more particularly shows a more detailed embodiment of the variable conductance amplifier shown in FIG.

In FIG. 4, p-channel MOS transistors MI1a, MI1b and MI2a, MI2 are
Given constant gate voltages Vpa and Vpb, the first and second constant current sources I1 and I2 are formed. The n-channel MOS transistors MI3 and MI4 are supplied with a constant gate voltage Vna and are supplied with the third and fourth constant current sources I.
3 and I4 are formed. The n-channel MOS transistor MI5 forms the variable constant current source (fifth constant current source) I5 by being supplied with the variable control voltage Vc as the gate voltage. n-channel MOS transistor MI6a,
MI6b and MI7a, MI7b are supplied with constant gate voltages Vna, Vnb and are supplied with the sixth and seventh constant current sources I.
6 and I7 are formed.

P-channel MOS transistor M7a,
M8a, M7b, and M8b are connected in series with the set of M7a and M7b and the set of M8a and M8b, respectively, and the gate and drain of M7b are connected in common to M8b.
The first current mirror circuit 21 that operates so that the currents in the two current forming paths are equalized to each other by being connected to the gate of M8a and the gate and drain of M8a are commonly connected to the gate of M7a. To form.

P-channel MOS transistor M9a,
M10a, M9b, and M10b are connected in series to the set of M9a and M9b and the set of M10a and M10b, respectively, and the gate and drain of M9b are commonly connected to the gate of M10b. A second current mirror circuit 22 that operates so as to make the currents in the two current forming paths equal to each other is formed by being commonly connected and connected to the gate of M9a.

In the embodiment shown in the same figure, the forming currents in the current mirror circuits 21 and 22 are determined with high precision by mutual mirror control, whereby the precision of the variable conductance amplifier can be further improved.

In the figure, the MOS transistor M
B1, MB2, MB3, MB4a, MB4b, MB5
a, MB5b, MB6, and MB7 are for generating the gate voltage of the MOS transistor forming the constant current source.

Further, M1, M2, M3 to M6, MI1
b, MI2b, M7a, M7b, M8a, M8b, M9
a, M9b, M10a, M10b, MI6b, MI7
b, MB4b and MB5B are low threshold type MOS transistors, and the others are standard threshold type MOS transistors.

FIG. 5 shows an embodiment of a second-order Butterworth low-pass filter using the variable conductance amplifier according to the present invention, although not particularly limited thereto.

In the figure, 101 is a first variable conductance amplifier, 102 is a second variable conductance amplifier, and 103 is a third operational amplifier. A Gm control circuit (Gm control circuit) 201 variably controls the mutual conductance Gm of the variable conductance amplifiers 101 and 102. C is a first capacitive element, and 2C is a second capacitive element.

Each variable conductance amplifier 101, 10
2 respectively converts the potential difference between the output terminal and the non-inverting input terminal into a current by negative feedback from the output side to the inverting input (-) side, and charges and discharges the capacitance.

The first capacitive element C is the third operational amplifier 10
3 is connected between the output (Vout) and the non-inverting input (+) of the second variable conductance amplifier 102, and the second capacitive element 2C is connected between the non-inverting input (+) of the third operational amplifier 103 and the reference potential. Is connected in between.

As described above, the low-pass filter having the mutual conductance (Gm) of the variable conductance amplifiers 101 and 102 and the capacitance values of the capacitive elements C and 2C as time constant parameters is formed. The transfer characteristics of this low-pass filter, especially the cutoff frequency (cutoff frequency)
Can be appropriately variably adjusted by Gm control of the variable variable conductance amplifiers 101 and 102 by the Gm control circuit 201.

Therefore, when the low-pass filter shown in FIG. 5 is formed in a semiconductor integrated circuit, it is easy to correct variations caused in the manufacturing process of the semiconductor integrated circuit and to accurately match target characteristics. .

FIG. 6 shows a preferred application example of the variable conductance amplifier according to the present invention.

The application example shown in the figure is a mobile phone using wireless transmission, and 3 is a baseband unit and 5 is a wireless transmission / reception unit.

The baseband unit 3 is a first digital signal processing unit (DSP1) 3 for processing low frequency signals such as voice.
1. A second digital signal processing unit (DSP2) 32 for performing intermediate frequency signal processing such as modulation and demodulation, a low frequency A / D conversion unit 33 for digitizing an audio signal from a microphone, digitally processed D / A converter 34 for analog conversion to output the demodulated signal to the speaker, D / A for analog conversion of the digitally modulated intermediate frequency signal
A conversion unit 35, A / D conversion unit 36 for converting the intermediate frequency received signal into a digital signal for digital demodulation processing, A / D conversion unit 36
Harmonic signal (noise) before and after D conversion and D / A conversion
Low pass filters 37 to 40 for removing the above, and a logic processing unit (MPU) 4 for centrally controlling the entire operation.
1 and the like.

Here, the low pass filter sections 37 to 4 are
If 0 is configured by the variable conductance amplifier according to the present invention, it is possible to easily and optimally control the frequency transfer characteristic with low power consumption according to the manufacturing variations of the semiconductor and the ambient temperature.

Although the invention made by the present inventor has been specifically described based on the embodiments, the present invention is not limited to the above embodiments, and various modifications can be made without departing from the scope of the invention. Needless to say.

For example, the conductivity type of a MOS transistor may be switched between the p-channel and the N-channel depending on the power supply voltage polarity of the equipment used.

In the above description, the case where the invention made by the present inventor is mainly applied to the low-pass filter which is the field of application which is the background of the invention has been described, but the present invention is not limited to this, and for example, a high-pass filter or It can also be applied to bandpass filters.

[0073]

The effects of the typical ones of the inventions disclosed in this application will be briefly described as follows.

That is, it is possible to obtain an effect that it is suitable for low power consumption and that the mutual conductance can be smoothly changed while keeping the transmission distortion low.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing a first embodiment of a variable conductance amplifier to which the technique of the present invention is applied.

FIG. 2 is a circuit diagram showing a variable conductance amplifier according to a second embodiment of the present invention.

FIG. 3 is a circuit diagram showing a variable conductance amplifier according to a third embodiment of the present invention.

FIG. 4 is a circuit diagram showing a fourth embodiment of the variable conductance amplifier of the present invention.

FIG. 5 is a circuit diagram showing an embodiment of a low pass filter using a variable conductance amplifier according to the present invention.

FIG. 6 is a block diagram showing a configuration of a mobile phone which is a preferred application example of the variable conductance amplifier according to the present invention.

FIG. 7 is a circuit diagram showing a first example of a variable conductance amplifier examined prior to the present invention.

FIG. 8 is a circuit diagram showing a second example of a variable conductance amplifier examined prior to the present invention.

[Explanation of symbols]

 1 1st circuit R Ohm resistance element Vin Input current IGmc output current I1 to I4 1st to 4th constant current source I5 Variable constant current source (5th constant current source) M1 to M6 1st to 6th MOS transistor 2 2nd 21 current mirror circuit (first) 211 CMOS switch circuit 22 current mirror circuit (second) 101 to 103 variable conductance amplifier 3 baseband unit of mobile phone 4 wireless transmission / reception unit of mobile phone 37 to 40 low-pass filter

Claims (7)

[Claims] 1. An ohmic resistance element, a MOS transistor whose gate voltage is feedback-controlled so that a current flowing through the ohmic resistance element by the input voltage becomes a drain current, and a mutual transistor having the gate voltage of the MOS transistor as an input voltage. A variable conductance amplifier comprising a conductance control circuit. 2. An ohmic resistance element, first and second MOS transistors for applying an input voltage applied between two inputs, inverting and non-inverting, to the ohmic resistance element, and an ohmic resistance element based on the voltage. The third and fourth gate voltages are feedback controlled so that the forming current of the gate flows as the drain current.
And the fifth and sixth MOS transistors whose gate voltages are applied to the gate voltages of the third and fourth MOS transistors.
And a current mirror circuit for controlling the current flowing through each of the fifth and sixth MOS transistors to a predetermined mirror ratio, and the input voltage from the drain of the fifth or sixth MOS transistor. And a variable conductance amplifier which variably controls the rate of change of the output current according to the input voltage by variably controlling the currents flowing through the fifth and sixth MOS transistors. 3. A variable constant current source for controlling a common source current of the fifth and sixth MOS transistors, and a change in output current according to the input voltage by controlling a current formed by the variable constant current source. The variable conductance amplifier according to claim 2, wherein the rate is variably controlled. 4. A first current mirror circuit for controlling a current flowing through each of the fifth and sixth MOS transistors to a predetermined mirror ratio, and an output current extracted from the drain of the fifth or sixth MOS transistor. The variable conductance amplifier according to claim 2 or 3, further comprising a second current mirror for buffer transmission, wherein the output current is taken out through the second current mirror. 5. One of a fifth MOS and a sixth MOS
5. The transistor and the MOS transistor in the current mirror circuit for controlling the drain current of the MOS transistor are each composed of a plurality of MOS transistors connected in parallel via a switch circuit. A variable conductance amplifier according to claim 1. 6. An ohmic resistance element, first and second MOS transistors for applying an input voltage applied between two inputs, inverting and non-inverting, to the ohmic resistance element, and an ohmic resistance element based on the voltage. The third and fourth gate voltages are feedback controlled so that the forming current of the gate flows as the drain current.
And the fifth and sixth MOS transistors whose gate voltages are applied to the gate voltages of the third and fourth MOS transistors.
MOS transistor, a current mirror circuit for controlling currents flowing through the fifth and sixth MOS transistors to a predetermined mirror ratio, and the fifth and sixth MOS transistors.
A low-pass filter having a variable conductance amplifier having means for variably controlling a current flowing through a transistor, wherein a filter time constant is formed by the variable conductance amplifier and a capacitive element. 7. An ohmic resistance element, first and second MOS transistors for applying an input voltage applied between two inputs, inverting and non-inverting, to the ohmic resistance element, and an ohmic resistance element based on the voltage. The third and fourth gate voltages are feedback controlled so that the forming current of the gate flows as the drain current.
And the fifth and sixth MOS transistors whose gate voltages are applied to the gate voltages of the third and fourth MOS transistors.
MOS transistor, a current mirror circuit for controlling currents flowing through the fifth and sixth MOS transistors to a predetermined mirror ratio, and the fifth and sixth MOS transistors.
A mobile phone having a variable conductance amplifier including means for variably controlling a current flowing through a transistor, and a low-pass filter formed by the variable conductance amplifier and a capacitive element being provided in a signal processing path.