JPH0617796B2 - Equipment for detecting and measuring physical parameters - Google Patents

Description

Description: BACKGROUND OF THE INVENTION This invention relates to rotation sensors such as gyroscopes, and more particularly to fiber optic rotation sensors with extended dynamic range.

Fiber optic rotation sensors typically include a loop of fiber optic material into which light waves are coupled to propagate around the loop in opposite directions. The rotation of the loop creates a relative phase difference between the waves propagating in opposite directions according to the well-known "Sagnac effect", the amount of which corresponds to the speed of rotation. When the counter-propagating waves are recombined, they interfere constructively or destructively to produce an optical output signal whose intensity varies with the rotational speed of the loop. The rotation detection is
This is generally done by detecting this optical output signal.

Various techniques have been devised to increase the sensitivity of fiber optic rotation sensors even at low rotation speeds. However, most of these techniques cannot be used to detect very high rotational speeds because their output functions tend to occur repeatedly at different rotational speeds. As a result, its output signal is one of those possible rotational speeds with the same output signal waveform:
It cannot be used to determine if it may respond to the particular output signal waveform observed.

Thus, it would be a great improvement in the art to provide a phase modulation technique in which rotation can be accurately and reliably sensed over a very wide range of rotational speeds. Techniques and means for accomplishing that are described herein.

SUMMARY OF THE INVENTION The present invention provides an apparatus for sensing and measuring physical parameters using an optical loop. In this system, the light source provides counter-propagating light waves in the loop, and the detector is responsive to the counter-propagating light waves and, in particular, to the phase difference between the counter-propagating light waves. respond. This phase difference is shifted according to the physical parameter being measured. Such physical parameters include, for example, elastic waves, static pressure, dynamic pressure, temperature or the speed of rotation of the optical loop in the case of an optical gyroscope.

In this system, according to the invention, the phase difference modulator periodically biases the phase difference between the light waves that propagate in opposite directions in the loop to counteract the phase difference generated by the physical parameter. . The controller adjusts the periodic bias of the phase difference modulator in response to the output of the detector to eliminate the phase difference generated by the physical parameter. The circuit measures the physical parameter in response to the periodic bias of the phase difference modulator.

In the particular embodiment of the invention described herein, additional circuitry is used to periodically blank the detector output at the same frequency that the modulator periodically biases the phase difference in the loop. . In addition, a circuit connects the gating device to the control device and compares the output of the gating device with a reference signal at the second frequency. The light waves propagating in opposite directions in the optical loop are also modulated at a second frequency which is substantially higher than the first frequency.

In a particular implementation of the invention,
The phase difference modulator applies a substantially DC phase bias to the light waves propagating in opposite directions. This substantially DC phase bias is achieved by combining the output of the signal generator with the harmonic output of the same signal generator to produce a repetitive dump signal that is used to provide a periodic DC phase bias. Made

The present invention includes a rotation sensor and method of operation thereof for accurately and reliably detecting a wide range of rotation speeds.
Rotational sensors include all fiber optic components, such as fiber optic directional couplers, which (a) direct light from a light source in opposite directions around a sensing loop. It splits into two propagating light waves, and (b) combines the propagating light waves in opposite directions to provide an optical output signal. The correct polarization of the given light, the counter-propagating waves and the optical output signal is established, controlled and maintained by the fiber optic polarizer and the fiber optic polarization controller. A second fiber optic coupler is provided for coupling the optical output signal from the successive strands to a photodetector that outputs an electrical signal proportional to the intensity of the optical signal.

The improved operational stability and sensitivity of the rotation sensor is
Phase modulation of light waves propagating in opposite directions to each other at a first frequency (bias phase modulation frequency) using the phase modulator of (1), thereby biasing the phase of the optical output signal. A phase sensitive detection system is used to measure the first harmonic component of the intensity of the optical output signal. In the disclosed detection system, the amplitude of this first harmonic component is proportional to the rotational speed of the loop.

The second phase modulated signal is provided at any frequency much lower than that of the bias frequency. The second phase modulation frequency is preferably not harmonically related to the first phase modulation frequency. This second phase modulated signal may be imposed on the system via the first phase modulation, or alternatively a second phase modulator may be used. The optical output signal is gated synchronously with the second phase modulation frequency so that the output signal represents, for example, a phase difference modulation produced during the positive half cycle of the second phase modulation, while During the negative half cycle of 2 phase modulation, a value of 0 is presented. By adjusting the amplitude of the second modulation signal, the effect of the Sagnac phase shift can be effectively removed, whereby the time averaged value of the optical output signal during the positive half cycle of the second modulation. Becomes 0. Therefore, it can be seen that the amplitude of the second modulated signal is used as a means for eliminating the Sagnac phase shift and thus this magnitude represents the amount of Sagnac phase shift present in the system.

To adjust the magnitude of the second phase modulation signal, the first phase modulation frequency component in the gated portion of the output signal is fed back to the feedback error to control the amplitude of the second phase modulation drive signal. Used to generate a signal.

The feedback error correction modulator controls the amplitude of the second modulation drive signal according to the feedback error signal,
It corresponds to the amplitude of the optical output signal produced by the Sagnac phase shift.

Rotational speed data associated with the transfer function for the amplitude of the second phase modulation that removes the first phase modulation frequency component in the optical output signal caused by the rotation is stored in memory. The "elimination" amplitude of the lower frequency signal, which is sufficient to eliminate or limit the first phase modulation frequency component caused by the Sagnac effect, is then used to access the memory using the amplitude of the elimination signal as an address. Is converted into rotation speed by. The rotational speed data so accessed can be used directly or converted into a signal that can be interpreted to extract the speed of the Sagnac phase shift or rotation.

It has been found that the amplitude modulation of the odd harmonics of the optical output signal (directly or indirectly through polarization modulation) caused by the phase modulator may be removed by operating the phase modulator at a particular frequency. There is. Since the detection system used detects only odd harmonics (eg, the first harmonic), the effects of amplitude modulation induced by the phase modulator may be eliminated by operating at such frequencies. This eliminates a significant source of error in rotation sensing, and thereby increases the accuracy of the rotation sensor.

In another preferred embodiment of the present invention, the phase difference modulation is performed by modifying the phase modulation waveform.
Its amplitude has the same wavelength dependence for a given drive signal as the Sagnac phase shift has for rotational speed. Therefore, a substantially linear scale factor is created. Linearized scale factor
Then, the amplitude of the phase difference modulation at the second frequency is proportional to the Sagnac phase shift of the light waves propagating in opposite directions in the rotation sensor loop. In this embodiment, the phase difference modulation includes a substantially constant DC value, which is a simplified linear function of the DC phase difference modulation (sagnac phase shift) of the light waves propagating in opposite directions to each other. Can be adjusted to eliminate

One means for producing a substantially linear scale factor is to use the phase tilt imparted to the waves propagating in opposite directions by a modulator located at an asymmetric position in the sensing loop of the gyroscope. Applying a phase ramp creates a DC differential phase shift between waves propagating in opposite directions. However, commonly used fiber optic phase modulators cannot provide a phase tilt. Therefore, the phase ramp is simulated using a periodic repetitive waveform with a ramp.

One such waveform is a sawtooth, which simulates a sawtooth modulation waveform by combining the phase modulation of one frequency with the phase modulation of its second harmonic frequency. May be simulated by adjusting the phase and amplitude relationships to Since the reset process is performed at the peak of each sawtooth modulation waveform, and the two optical paths of the waves propagating in the opposite directions are related to each other, the phase difference cannot be timely made constant. This problem is overcome by first using a sawtooth waveform instead of the second modulated signal in the embodiment described above. With such a permutation, the optical output signal is gated during the time period when the ramp portion of the sawtooth waveform is present, and the output signal is not gated at all other times. Therefore, by adjusting the amplitude or frequency of the second modulation in the manner described above, the DC Sagnac phase shift is nulled out by the DC phase difference modulation created by the phase ramp when the output signal is gated. And a 0 sagnac phase shift is also simulated during the period when the output signal is not gated.

The gating of the output signal may be done at the light source or at the detector or after the detector. The slope of the phase slope that determines the differential phase shift is controlled by adjusting the modulation amplitude of the sawtooth modulated signal. This is achieved by using an error feedback signal and an error correction modulator as described above.

Of course, triangular tooth waveform phase modulation is also another type of waveform that can be used in the same manner as sawtooth waveform modulation. Such a triangular tooth waveform may be created by the combination of the modulation frequency and the third harmonic of the modulation frequency.

These and other advantages of the invention are best understood with reference to the accompanying drawings.

Detailed Description of the Preferred Embodiments Before proceeding with a discussion of the preferred embodiments of the present invention, a discussion of the basic rotation sensor used in the present invention is necessary to fully understand this improvement. FIG. 1 shows a rotation sensor having a basic structure of the type used in the present invention. It includes a light source 10 for directing light to a continuous, elongated or strand of optical fiber 12, a portion of which is wound into a sensing loop 14.
As used herein, reference numeral 12 generally designates an entire continuous strand of optical fiber, while reference numeral 12 with a suffix (A, B, C, etc.) indicates an optical fiber 12 Shows the part of.

In the illustrated embodiment, the light source 10 comprises a gallium arsenide (GaAs) laser that produces light having a wavelength on the order of 0.82 microns. According to a particular embodiment, the light source 10 is a general purpose light source at 3005 Hurdley Road, South Plainfield, NJ.
It may be a model GO-DIP laser diode, commercially available from Optronics Corporation. Fiber optic strands, such as strand 12, preferably have an outer diameter of, for example, 80 microns, and 4
It is a single mode fiber with a core diameter of micron. Loop 14 comprises a plurality of turns of fiber 12 wound around a spool or other suitable support member (not shown). According to a particular embodiment, loop 1
4 may have about 1000 turns of fiber wound on a foam having a diameter of 14 centimeters.

Preferably, the loop 14 is symmetrically wound starting from the center so that the points of symmetry of the loop 14 are close together. This appears to reduce the environmental sensitivity of the rotation sensor because such symmetry has a similar effect on both waves propagating in opposite directions with time-varying temperature and pressure gradients. Because.

Light from the light source 10 is optically coupled to one end of the fiber 12 by abutting the fiber 12 against the light source 10. The various components for guiding and processing light are positioned or formed at various locations along the continuous strand 12. For purposes of illustrating the relative placement of these components, continuous fiber 12 is described as being divided into seven sections, each labeled 12A-12G, which sections 12A-12E are Located on the side of the loop 14 that is coupled to the light source 10 and is part 12F and 12
G is on the opposite side of loop 14.

The polarization controller 24 includes fiber sections 12A and 12B.
And is adjacent to the light source 10. The type of polarization controller suitable for use as controller 24 has been assigned to the assignee of the present invention. Co-pending U.S. Patent Application Serial No. 183,975 (filed Sep. 4, 1980) is described in detail in "Fiber Optic Polarization Converter", which is incorporated herein by reference. A brief description of the polarization controller 24 will be given later. However, it should be understood here that this controller 24 allows adjustment of both the state and direction of polarization of a given light.

Fiber 12 is then coupled to fiber sections 12B and 1
It passes through the ports labeled A and B of the directional coupler 26 located between 2C. Coupler 26 couples the optical output to a second strand of optical fiber passing through the ports labeled C and D of coupler 26, port C being on the same side of the coupler as port A, and Port D is on the same side of the coupler as port B. The end of the fiber 28 extending from port D is "NC" (not connect).
The end of the fiber 29 extending from port C is optically coupled to the photodetector 30 while terminating non-reflectively at the point labeled ed). According to a particular example, the photodetector 30 may be a standard, reverse biased, silicon, PIN type photodiode. Combiner 26 is 1980
United States Patent Application Serial No. 139,511, "Fiber Optic Directional Coupler", a partial continuation application filed on April 11, 1994, filed September 10, 1981, co-pending patent Application serial number 300,95
5, "Fiber Optic Directional Coupler", both of which patent applications are assigned to the assignee of the present application. These co-pending patent applications are incorporated herein by reference.

Fiber portion 12C extending from port B of coupler 26
Passes through a polarizer 32 located between the fiber sections 12C and 12D. Single-mode optical fibers have two polarization modes in the path for any light wave. Polarizer 32 allows light to pass in one of the polarization modes of fiber 12 while blocking light in the other polarization mode. Preferably, the polarization control device 2 described above
4 is used to adjust the polarization of the given light,
Thereby, such polarization is substantially the same as the polarization mode passed by the polarizer 32. by this,
The loss of optical power is reduced when a given light passes through the polarizer. A preferred form of polarizer for use in this invention was assigned to the assignee of the present invention, 1980, 1
Co-pending United States Patent Application Serial No. 195,934, filed on Oct. 10, "Polarizers and Methods," is described in detail and incorporated herein by reference.

After passing through the polarizer 32, the fiber 12 is arranged between the fiber portions 12D and 12E, the directional coupler 3
4 through ports labeled A and B. This combiner 34 is preferably of the same type as described above with reference to combiner 26. The fiber 12 is wound on the loop 14 with a polarization controller 36 disposed between the loop 14 and the fiber portion 12E. The polarization controller 36 may be of the type discussed with reference to the controller 24 and is formed by the interference of these counter-propagating waves. The optical output signal is utilized to adjust the polarization of the light waves propagating in opposite directions through the loop 14 to have polarizations that are effectively passed by the polarizer 32 at the initial optical power loss. .
Thus, by using both polarization controllers 24 and 26, the polarization of the light propagating in fiber 12 can be adjusted for maximum optical power output.

Phase modulator 38 driven by AC signal generator 40
Is mounted in the fiber portion 12F between the loop 14 and the second directional coupler 34. The modulator 38 comprises a PZT cylinder around which the fiber 12 is wound. Fiber 12
Is coupled to the barrel so as to expand the fiber 12 as it expands radially in response to the modulated signal from the generator 40.

An alternative type of modulator (not shown) suitable for use with the present invention is a PZT cylinder that longitudinally extends four sections of fiber 12 that are coupled to double length capillaries at both ends of the cylinder. including. Those skilled in the art will recognize that such alternative forms of modulator may provide a lesser degree of polarization modulation to the propagating optical signal as compared to modulator 38, however, however, It will be appreciated later that the modulator 38 may be operated at a frequency that eliminates the unwanted effects of polarization modulation. Therefore, any type of modulator is suitable for use in the present invention.

The fiber 12 then passes through the ports labeled C and D of the coupler 34, and the fiber section 12F receives the port D.
And a fiber portion 12G extends from port C.
The fiber portion 12G is non-reflectively terminated at a point labeled "NC" (not connected).

The output signal from the AC generator 40 is provided on line 44 as a reference signal to a lock-in amplifier 46, which is also connected by line 48 to the photodetector 3
Connected to receive a 0 output. This signal on line 44 to amplifier 46 provides a reference signal which causes amplifier 46 to synchronously detect the detector output signal at the modulation frequency of modulator 38, ie the first harmonic component of the optical output signal. While blocking all other harmonics of this frequency.

Lock-in amplifiers are well known in the art,
It is commercially available.

The magnitude of the first harmonic component of the detector output signal is determined by loop 1
It will be seen later that a rotational speed of 4 is proportional over a limited operating range. Amplifier 46
Outputs a signal proportional to this first harmonic component and thus directly displays the rotational speed, which is indicated by the display panel 4
7 may be displayed visually. However, the first
The detection method shown in the figure can only be used for relatively small rotational speeds, as can be seen in the discussion of FIG.

Couplers 26 and 34 A preferred fiber optic directional coupler for use as couplers 26 and 34 in the rotation sensor or gyroscope of the present invention is shown in FIG. The coupler is a single mode fiber optic material having a portion of the cladding removed from one side thereof, 50A, 50B of FIG.
It includes two optical fiber strands indicated by. The two strands 50A and 50B are mounted in respective arcuate slots 52A and 52B formed in respective blocks 53A and 53B. The strands 50A and 50B are positioned at the portions of the strand where the cladding has been closely spaced so as to form an interaction region 54 where light is transferred between the core portions of the strands. The amount of material removed is such that the core portion of each strand 50A and 50B is within the evanescent field of the other. The center-to-strand spacing between the strands at the center of the coupler is typically less than about 2-3 core diameters.

It is important to note that the light transferred between the strands of interaction region 54 is directional. That is, substantially all of the light provided to input port A is provided to output ports B and D without being backward coupled to port C. Similarly, substantially all of the light applied to input port C is output ports B and D.
Given to. Moreover, this directionality is symmetrical. Therefore, light supplied to either input port B or input port D is provided to output ports A and C.
Moreover, the combiner is substantially non-discriminatory with respect to polarization, and thus preserves the polarization of the combined light. Thus, for example, if a light beam with vertically polarized light is input to port A, the light coupled from port A to port D will be similar to the light passing straight from port A to port B. It remains vertically polarized.

From the above description, the combiner functions as a beam splitter,
Two waves W1 propagating a given light in opposite directions
It can be seen that it is divided into W2 (Fig. 1). In addition, the combiner additionally serves to recombine the counter-propagating waves after they have crossed the loop 14 (FIG. 1).

In the illustrated embodiment, each of the combiners 26, 34 has five
It has a coupling efficiency of 0%, since the selection of this coupling efficiency gives the maximum light output at the photodetector 30 (FIG. 1). As used herein, the term "coupling efficiency" is expressed as a percent,
It is defined as the power ratio between the combined power and the total output power. For example, referring to FIG. 2, if light is provided to port A, the coupling efficiency will be equal to the ratio of the power at port D to the sum of the power outputs at ports B and D. Furthermore, the 50% coupling efficiency for the coupler 34 ensures that the waves W1, W2 propagating in opposite directions are of equal magnitude.

Polarizer 32 FIG. 3 shows a preferred polarizer for use in the rotation sensor of FIG. This polarizer is positioned within the evanescent field of the light carried by fiber 12.
A birefringent crystal 60 is included. The fiber 12 is mounted in a slot 62 that communicates with an upper surface 63 of a generally rectangular crystal block 64. The slot 62 has an arcuately bent bottom wall, and the fiber is mounted in the slot 62 so that it follows the contour of this bottom wall. Block 6
The upper surface 63 of 4 is overlaid to remove some of the cladding from the fiber 12 in region 67. The crystal 60 is mounted on the block 64 with the lower surface 68 of the crystal facing the upper surface 63 of the block 64 so as to position the crystal 60 within the evanescent field of the fiber 12. The relative refractive index of the fiber 12 and the birefringent material 60 is such that the wave velocity of the desired polarization mode is greater in the birefringent crystal 60 than in the fiber 12 and the wave velocity of the other undesired polarization mode is higher. Is larger in the fiber 12 than in the birefringent crystal 60. The light of the desired polarization mode is transmitted to the fiber 1
The light of the undesired polarization mode remains guided by the core part of the fiber 2, while the light of the undesired polarization mode is transmitted from the fiber 12 to the birefringent crystal 6.
Tied to 0. In this way, the polarizer 32 can pass light in one polarization mode, while preventing light from passing in the other polarization mode. As previously indicated, the polarization control devices 24, 36 (FIG. 1) each have the following features to minimize optical power loss by the polarizer:
It can also be used to adjust the polarization of a given light and optical output signal.

Polarizer Controllers 24,36 One form of polarization controller suitable for use in the rotation sensor of FIG. 1 is shown in FIG. The controller includes a base 70 on which a plurality of upright blocks 72A-72D are mounted. Between adjacent blocks 72, spools 74A-74C are mounted tangentially on shafts 76A-76C, respectively. The shafts 76 are axially aligned with each other and are rotatably mounted between the blocks 72. Spool 74
Is generally cylindrical and is tangentially positioned with respect to the shaft 76.

Strands 12 extend through axial bores in shaft 76 and are wound around each of spools 74 to form three coils 78A-78C. The radii of the coils 78 are such that the fiber 12 is stressed to form a birefringent medium in each of the coils 78. The three coils 78A to 78C are
Each is independently rotated about the axis of shafts 74A-74C to adjust the birefringence of fiber 12 and thus control the polarization of light passing through fiber 12.

The diameter and number of turns of the coil 78 depends on the outer coil 7
8A and 78C provide a quarter wavelength spatial delay, while the central coil 78B provides a half wavelength spatial delay. Quarter-wave coil 78
A and 78C control the eccentricity of the polarized light, and
The wavelength coil 78B controls the polarization direction. This gives the full range of polarization adjustment for the light propagating in the fiber 12.

The polarization direction (otherwise provided by the central coil 78B) may be indirectly controlled by properly adjusting the eccentricity of the polarization by two quarter-wave coils 78A and 78C, so that polarization control It will be appreciated that the device may be modified to provide only two quarter wave coils 78A and 78C. Accordingly, polarization controllers 24 and 36 are shown in FIG. 1 as including only two quarter wave coils 78A and 78C. This configuration reduces the overall size of the controller 24-36, which may be advantageous for certain space-constrained applications of the present invention.

Thus, the polarization controllers 24 and 36 provide a means for establishing, maintaining and controlling the polarization of both the given light and the waves propagating in opposite directions to each other.

Operation Without Phase Modulation or Polarization Control In order to fully understand the function and importance of the polarizer 32 (FIG. 1) and the phase modulator 38, for the operation of the rotation sensor, these components are first removed from the system. Explain as if it were. Thus, FIG. 5 shows, in schematic block diagram form, the rotation sensor of FIG. 1 with modulator 38, polarizer 32 and associated components removed therefrom.

Fiber 1 from the optical laser source 10 to propagate therethrough.
Is bound to 2. The light enters port A of coupler 26 where some of the light is lost via port D. The remaining part of the light propagates from port B of the coupler to port A, where it is split into two counter-propagating parallels W1, W2 of equal amplitude. Wave W1 propagates around loop 14 clockwise from port B, while wave W2 propagates around loop 14 counterclockwise from port D.

After the waves W1, W2 have traversed the loop 14, they are recombined by the combiner 34 to form an optical output signal, which propagates from port A of the combiner 34 to port B of the combiner 26. Part of the optical output signal is port B of the combiner 26.
From C to port C and propagates along fiber 29 to photodetector 30. The photodetector 30 outputs an electrical signal proportional to the intensity of light applied thereto by the optical output signal.

The strength of the optical output signal depends on the amount and form of interference between the waves W1, W2, ie constructive or destructive, when the waves W1, W2 are recombined or interfered in the combiner 34. ,Change. For now, ignoring the effects of fiber birefringence, the waves W1, W2 follow the same optical path around the loop 14. Therefore, assuming that the loop 14 is stationary, the waves W1, W2 are
When recombined at 4, they interfere constructively without any phase difference between them, and the intensity of the optical output signal is maximized. However, when the loop 14 is rotated, the counter-propagating waves W1, W2 are phase-shifted according to the Sagnac effect, so that when they are superposed in the combiner 34, they interfere destructively. To reduce the intensity of the optical output signal. Such a Sagnac phase difference between the waves W1, W2 caused by the rotation of the loop 14 is defined by the relationship:

Where A is the region coupled by the loop 14 of optical fiber, N is the number of turns of the optical fiber around that region A, and Ω is the angular velocity of the loop about an axis perpendicular to the plane of the loop. And λ and c are the free-space values of wavelength and velocity, respectively, of the light provided to the loop.

The intensity of the optical output signal (I _T) is a function of the Sagnac phase difference ([Delta] [phi _R) between the waves W1, W2, defined by the following equation.

Here, I ₁ and I ₂ are respectively waves W1,
It is the individual intensity of W2.

From equations (1) and (2) it can be seen that the intensity of the optical output signal is a function of the rotational speed (Ω). Thus, such an indication of rotational speed is obtained by measuring the intensity of the optical output signal using detector 30.

FIG. 6 shows a curve 80, which is the Sagnac phase difference (Δφ _R ) between the waves W1 and W2 propagating in opposite directions.
And the intensity ( _IT ) of the optical output signal.
The curve 80 has the shape of a cosine curve, and the intensity of the optical output signal is maximum when the Sagnac phase difference is zero. If the phase difference between the counter-propagating waves W1, W2 is caused entirely by the rotation of the loop 14, the curve 80 changes symmetrically about the vertical axis. However, polarized light, as discussed in co-pending US patent application serial no. 288,212, "Fiber Optic Rotation Sensor Using Unpolarized Light," filed July 29, 1981. Then, the waves W propagating in opposite directions
An additional, non-reciprocal phase difference between 1 and W2 may be caused by the residual birefringence of the optical fiber 12. This application is incorporated herein by reference. This additional non-reciprocal phase difference does not occur if completely unpolarized light is used.

Since the light traveling in each of the two polarization modes of the single mode fiber 12 travels at different speeds, there is no phase difference induced by birefringence. Birefringence couples some of the light traveling in one polarization mode into the other mode.
This creates a non-rotationally induced phase difference between the waves W1, W2, which causes the waves W1, W2 to interfere to distort or shift the curve 80 of FIG.
Such a shift is represented by the curve 8 shown in perspective in FIG.
Indicated by 2.

The non-reciprocal phase difference induced by such birefringence is indistinguishable from the rotationally induced Sagnac phase difference, and is an environment that changes the fiber birefringence such as temperature and pressure. Depends on the actual factor. Therefore, fiber birefringence is the main source of error in fiber optic rotation sensors.

Operation with Polarizer 32 The problem of nonreciprocal operation due to fiber birefringence is that polarizer 32 allows only one polarization mode to be used.
(FIG. 1) solves the problem in the rotation sensor of the present invention. When the polarizer 32 is introduced into the system at the point indicated by reference numeral 84 in FIG. 5, light passing through the polarizer 32 propagates to the loop 14 in one selected polarization mode. Further, the counter-propagating waves, when recombined to form the optical output signal, prevent any light that is not of the same polarization as the light provided to the loop from reaching photodetector 30. Is because the optical output signal passes through the polarizer 32. Thus, the optical output signal has exactly the same polarization as the light provided to the loop as it travels from port A of coupler 34 to port B of coupler 26.

Therefore, by passing the input light and the optical output signal through the same polarizer 32, only one optical path is used, which results in birefringence caused by different propagation velocities in the two possible polarization modes. Eliminates the problem of phase difference induced by. That is, by propagating all the light transferred by the birefringence in the fiber from the selected mode to the non-selected mode, the propagation speeds are different, thus gaining the phase for the selected mode or All light waves can be rejected in unselected modes that may be lost. Further, the polarizer control devices 24 and 36 (first
The figure) reduces the optical power loss in the polarizer 32 and thus maximizes the signal strength at the detector 30,
It should be noted that each can be used to adjust the polarization of a given light and optical output signal.

Operation with Phase Modulator 38 Referring again to FIG. 6, since the curve 80 is a cosine function, the intensity of the optical output signal is a small Sagnac phase difference (Δφ _R ) between the waves W1 and W2. It can be seen that it is nonlinear with respect to. Furthermore, the strength of the optical output signal is Δφ
_It is relatively insensitive to changes in phase difference for small values of _R. Due to such non-linearity and insensitivity, the optical intensity ( _IT ) measured by the detector 30.
To a signal representing the rotational speed of the loop 14 (Equation 1
It's hard to do).

Further, as described above using the polarizer 32, the waves W1,
The birefringence induced phase difference between W2 is eliminated, but there is cross-coupling between polarization modes caused by fiber birefringence. This cross-coupling prevents the cross-coupled light from reaching the photodetector 32 on the polarizer 32, so
Decrease the optical intensity of the optical output signal. Thus, due to changes in fiber birefringence, the amplitude of curve 80 in FIG. 6 changes, for example, as shown by curve 84. It will be appreciated that the curves 80, 82, 84 in FIG. 6 are not drawn to scale.

The aforementioned problems are solved by a sync detection system using the phase modulator 38, signal generator 40 and lock-in amplifier 46 shown in FIG.

Referring to FIG. 7, the phase modulator 38 modulates the phase of each of the propagating waves W1 and W2 at the frequency of the signal generator 40. However, as seen in FIG. 1, the phase modulator 38 is located at one end of the loop 14. Thus, the modulation of wave W1 does not necessarily have to be in phase with the modulation of wave W2. In fact, for proper operation of this sync detection system, the modulation of the waves W1, W2 is preferably 180 ° out of phase. Referring to FIG. 7, the modulation of the wave W1 represented by the sinusoidal curve 90 is represented by the wave W represented by the curve 92.
It is preferably 180 ° out of phase with the modulation of 2. Using a modulation frequency that provides such a 180 ° phase difference between the modulation of wave W2 and the modulation of wave W1 means that it is induced by the modulator in the optical output signal measured by detector 30. It is particularly advantageous in that it eliminates amplitude modulation. The modulation frequency (f _m) is calculated using the following equation.

Here, L is waves W1 and W propagating in opposite directions.
The differential fiber length between the coupler 34 and the modulator 38 for 2, ie the distance between the modulator 38 and the point of symmetry on the other side of the loop 14, measured along the fiber, n _eq is the equivalent refractive index for the single mode fiber 12 and c is the free space velocity of the light provided to the loop 14.

"Suitable" This modulation frequency is called frequency _(f m)
And the phase difference (Δφ ₁ ) between these waves W1 and W2, which branch from the phase modulation of these counter-propagating waves W1 and W2 according to curves 90 and 92, is a sinusoidal curve in FIG. Denoted at 94. This curve 94 is obtained by subtracting curve 92 from curve 90 to obtain the phase difference between W1 and W2. Waves W1, modulation of the phase difference between W2 also just Sagnac phase shift modulate the intensity of the optical output signal (I _T) according to Figure 6 of curve 80, as is the case, since such phase modulation Δφ ₁ is the rotationally induced Sagnac phase difference Δφ
_This is because it cannot be distinguished from _R.

(A) Phase modulation Δ defined by the curve 94 in FIG.
and phi _1, by referring to FIG. 8 and FIG. 9 shows schematically the effect of the Sagnac phase difference △ phi _R when the (b) the intensity of the optical output signal (I _T), the foregoing description Be better understood. Before proceeding with the discussion of FIGS. 8 and 9, the intensity of the modulated optical output signal ( _IT ) is
It should be understood that it is a function of the total phase difference between 1 and W2. Such a total phase difference consists of both the rotationally induced Sagnac phase difference Δφ _R and the phase difference Δφ ₁ induced by the time-varying modulation. Wave W1, W
The total phase difference Δφ between the two is expressed as follows.

Δφ = Δφ _R + Δφ ₁ (4) Therefore, not only the rotationally induced phase difference Δφ _R, but also the effect of the phase difference Δφ ₁ induced by modulation is shown in FIGS. 8 and 9. As will be considered with reference to the figures, the horizontal axis for curve 80 is that the total phase difference is considered, rather than only rotationally induced phase differences, as in FIG. In order to indicate that, a symbol is newly added as Δφ.

Now referring to FIG. 8, the effect of the phase modulation Δφ ₁ (curve 94) on the intensity I _T of the optical output signal will be discussed. Curve 8
Zero indicates the relationship between the optical output signal strength resulting from two interfering coherent waves and the phase difference Δφ between these waves. The relative phase angle between them is 193
When 0, the resulting intensity of the combined wave is maximum, as shown at 195. When the relative phase between the waves W1 and W2 is non-zero, the combined optical signal has a smaller intensity depending on the magnitude of the phase difference Δφ. The relative phase difference is 19
It continues to decrease as Δφ increases until it becomes + or −180 ° as shown by 7 and 199. With a phase difference of + or -180 °, the two counter-propagating waves interfere completely destructively and the resulting intensity is 0, as indicated by 197 and 199.

In FIG. 8 it is assumed that the loop 14 is stationary and thus the optical signal is unaffected by the Sagnac effect. In particular, it can be seen that the phase difference curve 94 induced by the modulation causes the optical output signal to change as shown by curve 96. The curve 96 is
Obtained by transforming the point on curve 94, which represents the instantaneous phase difference Δφ ₁ between W1 and W2, onto curve 80, which represents the resulting optical intensity for a phase difference of that magnitude. . All points on curve 94 are curve 80
Once converted up and the corresponding intensities plotted, the curve 96 results. The transition through curve 80 to curve 94 is symmetrical with respect to the vertical axis of curve 80 so that the optical intensity measured by detector 30 is
As indicated by 6, it varies periodically at a frequency equal to the second harmonic of the modulation frequency.

As described above, the lock-in amplifier 46 serves as the signal generator 4
Since it is tuned by the reference signal from 0 (FIG. 1) at the modulation frequency f _m , the lock-in amplifier synchronously detects only the detector output signal at the modulation frequency f _m of the modulator 38, ie the first harmonic. To do. However, because the detector output signal is at the second harmonic of the modulation frequency, as shown by curve 96, the output signal from amplifier 46 is zero and display 47 shows a rotational speed of zero.

Even though the birefringence-induced amplitude variation occurs in the optical output signal, as discussed with reference to curve 84 in FIG.
It should be noted that the curve 96 in FIG. 8 remains at the second harmonic frequency. Therefore, the fluctuation of the amplitude induced by such birefringence has no influence on the output signal of the amplifier 46. Therefore, the detection system described thus far provides a substantially more stable operating point that is insensitive to changes in birefringence.

When the loop 14 is rotated, the counter-propagating waves W1, W2 are phase-shifted as described above according to the Sagnac effect. The Sagnac phase shift provides a constant phase difference Δφ _R for a constant rotational speed. This Sagnac phase shift increases the phase difference Δφ ₁ produced by the modulator 38 so that the total curve 94 becomes
As shown in FIG. 9, an amount equal to Δφ _R
The phase is shifted from the position shown in. This causes the optical output signal to change asymmetrically along the curve 80 between points 99 and 101. This produces an optical output signal as shown by curve 96.

The points on curve 96 are derived as follows. The combined phase difference indicated by 103 on curve 94 moves through point 101 on curve 80 to point 105 on curve 96. Point 107 on curve 94 moves to point 111 on curve 96 via point 109 on curve 80. Similarly, point 113 moves to point 115 via point 99 and point 117 moves to point 119 via point 109. Finally, point 121 moves to point 123 via point 101.

The optical output signal 96 has a first harmonic component, as shown by the perspective of sinusoidal curve 98. The peak amplitude of the first harmonic component 98, however, need not exactly match the amplitude of the optical output signal at point 115, but in some cases it may.

It will be seen later that the RMS value of this sinusoidal curve 98 is proportional to the sine of the rotationally induced Sagnac phase difference Δφ _R. Since amplifier 46 synchronously detects the signal having the fundamental frequency of modulator 38, amplifier 46 outputs a signal proportional to the RMS value of curve 98. This signal can be used to indicate the rotational speed of the loop.

The drawing of FIG. 9 shows the intensity waveform of the optical output signal for one direction of rotation of the loop 14 (eg clockwise). However, if the loop 14 is rotated at the same rate and in the opposite direction (eg, counterclockwise), the intensity waveform 96 of the optical output signal will cause the curve 98 to shift 180 degrees from the position shown in FIG. It will be appreciated that it would be exactly the same as shown in FIG. 9 except that it would be translated as described.

The lock-in amplifier 46 detects this 180 ° phase difference with respect to the curve 98 by comparing the phase of the first harmonic 98 with the phase of the reference signal from the signal generator 40 and the rotation of the loop counterclockwise. Determines if it is clockwise. Depending on the direction of rotation, the amplifier 46 outputs either a positive or negative signal to the display device 47. However, regardless of the direction of rotation, the magnitude of the signal is loop 1
The same is true for equal speeds of rotation of four.

The waveform of the output signal of the amplifier is shown as curve 100 in FIG. This curve 100 is a sine wave and loop 14
12 depending on whether the rotation is clockwise or counterclockwise
It can be seen that the zero rotation speed output voltage indicated by 5 changes in the positive direction or in the negative direction. Furthermore, the curve 100
Has a substantially linear part 102, which part 102
Provides a relatively wide range of motion for measuring rotation that changes symmetrically with respect to the origin. Furthermore, the curve 10
A zero slope provides excellent sensitivity through its linear operating range 102, even for small Sagnac phase shifts.

Thus, by using a sync detection system,
The above-mentioned problems of non-linearity, insensitivity to small Sagnac phase shifts, and amplitude fluctuations induced by birefringence can be seen at some point in the curve 80 between points 97 and 95, at points 99 and 101 in FIG. Loop 14 to maintain
Is reduced or eliminated for the rotational speed of.

Thus, a further advantage of the previously disclosed detection systems is that, in the state of the art, phase modulators such as modulator 38 induce amplitude modulation in the optical output signal, either directly or indirectly through polarization modulation, That is, it relates to the fact that the phase modulator also shifts some of the light passing through it into unselected polarization modes. However, as will be recalled from the discussion made with reference to equation (3), by operating at a specific or "appropriate" frequency where the phase difference between the modulation of waves W1 and W2 is 180 °, The odd harmonic frequency components of this amplitude modulation, induced in each of the counter-propagating waves W1, W2 by 38, cancel each other out when the waves are superposed to form the optical output signal. Therefore, the detection system described above detects only the odd harmonics of the optical output signal, ie the fundamental frequency, thus eliminating the effects of unwanted amplitude modulation. Therefore, by operating at a particular frequency defined by equation (3), and by detecting only the odd harmonics of the optical output signal, the rotation sensor of the present invention provides a modulator-induced amplitude and polarization. It can operate independently of modulation.

Yet another advantage of operating at the proper frequency is that the modulator 38 in each of the counter-propagating phases W1, W2.
Induced by. The even harmonics of the phase modulation are that they cancel when they are superimposed to form the optical output signal. These even harmonics, by superimposing, generate spurious odd harmonics in the optical signal that might otherwise have been detected by the detection system, so their removal could improve the accuracy of rotation detection. Become.

Phase modulator 3 at the frequency defined by equation (3)
In addition to activating 8, it is also preferable to adjust the magnitude of the phase modulation so that the amplitude of the detected first harmonic of the intensity of the optical output signal is maximized, since this is the sensitivity of rotation detection. And to improve accuracy. Figure 7,
The optical output when the amplitude of the modulator-induced phase difference Δφ ₁ between the waves W1, W2 is 1.84 radians, as indicated by the dimension labeled Z in FIGS. 8 and 9. It has been found that the first harmonic of the signal strength is at a maximum for a given rotational speed. This is the total intensity ( _IT ) of two superposed waves with individual intensities of I ₁ and I ₂ , respectively, with a phase difference of Δφ.
It may be more fully understood by reference to the following equation for.

Here, Δφ ₁ = Δφ _R + Δφ ₁ (6) Δφ ₁ = Z sin (2πf _m t) (7) Therefore, Δφ = Δφ _R + Z sin (2πf _m t) ( 8) The Fourier expansion of cos (Δφ) is as follows.

Here, J _n (z) is the n-th order Bessel function of the variable z, and z is the peak amplitude of the phase difference induced by the modulator between the waves W1 and W2.

Therefore, the following equation is obtained by detecting only the first harmonic of I _T.

Thus, the amplitude of the first harmonic of the intensity of the optical output signal depends on the value of the first Bessel function J ₁ (z). J
_{Since 1} (z) is maximum when z equals 1.84 radians, the amplitude of the phase modulation is preferably the waves W1, W2.
In between, the amplitude (z) of the phase difference Δφ ₁ induced by the modulator must be chosen to be 1.84 radians.

Reduction of Backscattering Effects As is well known, current state of the art optical fibers are not optically perfect and have imperfections such as variations in density in the fiber's base material. These imperfections cause variations in the index of refraction of the fiber, which results in a small amount of light scattering. This phenomenon is generally called Rayleigh scattering. Due to such scattering, some light is lost from the fiber, but such loss is relatively small,
Therefore it is not the main concern.

The main problem associated with Rayleigh scattering is not with the scattered light lost, but rather with the light reflected back in the fiber in the direction opposite to the original direction of propagation. This is commonly referred to as "backscattered" light. Since such backscattered light is coherent with light composed of counter-propagating waves W1, W2, it must interfere constructively or destructively with such counter-propagating waves. And thereby causes variations in the intensity of the optical output signal as measured by detector 30.

The portion of the backscattered light from one wave coherent with the wave propagating in the opposite direction is scattered within the coherency length of the center of the loop 14. Therefore, by reducing the coherency length of the light source, the coherency between backscattered light and waves propagating in opposite directions is reduced. The rest of the backscattered light is not coherent with the waves propagating in opposite directions to each other, and therefore the interference between them varies randomly as it is averaged. Therefore, the incoherent portion of the backscattered light is of substantially constant intensity, and thus it does not cause any significant variation in the intensity of the optical output signal.

Therefore, in the present invention, the influence of backscattering is
It is reduced by using as the light source 10 a laser having a relatively short coherency length, for example 1 meter or less. According to a particular example, the light source 10 is, as described above, a general optronics light source.
Model GO-commercially available from Corporation
It may include a DIP laser diode.

An alternative way to inhibit destructive or constructive interference between backscattered and propagating waves is to include an additional phase modulator in the system at the center of fiber loop 14. . This phase modulator is not synchronized with modulator 38.

The propagating wave passes through this additional phase modulator only once as it travels around the loop. In contrast to the backscatter that results from the wave propagating before the wave reaches the additional modulator, that backscatter will not be phase modulated by this additional modulator because the light source propagating the wave also This is because the backscatter itself did not pass through the additional modulator.

On the other hand, for backscattering resulting from a wave propagating after the wave has passed through this additional phase modulator, once when the propagating wave has passed through the additional phase modulator, and Once the backscatter passes through the additional modulator, the backscatter will effectively be phase modulated by a factor of 2.

Thus, if the additional phase modulator induces a phase modulation of φ (t), the backscattered wave starting at any point, except at the center of loop 14, will have 0 or 2
Any of them has a phase shift of φ (t), which is time varying with respect to the propagating wave with respect to the φ (t) phase shift. This time-varying interference is time-averaged, effectively eliminating the effects of backscatter.

In yet another alternative method of inhibiting destructive or constructive interference from backscatter, an additional phase modulator, not synchronized with modulator 38, may be induced at the output of light source 10.

In this case, the backscatter that occurs at any point other than the center of the loop 14 has a different optical path length from the light source 10 to the detector 30, unlike the backscattered propagating wave.

Thus, the propagating wave will traverse the loop 14 once, but the backscattered wave and the propagating wave resulting therefrom will traverse a portion of the loop 14 twice. If this part is not half of the loop, the path length is different.

Because of the different path lengths, the propagating waves that reach the detector 30 will be generated at the light source 10 at different times than the backscattered waves that reach the detector 30 at the same time.

The phase shift induced in the light source 10 by the additional phase modulator is the phase shift φ (t) for the propagating wave.
, Which induces a phase shift of the phase shift φ (t + K) for backscattered waves, where K is the time difference between the paths of the waves passing through the modulator. φ (t +
Since K) is time varying with respect to φ (t), the backscattered interference is averaged over time, effectively eliminating the backscattering effect.

Extended Dynamic Range Detection System Using Gated Waves The detection system described above with reference to FIGS. 1-10 is a very effective rotation sensing system within a range of rotational speeds for loop 14. Is. However, the dynamic range is limited by certain phenomena. With reference to FIG. 9, it can be seen that the curve 80 is periodic.
Therefore, if a large rotational speed causes Δφ _R to be large enough to move curve 94 through either point 97 or point 95, then function 96 will yield a second, higher rotational speed. Repeat. This second rotational speed is substantially greater than the rotational speed that produced the Sagnac phase shift Δφ _R shown in FIG. 9, but would be indistinguishable from the lower speed using the output optical signal 96. . That is, one could move curve 94 such that Δφ _R from some higher rotational speed was sufficiently large to operate between two new points 99 'and 101' on the second protrusion of curve 80. For example, the output optical signal 96 would be indistinguishable from the case shown in which curve 94 operates between points 99 and 101 in such a case.

The present invention includes a novel method and associated apparatus for extending the detection range of a fiber optic gyroscope. Upon achieving this method, an optical fiber gyroscope described above, opposite directions through the above-mentioned "appropriate" frequency or bias frequency (f _m) much lower additional frequency level than (f _m) by Equation 3 Is modified to include the modulation of the light wave propagating in the.

In an asymmetrically arranged reciprocal phase modulator in the sense loop, applying a signal to the modulator creates a differential shift Δφ _c between the phases of two counter-propagating waves in the loop. Can be This △ φ
_c is the modulation frequency f _c and changes with time, and what DC
It also does not include the condition, because the phase shift produced by the modulator in one half of its modulation cycle is canceled by the one produced in the next half cycle.

In contrast, the differential phase shift Δφ _R caused by rotation is D
It may be a C amount, so Δφ _c cannot be used directly to eliminate Δφ _R.
However, if the gyroscope is gated off every other half-cycle of the modulation waveform, the average Δφ _c produced in the remaining half-cycles will directly affect the signal produced by rotation in these same half-cycles. It can be used to eliminate. By monitoring the amplitude of the signal that produces Δφ _c during the gate-on half cycle, the rotational speed of the sensor can be determined.

As mentioned above, by imposing an additional modulation frequency f _m, which is much lower than the bias phase modulation frequency, at the bias frequency f _m for biasing the operating point, and then another half cycle of the f _m waveform. By gating the gyro off every time, a phase difference modulated waveform is created, the time average of which has a net DC level. By adjusting the amplitude of the second phase modulation at the frequency f _m , this time-averaged DC value of the phase difference modulation may be adjusted to effectively eliminate the Δφ _R generated in these same half cycles. Good. The techniques described above serve to effectively eliminate the effects of rotation on the output signal. Because no rotation is discerned during the time period when the gated signal is off, and the effect of Δφ _R is canceled by phase-modulating at the frequency f _m when the gated signal is on. To be done. Rotation amount is proportional to the amplitude of the phase difference modulation at f _m was necessary to eliminate the influence of △ phi _R, the rotational speed can be determined by monitoring the amplitude of the second modulated signal You can This method is described in more detail below with respect to the description of the apparatus used to implement the method.

Referring to FIG. 11, a preferred embodiment of an apparatus is provided which, when used in connection with the method described herein, provides a significant increase in detection range and improves the reliability of the results provided by such detection. To be The detection system of FIG. 11 implements many of the components of the system shown in FIG. Therefore, for simplicity,
Corresponding reference numerals are assigned to those components of FIGS. 1 and 11 that have the same structure and function.

In the circuit of FIG. 11, the optical output signal from detector 30 is transmitted via line 48 through amplifier 300, where its intensity is magnified so that it can be fully used in electrical circuits. It Amplifier 3
From 00, the output signal passes through line 302 to a conventional electronic gate 304. The operation of gate 304 is controlled by the gating signal received from AC signal generator 308 on line 306. The phase of the signal on line 306 can be adjusted using a conventional phase delay device on line 306.

Signal generator 308 generates a second phase-modulated signal at the frequency f _m, although this frequency f _m is selected arbitrarily, as described above, "appropriate" is typically at the frequency f _b is set Must be much lower than that of bias phase modulation.

The signal from gate 304 is synchronously gated onto line 310 at a second phase modulation frequency f _m generated in signal generator 308. The signal is then transmitted to bandpass filter 312, which passes only the f _b frequency component of the signal received on line 310 onto line 314. The signal at frequency f _b on line 314 represents the amount of rotation experienced by loop 14 in the absence of any other signal to polarize its magnitude.

As described below, the signal on line 314 is used with lock-in amplifier 46 to generate a feedback signal that controls the amplitude of the second phase modulation at frequency f _m . With the appropriate amplitude adjustment of this second phase difference modulation, a signal is generated, which signal causes the phase modulator 3
8 affects the waves propagating in opposite directions in the loop such that the signal at frequency f _b on line 314 is driven in a time average, towards zero, regardless of loop rotation speed.

The signal on line 314 is transmitted to lock-in amplifier 46 to generate the feedback signal described above.
In addition, the lock-in amplifier receives a reference signal from line 316 corresponding to the bias modulation frequency f _b generated by the AC signal generator 40. Generally, this frequency f
_b corresponds to the "appropriate" frequency, as previously calculated using equation (3).

In response to the signals received on lines 314 and 316,
The lock-in amplifier 46 produces an "error signal" which is proportional to the amplitude of the input signal from line 314 and matches the frequency of the reference signal from line 316. This error signal is the curve 10 in FIG.
Lying somewhere in 0. In a particular case, the error signal is some D on curve 100 for a fixed rotational speed that produces a fixed amplitude of the first harmonic component on input line 48.
It will be C level. If the amplitude of the first harmonic component changes, the DC level of the error signal is 100 ° at the operating point.
It will change as you shift along.

As explained previously, without the second phase modulation at the frequency f _m , the curve 100 would be periodic, since the curve 80 of FIG. 9 is periodic. Therefore, the optical output signal 9
The magnitude of the f _b frequency component of 6 will change periodically as increasing the Sagnac phase shift pushes the entire phase shift curve 94 to the other protrusions of curve 80. That is, on curve 100, 134 (FIG. 10) is such that the resulting global phase shift curve maxima and minima transition through the symmetrically balanced points on the second protrusion of curve 80. The Sagnac phase effect represents the case of pushing the curve 94 far enough. The resulting output waveform 96 will appear similar to the output optical signal 96 depicted in FIG. 8 for the zero rotation case and will have no first harmonic component. Wave 9
Since 6 has no first harmonic component in this case, the output of the lock-in amplifier will be zero despite the fact that the rotational speed is not zero.

The detection system of the present invention addresses this problem by using the feedback error signal by adjusting the amplitude of the second phase modulated signal at frequency f _m in response to changes in the first harmonic signal on line 314. Has been resolved.
The adjusted second phase-modulated signal is then phased with the opposite propagating waves of the loop so that the signal at frequency f _b on line 314 is effectively canceled, as described below. Used to adjust the modulation. As a result, even if the rotation is such that the Sagnac phase shift is pushed to point 134 on curve 100 of FIG.
The amplitude of the second phase modulated signal, in the absence of such feedback, provides a readiness indication of the actual rotational speed at the high speed that would place curve 94 beyond the point indicated by 134 in FIG.

The function of adjusting the amplitude of the second phase modulation signal in response to the feedback error signal is performed by the error correction modulator 130. To achieve this, error correction modulator 130 uses lock-in amplifier 4 via line 318.
6 receives the error signal, and also on line 320,
The second modulated signal from the signal generator 308 is received. Preferably, the second modulated signal defines a sinusoidal waveform.

Upon receiving a non-zero error signal on line 318, error correction modulator 130 may reduce the magnitude of the error signal on line 318 to zero, or to reduce it within a predetermined zero range. , Increase or decrease the amplitude of the second phase modulation signal in response to the magnitude and sign of the error signal. When the predetermined level for the error signal on line 318 is reached, the modulator 130 maintains the amplitude of the second phase modulated signal until the error signal changes again.

Upon detecting a change in the error signal, the modulator 130 again outputs the second signal until the error signal on line 318 is again reduced to zero or within a predetermined zero range.
The amplitude of the phase modulation signal of is changed. The adjusted second phase modulation signal is output from the error correction modulator 130 to the line 322.
Transmitted up. The feedback approach described here is also applicable to other types of gyroscopes, such as those made from highly birefringent fibers.

The adjusted second phase modulation signal on line 322 is combined with the bias modulation signal from signal generator 40 on line 324. This combined signal from line 324 is provided to phase modulator 38 to affect the counter-propagating waves and thus the output signal from detector 30 in accordance with the method described above.
As described above, the second phase modulation signal serves to bias the phase difference between the light waves propagating in the opposite directions in order to substantially eliminate the phase shift generated in the phase difference between the light waves propagating in the opposite directions due to the rotation speed. do. In this context, the bias provided by the second phase-modulated signal not only serves to compensate the component of the output signal at frequency f _b caused by the rotational speed, but it also affects the phase difference signal generated by that rotational speed. Effectively eliminates, thereby removing its associated component from the output signal.

The speed of rotation may be determined by using the output display 208 connected to the line 324 via a bandpass filter 326. In particular, the modulated signal from line 324 is connected by line 330 to a filter 326 that can only pass signals at the second modulation frequency f _m . The signal from filter 326 is on line 33.
2 to the output display device 208. Display device 20
The signal on 8 corresponds to the amplitude of the second modulated signal of frequency f _m and may therefore be used to determine the speed of rotation. The display 208 and associated circuitry for determining rotational speed are described in detail below.

By referring to FIG. 12, as a result of loop rotation and phase modulation in the apparatus shown in FIG.
The resulting relative phase shift experienced between waves propagating in opposite directions can be illustrated graphically. 12th
In the figure, the optical output signal (not shown) taken at the photodetector 30 is the resulting or total phase shift curve 35.
0, this curve 350 includes a Sagnac phase shift Δφ _R (indicated by a constant bias 352 for a constant rotational speed) and a sinusoidally time-varying second phase difference modulation signal Δ indicated by curve 354. φ _m (cos
ω _m t) and the bias phase difference modulation difference signal Δφ _b (cos ω _b t) which changes sinusoidally with time are represented. The resulting phase shift Δφ is therefore defined as

△ phi = the time average value of _{(△ φ b cos ω b t} ) + (△ φ m cos ω m t) + △ φ R ... (11) △ φ is approximately 0 by gating off a portion of the signal Can be adjusted to a value. Therefore, as illustrated in FIG. 12, the second of the frequency f _m is
All other half-cycles of the phase-difference modulation 354 of <RTIgt; By adjusting the amplitude of the second phase difference modulation 354 in this situation, the portion of the bias modulation signal 350 that is gated on can be positioned with respect to the vertical axis 355.

In the circuit shown in FIG. 11, the gate 304 is turned on and off in synchronization with the second phase modulation signal by using the gate signal from the signal generator 308. Thus, the waveform of FIG. 12 can be generated with the gate signal on line 306 synchronized to switch gate 304 at each half cycle of the second phase modulation signal frequency f _m . It will be appreciated that a value of 0 is present at the output of detector 30 during the time the signal is gated off, as shown in FIG. 12 (FIG. 11).

The output signal at frequency f _b generated as a result of the above conditions is shown at 360 in FIG. Note that for purposes of illustration, line 360 is not drawn to scale with respect to the second phase modulation waveform 354. Therefore, the frequency f
It can be seen that the rotational speed of the loop can be determined by adjusting the amplitude of the second phase modulated signal 354 until the time averaged value of the gated portion of the output signal of _b is equal to zero. In particular, the rotation speed is determined by observing the amplitude of the second phase modulation signal that produced the O error signal.

If no modulation signal, the detector output amplitude I _b of the bias modulation frequency f _b can be mathematically explained as follows.

I _b = CP ₀ J ₁ (Δφ _b ) sin Δφ (12) where C is a constant, P ₀ is the optical power incident on the detector, and Δφ _b propagates in opposite directions. Is the amplitude of the phase difference modulation between the waves, J ₁ is the first-order Bessel function of the first kind, and Δφ is the phase difference between the waves propagating in opposite directions of the sensing coil.

The second phase modulated signal is further applied with a frequency f _m much lower than the bias modulation frequency f _b , and the waveform of the phase difference modulation in the presence of the rotation-induced non-reciprocal phase shift Δφ _R is This is as shown in FIG. When the signal from photodetector 30 is switched off for 50% of each cycle of phase modulation of frequency f _m ,
The demodulated output power of the bias modulation frequency f _b is the first
It is shown in FIG. The demodulated output power can be zeroed by adjusting the amplitude of the phase difference modulation Δφ _m , provided that the lock-in amplifier integrates its signal over many cycles of the phase modulation of frequency f _m. it can. This means that non-reciprocal phase shifts induced by rotation can be offset with respect to time average by phase modulation with gating. The demodulated output power is illustrated at 360 in FIG. This output signal 36
The time average of 0 is explained as follows.

Here, T = 1 / f _m ; ω _m = 2πf _m ; and Δ _m are the amplitudes of the phase difference modulation at the frequency f _m .

And size [Delta] [phi _m of the second modulated signal of the demodulated power at the frequency f _m to zero, Sagnac phase shift △ phi
The relationship with _R can be obtained from the equation:

Here, J _n is an n-th order Bessel function.

It is described by equation 14. Second for the value of Δφ _R
The amplitude relationship of the phase difference modulation is shown diagrammatically in FIG. Curve 370 of FIG. 14 shows the response of the sensor to rotation when the gyro is operated electronically in a closed loop configuration. Curve 370 is a transfer function or scale factor that describes the amplitude of the second phase difference modulation Δφ _m required to bring the Sagnac phase shift (Δφ _R ) to substantially zero in the gated device shown in FIG. Is shown graphically.

It can be seen that the scale factor of FIG. 14 has a monotonic characteristic that provides a dynamic range for gyroscopic operation which is limited only by that of the phase modulator used.
The small deviation of the magnification curve 370 from perfect linearity is
This results from the fact that the net non-reciprocal phase shift is averaged to zero with a time varying phase shift instead of a DC phase shift. Thus, the present invention is higher than prior art sensors would be unable to know which of several possible rotational speeds are producing the detector output with its particular characteristics. Means are provided for disambiguating the detector output signal with respect to rotational speed.

Since the frequency of the second phase modulated signal at frequency f _m is arbitrary, it is necessary that its frequency and phase have no fixed relationship to the bias phase modulation operated at the appropriate frequency. As a result, the components that generate and control the two excitation signals are less demanding on stability. Furthermore, the lack of a frequency and phase relationship between the two modulations makes it possible to apply them to signal phase modulators without compromising the electrical combination of the two excitation signals and the sensitivity of the rotation sensor. Become.

Some of the other components of the device of Figure 11 are described in more detail below.

FIG. 15 shows an embodiment of the error correction modulator 130.
In this embodiment, the error signal on line 318 is coupled to the inverting input of an operational amplifier connected as an integrator. The exact structural details of practical integrators are well known to those skilled in the art and will not be described in further detail.

As is well known in the operational amplifier art, the negative feedback voltage across the capacitor tends to keep point 170 at virtual ground. That is, the voltage at point 170 is held at or near 0 volts by negative feedback. However, no current flows to ground due to this virtual short circuit. Impedance R ₀
The input current i _in flowing through the output impedance of the lock-in amplifier 46 to the operational amplifier 169, indicated by 172, is a lock-in divided by its output impedance R ₀ because the impedance from point 170 to ground is 0. Equal to the output error voltage of amplifier 46. However, since no current flows from the connection point 170 to ground, the input current i _in flows through the capacitor 168,
And the output voltage V _{0 with} respect to ground is formed on line 174 as a function of time. Output voltage V as a function of time
The expression for ₀ is:

Here, C is the value of the capacitor 168.

Referring to FIG. 16, the response characteristic for the operational amplifier integrator 169 is shown. FIG. 16A shows a hypothetical error signal on line 318. The output voltage V ₀ of the integrator on line 174 is plotted in FIG. 16 (B).

As can be seen from FIG. 16 (B), for a 0 error signal, the output voltage curve has a slope of 0, and as the magnitude of the non-zero error signal is increased, it becomes V ₀ . The magnitude of the output voltage curve slope increases. That is, the sign of the slope depends on whether the error signal is positive or negative, and the slope of the slope at any instant in time depends on the magnitude of the error signal at any instant in time.

As the error signal increases from the origin to point 176, the integrator output signal V ₀ increases to point 176B. Once again, the 15th
Referring to the figure, MC149 manufactured by Motorola
Conventional balanced modulators, such as the 6L, and related circuits,
This input voltage V ₀ on line 174 is converted to a corresponding change in the envelope of the drive signal on line 322. That is, the amplitude of modulator 188 modulates the fixed amplitude signal on line 320 with the signal on line 174. This drive signal on line 322 is then sent to line 32
4 (FIG. 11), on line 324, it is combined with the bias modulation signal from generator 40 and provided to phase modulator 38.

As the amplitude of the drive signal on line 322 increases, the amplitude of the low frequency component in the optical output signal begins to increase. When it rises far enough, the time averaged value of the gated signal attempts to cancel the first harmonic component caused by the rotation. This is the point 176 and 177 in FIG. 16 (A).
Try to reduce the error signal shown in between. The decreasing error signal changes the slope of the slope of the integrator output voltage V ₀ in FIG. 16 (B), as shown between points 176B and 177B. At point 177 in FIG. 16A, the magnitude of the drive signal is just sufficient to cancel out all of the first harmonic components caused by the rotation of the optical output. This is points 177B and 178
Between B, it is reflected by the flat point non-zero portion of the integrator output voltage curve for V ₀ .

At time 178 in this hypothetical situation, the speed of rotation of loop 14 begins to increase in magnitude as the error signal changes sign and is shown between 178 and 180 in FIG. 16 (A). Changes to. by this,
A decrease in output voltage V ₀ occurs because current i _in changes direction and the voltage on capacitor 168 begins to change. This is point 1 in FIG. 16 (B).
Shown between 78B and 180B. The effect is to reduce the amplitude of the drive signal, which tends to cause the error signal to return to the zero direction as seen between points 180 and 182 in FIG. 16 (A).

At time 182 in the hypothetical situation, the rotation of loop 14 again causes more of the first harmonic component to become at points 182 and 1
84 and 84 to vary as generated by the Sagnac phase shift to flatten the error signal curve. This causes the integrator output voltage to ramp downward with a constant ramp to reduce the amplitude of the second or low frequency phase modulated signal between points 182B and 184B.

At time 184, the rotational speed of the loop changes again, but the error signal is still negative and non-zero. Non - by a zero error signal, the integrator output voltage V ₀ continues to decrease, it by and the error signal changes the amplitude of the drive signal, the zero direction as shown between points 184 and 186 move.

Once the error signal reaches zero, the integrator output voltage is
It holds steady any amplitude that cancels all or substantially all of the first harmonic component generated by the Sagnac. The situation at time 186 is line 140
The first drive signal amplitude generated by the backscattering of
3 shows a non-zero constant rotational speed in loop 14, adjusted to an appropriate level just to cause cancellation of harmonic components.

Those skilled in the art will appreciate that if the rotation continues to accelerate in one direction, the output voltage V ₀ will rise to the safe level above, causing a component failure in the amplitude modulator 188 for the circuit of FIG. 15, for example. Let's understand that.
In order to avoid such a situation, the voltage limiting device must be coupled to the integrator for limiting the excursions of the maximum positive and negative voltage V _0.

Referring to FIG. 17, there is shown a preferred embodiment for a portion of the error correction modulator circuit 130 to replace the integrator 190 in FIG. In this embodiment, the inverting input of differential amplifier 192 is coupled to the error signal on line 318 and its output is coupled to amplitude modulator 188 by line 174.

The manner in which the system shown in FIG. 17 works is better understood with reference to FIG. 18, which schematically shows a general rotation sensor, which is a 3-port network 1
The components of the sensor represented by 96 are coupled to differential amplifier 192. The optical part and most of the electronic components of the sensor are represented by a voltage division impedance network 196 having two inputs coupled to either end of the _two impedances Z ₁ and Z ₂ . The midpoint of this divider is coupled to the inverting input of differential amplifier 192.

When rotation is applied to the loop, the rotation signal (symbol) is applied to the second input of the 3-port network 196, which network 1
96 produces an error signal provided on line 318 which is coupled to the inverting input of differential amplifier 192. The difference between this input error signal and the reference signal on line 133, which is ground potential in this case, is filtered and inverted by the differential amplifier 192 and the amplified difference signal is output line 194. Given to. This output line is also coupled to the first input of network 196, which causes negative feedback through impedance Z ₁ which tends to cancel the voltage at point 198 caused by the rotation signal.

The signal on line 194 then seeks to minimize the voltage swing at point 198. Point 198 physically represents the output of lock-in amplifier 46 of FIG. Impedances Z ₁ and Z ₂ are virtual impedances that represent the overall transfer function and loop gain of the optical and electronic parts of the system.

System time response, phase margin, bandwidth and sensitivity are application-dependent design choices, and standard feedback system folds can be used to establish system parameters.

The effect of the feedback via impedance Z _{1 is} to constrain the error signal swing on the lock-in amplifier output line 318 to the small range indicated by box 200 in FIG. This range is a design choice and depends on the gain of the differential amplifier 192. Higher gains result in a smaller range of variation of the input signal, ie smaller boxes, but less stability.

Any structure responsive to the non-zero error signal to reduce the error signal to zero or substantially zero by increasing or decreasing the magnitude of the second phase modulated drive signal on line 322 is contemplated by the present invention. Will be enough for the purpose. For certain embodiments, it is desirable to maintain the level of the second phase modulation drive signal with a cancellation amplitude that reduces the error signal to zero or near zero. The exact circuitry used to accomplish this function is not a problem for this invention.

An alternative circuit that can be used for the error correction modulator is as shown in FIG. In this embodiment, the error signal on line 318 is coupled to the input of compare processor 201. The comparison processor has a reference voltage applied to its reference input 203, which in this case is ground potential. The compare processor compares the error signal on line 318 with the reference signal on line 203 and produces one of three outputs. If the error signal is positive and non-zero, then output line 205 is driven as a logic 1 level. If the error signal is negative and non-zero, line 207 is driven. Finally, if the error signal is equal to the reference signal, line 205 is driven.

Up-down counter 211 has an up input coupled to line 205 and starts counting from zero when line 205 is active. If the data on the bus 213 at any moment represents a binary representation of the count,
As the count progresses, a binary count is output on the bus 21.
The digital pattern on 3 is continuously changed.

Digital-to-analog converter 215 continuously or periodically samples the value of the binary count on bus 213 and converts the digital data to an analog output signal on line 174. This analog signal amplitude-modulates the second phase-modulated drive signal on line 320 and provides it on line 322, so that the conventional amplitude modulator 1
Used by 88.

The varying amplitude of the second phase modulation drive signal is on line 318.
This is reflected in the changing error signal above. That is, the error signal will tilt toward the reference signal voltage.

When the error signal reaches the reference voltage, comparator processor 201 drives line 209, which is coupled to the stop input of counter 211, thereby stopping counting. The D / A converter then changes the amplitude level of the second harmonic drive signal until the error signal changes again.
The level existing at that time is constantly maintained.

When the error signal goes negative and goes to non-zero, the process repeats, but the counter 211 begins counting down from zero or from the positive count present. If the count was zero when line 207 was driven, decoder 217 drives change sign line 219 which causes D / A converter 215 to sign the analog output voltage on line 174. Change. If the count is not zero when line 207 is activated, decoder 217 will not drive line 219 and D / A converter 215 will be at the same sign as when line 205 was driven.
Holds the analog signal above, but begins to lower its amplitude as the count decreases. This process is line 2
It continues until 09 is activated.

Since the transfer function is not linear in the same region, the linear element used to transfer the amplitude of the second harmonic drive signal to the magnitude of the Sagnac phase shift introduces error. A device may be used at the output for storing the transfer function or for solving the transfer function for the rotational speed or the Sagnac phase shift given the canceling amplitude of the second harmonic drive signal. That is, it is advantageous to convert the amplitude of the drive signal on line 322, which cancels the first harmonic component of the output due to the Sagnac phase shift, to the rotational speed or the Sagnac phase shift itself. This is the eleventh
This is the purpose of the output display circuit 208 shown.

FIG. 20 shows a preferred circuit for output display 208. The first harmonic of the drive signal, passed through bandpass filter 326 on line 322, is coupled to the input of lock-in amplifier 210. The lock-in amplifier is tuned to the drive signal, ie it has as its reference signal an unmodulated signal on line 320 from the signal generator 308 of FIG. Lock-in amplifier 210
The purpose of is to filter out all the noise on line 322 that scatters the desired waveform. This noise can result from noise due to power lines, electromagnetic interference, crosstalk with drive signals on line 324, and other miscellaneous sources.

The output signal on line 212 is proportional to the amplitude of the filtered drive signal at the output 212 of the lock-in amplifier and is coupled to an analog-to-digital (A / D) converter 214, which converts it to digital data. To be converted. This data is a memory 2 that stores digital data relating to the rotational speed corresponding to each amplitude of the drive signal determined by the transfer function of equation (15).
Used by the microprocessor or computer 216 to address the look-up table at 18.

The digital data at the output 217 of the A / D converter 214 is used by the microprocessor 216 to
The correct address of ROM 218 is accessed which stores digital data indicative of the rotational speed or corresponding Sagnac phase shift of the drive signal on line 332 for that particular amplitude. The program for the microprocessor 216 to perform this address function will be apparent to those skilled in the art, and any program for performing this function will suffice. The digital data output from the ROM can be converted to analog form by digital-to-analog converter 220, or it can be used in digital form.

In another embodiment, the microprocessor 216 uses the variable Δ.
It can be programmed to solve the transfer function of equation (14) by using the data from the A / D converter 214 as φ _m . In these examples, ROM 218 would contain programs to perform the calculations required by equation (14). The exact program used to make this calculation does not matter, and the program for making this calculation will be known to those skilled in the art. Any program capable of performing this calculation would be suitable for the purposes of this invention.

In another embodiment, R.R.
M. S. A voltmeter may be used, but such a structure seems to lead to errors. Because some noise on line 332 may be averaged and misinterpreted as the amplitude of the error in the drive signal. R. M. S. The voltmeter has its input at the midpoint of the voltage divider shown in FIG. The drive signal is applied to the connection point 221 of the voltage divider formed by the resistors R1 and R2. The resistors R1 and R2 reflect the slope of the transfer function in the linear region so that for a given amplitude of the drive signal at the connection point 221, a signal having an amplitude proportional to the rotational speed is generated at the connection point 222. Chosen to let. This signal is the R. M.
S. Coupled to the input of the voltmeter and read as a Sagnac phase shift or rotational speed.

Furthermore, the oscilloscope also detects the amplitude of the drive signal, as shown in FIG. M. S.
It can be used instead of a voltmeter. A linear scaling network of resistors R3 and R4 is also used to scale its input to the oscilloscope. The embodiments of Figures 21 and 22 are most accurate in the linear region of the transfer function.

Any other device capable of measuring lower modulation frequency waveforms can also be used for the output display circuit 208. For example, an analog curve matching device
It can be used to compensate the transfer function curve and to provide an output proportional to the rotational speed. further,
The ROM lookup table and microprocessor of FIG. 20 is distributed in the approximately linear region of the transfer function curve so that the simplified FIG. 20 embodiment can also be used in the approximately linear region for suitable results. Can be done.

Another preferred embodiment of a method and apparatus for sensing rotation with a generally linear scale over an extended dynamic range may be described with reference to FIGS. 23-29.

In a device such as that shown in FIG. 1, by introducing a time-varying signal through an asymmetrically positioned phase modulator 38, the signals will be in opposite directions when measured at the output of detector 30. A phase difference occurs between propagating waves. Such induced differential phase shift Δφ
(T) is defined as follows.

Δφ (t) = φ (t) −φ (t−τ) (16) where φ (t) is the phase shift generated by the phase modulator at time t, and τ is the phase modulator. The time difference between the interfering waves passing through 38.

Referring to FIG. 23, the DC phase difference between the waves propagating in opposite directions at a given time Δφ (t) has a linear phase slope, such as that shown at 400, through the phase modulator 38. It is achieved by giving a wave propagating in the opposite direction. In particular, the linear phase slope 400 shows the effect of the slope signal input through the modulator 38 on waves propagating counterclockwise through the sense loop. The effect of the same input signal on the signal propagating counterclockwise is shown by line 402. The amount of difference τ between slopes 400 and 402 depends on the asymmetrical placement of phase modulator 38 in the sense loop.

The phase difference signal Δφ (t) between the counterclockwise and clockwise propagating waves is shown at 404. It is of particular interest that this phase difference is a DC value whose magnitude can be changed by adjusting the slope of the slope signal. It thus becomes apparent that the tilt signal is provided via the phase modulator 38 to produce a DC phase difference magnitude that can be adjusted to effectively eliminate the rotation induced Sagnac phase shift. .

One means of generating such a ramp function would be to use frequency shifters placed asymmetrically in the sense loop. In this case, Δφ would be defined as:

Δφ = 2πτΔf (17) Here, Δf is the frequency shift amount. The use of a frequency shifter would have the additional advantage of allowing the frequency output to be used as a measure of rotational speed. However, no fiber type frequency shifter suitable for gyroscope application has been reported.

Commonly used fiber optic phase modulators, such as the modulator 38 that modulates the fiber length, cannot provide a continuous phase tilt to create a DC differential phase shift between waves propagating in opposite directions. Therefore, in order to use the phase modulator in this application, it is necessary to simulate the phase tilt.

FIG. 24 illustrates one waveform that may be used to simulate phase tilt. In particular, FIG. 24 (A)
Line 406 shows the case where a sawtooth wave is applied to the signal propagating counterclockwise in the sensing loop. Figure 6 shows the effect of the same sawtooth waveform on the clockwise propagating signal of the sensing loop from an asymmetrically positioned phase modulator. The line 410 in FIG.
The phase difference signal (Δφ (t)) generated by the phase difference between the interfering waves shown in FIG.

As can be seen from the waveform represented by line 410, the phase difference may not always be constant due to the reset process and interrelationship of the two optical paths. However, during these periods, shown at 412 when line 410 defines the DC value, the DC Sagnac phase shift can be eliminated by adjusting the amplitude or frequency of the phase modulation. Therefore, by applying a substantially DC phase bias to the phase difference between the light waves propagating in opposite directions, the Sagnac phase shift received by the phase difference between the light waves propagating in opposite directions is made substantially zero. It is further noted that during the period not included in the segment shown at 412, a zero sagnac phase shift can be simulated by turning off the rotation signal received from the detector 30 of FIG. See.

As a result, the rotation-induced Sagnac phase shift is due to the phase modulation induced phase difference during a portion of time and at or after detector 30 for the rest of the time at the signal of light source 10. It can be effectively eliminated by turning off. The slope of the slope, which determines the differential phase shift, can be controlled by adjusting the amplitude of the modulation signal. Of course, it will be appreciated that other waveforms having a ramp-type morphology can also be used to produce a similar effect. For example, with the understanding that the generation of a DC phase modulated output signal would require that the signal be turned off for a longer period than in the case of a sawtooth wave due to the shorter ramp length of the triangular wave shape. , Triangular wave phase modulation can be used.

One of the most commonly used fiber optic phase modulators
The first is a piezoelectric cylinder with several turns of fiber wrapped around it, as explained previously. Unfortunately, the frequency response of this device is not uniform over a wide frequency range. As a result, it is almost impossible to achieve a sawtooth waveform phase modulation of the type shown in FIG. 24 unless the amplitude and phase of each Fourier component of the waveform is controlled.

One way to overcome the non-uniformity problem described above is to generate a sawtooth or triangular waveform in an approximate manner by combining sinusoidal phase modulation in all fiber optic rotation sensors. For example, a sawtooth waveform may be used for phase difference modulation of one frequency and second phase of that frequency.
Simulated by combining with the harmonics, the amplitudes of the second harmonics and the phase relationships of those waves are adjusted appropriately. Similarly, a triangular waveform can be generated by combining the phase difference modulation of a frequency with the third harmonic of the frequency, which was properly adjusted for the amplitude and phase relationship. .

FIG. 25 shows one preferred set of waveforms used in all fiber optic gyroscopes to simulate a sawtooth waveform. In particular, the first phase modulated signal for simulating a sawtooth wave comprises the sine wave 450 of FIG. 25 (A) defining φ ₁ (t). Line 45
0 indicates the effect of a sinusoidal phase-modulated signal on a counterclockwise propagating wave in the sense loop, and line 4
52 shows the effect of this same sinusoidal modulation signal on a wave propagating clockwise.

In FIG. 25B, line 454 illustrates the effect on the counterclockwise propagating wave of the second phase modulated signal at the second harmonic frequency of the sinusoidal modulated signal 450.
This second harmonic phase modulated signal is shown as φ ₂ (t). Line 456 in FIG. 25 (B) illustrates the effect of the second harmonic phase modulated signal on the clockwise propagating wave.

FIG. 25 (C) shows a waveform composed of the sum of the modulated signals of FIG. 25 (A) and FIG. 25 (B). Especially 4
The sawtooth waveform shown at 58 has waveforms 450 and 4
The sum of 54 and the response of the counterclockwise propagating wave to this modulated signal is shown. Similarly, a sawtooth waveform, shown at 460, describes the sum of waveforms 452 and 456 and shows the effect of this waveform on the clockwise propagating wave at the rotation sensor.

FIG. 25 (D) shows phase difference modulation with respect to time. Four
This signal, shown at 62, therefore has a waveform 458.
(Φ (t)) and the waveform 460 (φ (t−τ)), where τ is the time difference between the interfering waves passing through the phase modulator. The waveform of FIG. 25 (D) can be explained as follows.

Δφ = cos ω _mt + 0.3 cos 2ω _mt (18) As described with reference to FIG. 24, the waveform 462 (Δφ
It should be noted that (t)) includes the portion indicated at 464, which is generally linear. As explained above, by gating the phase difference signal, the generally linear or DC portion of the phase difference modulation 462 can be used to effectively eliminate rotation-induced Sagnac phase shifts. As in the sawtooth waveform of FIG. 24,
The amplitude of the DC portion 464 can be controlled by adjusting the amplitude or frequency of the phase modulation. Thus, the DC-like portion of the phase difference modulation 464 can be used to eliminate the Sagnac phase shift Δφ _R , and the signal turn off during the period not included in the 464 portion is the rest of the time. Can be used to simulate a zero Δφ _R.

FIG. 26 shows the lower frequency (f _m ) second modulation signal in the rotation sensor of FIG. 11 if FIG. 25 (C).
5 schematically illustrates the combined effects of phase modulation that may be present if the sawtooth waveform 458 of ## EQU1 ## was introduced. FIG. 26 also shows the output signal that would be detected as a result of phase modulation under these circumstances.

In particular, the DC value for the phase shift resulting from the Sagnac phase shift at a fixed rotational speed is shown at 352. The phase modulation signal generated by the sawtooth second modulation signal is shown at 354. In addition, the phase modulation produced by the bias modulation signal (f _b ) is shown at 350. As with the sinusoidal modulation waveform used in the embodiment of FIG. 11, the sawtooth modulation signal has a bias modulation frequency f _b.
It is noted that it must be at a much lower frequency than.

It can be seen in FIG. 26 that the phase modulated signal described above oscillates for a DC phase shift 352 that is Δφ _R. At a lower frequency, the amplitude of the second phase modulation 354 is such that the generally flat or DC portion of its line 354 has a vertical axis 35.
It can be seen that it has been adjusted to be positioned on. Therefore, by gating either the light source 10 or the detector 30 output of the apparatus of FIG.
A lower frequency modulated signal 354 as shown at 464.
It is possible to output only those parts of the resulting output signal that are generated during the DC part of the. During this gated period 464, the resulting signal is the vertical axis 35.
Vibrate about 5. For the rest of the period, the output signal is equal to zero, thus simulating a situation where the Sagnac phase shift is zeroed.

The output resulting from gating the rotation sensor as described above and for the period shown at 464 in FIG. 26 produces an output signal having a waveform approximated by the waveform shown at 466 in FIG.

The output signal 466 does not contain the first harmonic, indicating that the Sagnac phase shift Δφ _R was substantially zero during the gated period and is not monitored during the off period. I am particularly interested in the fact that
As described above, by monitoring the amplitude of the second phase modulation signal, it is possible to determine the amount of rotation that the gyroscope receives even under a dynamic condition in which high rotation is expanded. A preferred circuit for detecting the amplitude of this signal and determining its rotational speed has been previously described with respect to the sensor shown in FIG.

FIG. 27 shows a preferred embodiment of a rotation sensor used to monitor rotation using a simulated ramp modulation signal. Many of the components of the apparatus shown in FIG. 27 are first in configuration and operation.
Note that it corresponds to the elements contained in the device of FIG. Corresponding elements are therefore indicated with corresponding reference numerals.

Based on its construction, it becomes apparent that the rotation sensor shown in FIG. 27 functions in a manner that is substantially identical to the sensor of FIG. However, the sensor shown in FIG. 27 generally replaces the sinusoidal second modulation signal with a low frequency modulation signal shaped like a sawtooth wave. The signal generator 308 transmits a sinusoidal waveform onto line 500 to generate a sawtooth modulated signal. This sinusoidal waveform is substantially the same as the waveform transmitted on line 320 from generator 308 of FIG. Further, the signal generator 308
The sinusoidal waveform from is also transmitted on line 502 to frequency multiplier 504, which receives the sinusoidal modulated signal of frequency f _m and doubles its frequency for transmission to amplitude adjuster 506. And generate a second harmonic of frequency 2f _m .

The device 506 may include conventional means for adjusting the amplitude of the signal, such as a potentiometer. From the amplitude adjuster 506, the modulation is transmitted to a phase shift circuit 508, where its phase generally corresponds to the relationship between the waveforms shown in FIGS. 25 (A) and 25 (B). It is shifted with respect to the first harmonic phase modulation signal from generator 308. The amplitude adjustment circuit 506 and the phase shift circuit 508, as long as the sinusoidal modulation waveform generated by the signal generator 308 is maintained at a constant frequency f _m, it is manually set by timing the one.

The second harmonic waveform from the phase shift circuit 508 is line 5
00 on line 510. Therefore, the first harmonic signal on line 500 and line 510
The second harmonic signal above is combined to produce a phase modulation waveform having a generally sawtooth shape, such as that illustrated in FIG. 25 (C). The signals from lines 500 and 510 are combined and transmitted via line 320 to error correction modulator 130, where the combined signals are first
Processing is performed in the manner described with reference to the rotation sensor illustrated in FIG.

As explained above, the rotation sensor of FIG. 27 gates the output signal so as to detect only the portion of the output resulting from the phase modulation produced by the sloping portion of the sawtooth wave, thereby providing the Sagnac effect. It works to reduce the DC effect of. As a result, the gating signal on line 306 from signal generator 308 must be adjusted so that gate 304 is only turned on during the ramped portion of the sawtooth wave. It has been found that the gating signal from signal generator 308 must be set to gate about 30% of each period of the modulating signal on line 320. The portion of the waveform of gated line 320 is
It is identified by simply extrapolating upwards the gated period identified at 464 in FIG.

By referring to FIG. 28, it can be seen that the transfer function or magnification resulting from the use of the rotation sensor of FIG. 27 is substantially linear. This result is obtained due to the fact that the Sagnac phase shift (Δφ _R ) is zeroed out by the phase difference modulation (Δφ _m ) which defines a substantially DC signal. Thus, as shown in the graph of FIG. 28, the increase in the magnitude of the phase difference modulation generated by the Sagnac effect corresponds to the magnitude of the phase difference modulation generated by the ramp portion of the sawtooth modulated signal. The increase can effectively eliminate it.

As with the rotation sensor of FIG. 11, the bandpass filter 326 provides a signal at frequency f _m from line 330,
This output device 208 is passed through to the output display device 208.
Is used to determine the rotational speed by identifying the amplitude of the phase modulation signal required to cancel the Sagnac phase shift.

The linearity of the magnification shown in FIG. 28 practically eliminates the wavelength dependence of the light source of the gyroscope magnification. this is,
The amplitude of the phase difference modulation is possible because it has the same wavelength dependence for a given signal that the Sagnac phase shift has for rotational speed (1 / λ). Considering the fact that the wavelength of the light source is difficult to control, this phase modulation approach can improve the stability of the scale factor. The stability of the system is further improved if the feedback modulation frequencies f _m and 2f _m do not match the resonant frequency of the phase modulator. Moreover, if the harmonic frequency of f _m does not match the bias modulation frequency f _b , then additional offset or noise in the rotation signal is also removed.

In the rotation sensor of FIG. 27, the gated time intervals and relative amplitudes of the two frequency components are on the order of 10 ⁻⁵ to 20 radians maximum of the Sagnac phase shift, which assumes the linear response of the phase modulator for a given signal. Can be adjusted to give linearity of the magnification of (eg, Δφ (t) α (cos (ω _mt ) +0.4 cos (2ω
_mt )).

In both the rotation sensor of FIG. 11 and the rotation sensor of FIG. 27, gating induces a loss of optical power and potential loss of rotation information during the time the sensor is gated off. The device of FIG. 11 (A) contains a loss of half the optical power, since the device is typically gated off for approximately half of its time. In the device of FIG. 27, at about 30% of the gate on the waveform, the loss of optical power occurs for about 70% of the time. This loss of information can cause errors in the measured rotation angle θ when abrupt changes in the rotation angle occur within the gate-out time interval. As an example, a value used to represent the maximum expected acceleration in many applications,
Consider the case of a full cycle of square wave angular acceleration using an acceleration of | t ² θ / dt ² | = 1,000 ° / sec ² . For a typical gating frequency f _{m of} 15 kHz and with gating occurring for half of that time, the above-mentioned amount of acceleration and subsequent acceleration within the first half of the time interval being gated out. The same magnitude of deceleration within the other half of the interval causes an error of θ of approximately 2.8 × 10 ⁻⁷ °. Thus, it becomes clear that the phase sensing device described herein is very reliable and that the effect of the gate configuration makes the error tendency in measuring the rotational speed very small.

FIG. 29 shows another embodiment of the rotation sensor using the sawtooth waveform. In this example, the signal generator 308 produces a modulated signal of frequency f _m , which comprises a train of square wave pulses. These square-wave pulses include 2f _{m and} have a frequency f _m
Including harmonics of. These square wave pulses are transmitted to the error correction modulator 130 via line 320 and processed in the manner previously described for the sensor illustrated in FIGS. 11 and 27.

The square wave signal generated by modulator 130 is on line 5
It is transmitted to the low pass filter 532 via 30. The filter 532 removes all but the first and second harmonics of the modulation transmitted from the error correction modulator 130. The filtered signal is then transmitted via line 534 to phase adjustment circuit 536. Phase adjustment circuit 53
One particular embodiment of No. 6 is a tunable bandpass filter used to modify the phase of the second harmonic with respect to the first harmonic to produce the desired sawtooth waveform for phase modulation. Including.

The sawtooth waveform modulation signal from the phase adjustment circuit 536 is line 5
38, where it is generated by the signal generator 40 on line 324, the sinusoidal modulation frequency f _b.
Combined with. The generated signal is given to the phase modulator 38 as a modulation signal. Second in all other respects
The sensor of Figure 9 functions in the same manner as the sensor of Figure 27.

One particular embodiment of the rotation sensor shown in FIG. 29 is constructed and evaluated as follows. The fiber length and radius of the sensing coil are approximately 580 meters and 7 centimeters, respectively. The wavelength of the light source used is about 8
It is 30 nanometers. The phase modulator 38 comprises a piezoelectric hollow cylinder with several turns of fiber wrapped around it. The first resonance frequency of the piezoelectric cylinder is about 20 kHz. The bias modulation frequency f _b generated by the signal generator 40 is 172 kHz, resulting in a phase difference modulation (Δφ _b ) amplitude equal to about 1.8 radians.

The sawtooth frequency modulation may be generated as follows. A square wave pulse train is generated by a signal generator 308, which includes a pulse generator with a repetition frequency f _m of 15 kHz. The frequency spectrum of this signal contains harmonics of the fundamental frequency f _m . Low pass electrical filter 532
Suppresses all frequency components, leaving only the first and second harmonics of f _m . The relative amplitudes of these frequency components (15 kHz and 30 kHz) are adjusted by varying the width of the square wave pulse from the pulse generator. The variable bandpass filter includes a phase adjustment circuit 536 used to adjust the relative phase of the two frequency components. This signal, combined with the bias modulation signal from the generator 40, is provided to the phase modulator 38.

The electrical signal from the silicon photodetector 30 is on line 306.
The synchronization signal from the pulse generator 308 transmitted above is gated on the electrical switch or gate 304. The gating phase may be adjusted by adjusting the pulse delay of the trigger signal. About 30% of the signal from detector 30 is allowed to pass through gate 304 to obtain a linearized scale factor. The signal from gate 304 is transmitted across line 310 and through bandpass filter 312, which allows it to pass only above bias modulation frequency f _b . This signal is then measured in the lock-in amplifier 46 against the reference signal at f _b frequency from the signal generator 40. The comparison of the signal from the filter 312 to the reference signal produces an error signal from the lock in the amplifier 46, which is communicated to the error correction modulator 130 described in the first part of the specification. The actual scale factor resulting from the operation of the circuit of FIG. 29 corresponds to the scale factor indicated by 520 in FIG.

Although the rotation sensor described here illustrates the use of a single phase modulator, it should be understood that another phase modulator can be used for the bias phase modulator and the lower frequency second phase modulator. Will be understood by those skilled in the art. It will also be appreciated that other waveforms can be used with the gating circuits described herein, with acceptable results. Such alternative embodiments are also considered to be within the scope of the invention described and claimed herein.

In summary, not only does the invention described herein include a significant improvement over the prior art in extending the dynamic range for rotation sensing over a very wide range of rotation speeds, the invention also includes (1) While using only one phase modulator selectively, it provides a means to obtain extended dynamic rotation detection, and (2) greatly improves the stability by substantially suppressing the source wavelength dependence of magnification. Such rotation is detected by the sex, and (3)
By linearizing the magnification or transfer function and thereby significantly simplifying the signal processing required in the sensing device, other long-standing problems in industry are overcome.

The present invention may be embodied in other specific forms without departing from its spirit or essential characteristics. The described embodiments are in all respects only schematic and should not be considered in a limiting sense. The scope of the invention is, therefore, indicated by the appended claims rather than by the foregoing description. All changes that come within the meaning and range of equivalency of the claims are to be embraced within their scope.

[Brief description of drawings]

FIG. 1 is a schematic of a basic rotation sensor, showing fiber optic components positioned along a continuous, uninterrupted strand of fiber optic material, and further associated with a detection system. A signal generator, a photodetector, a lock-in amplifier and a display device are shown. 2 is a cross-sectional view of one embodiment of a fiber optic directional coupler for use in the rotation sensor of FIG. FIG. 3 is a sectional view of an embodiment of a fiber optic polarizer for use in the rotation sensor of FIG. FIG. 4 is a perspective view of an embodiment of a fiber optic polarization controller for use in the rotation sensor of FIG. FIG. 5 is a schematic diagram of the rotation sensor of FIG.
The polarization controller and the phase modulator have been removed therefrom. FIG. 6 is a graph of the optical output signal intensity measured by a photodetector as a function of rotationally induced Sagnac phase difference, showing the birefringence induced phase difference and the birefringence induced amplitude variation. Illustrates the effect of. FIG. 7 is a graph of phase difference as a function of time showing the phase difference between counter-propagating waves and the phase modulation of each counter-propagating wave. FIG. 8 is a schematic diagram showing the effect of phase modulation on the intensity of the optical output signal as measured by the detector when the loop is stationary. FIG. 9 is a schematic diagram showing the effect of phase modulation on the intensity of the optical output signal measured by the detector when the loop is rotating. 10 is a graph of amplifier output signal as a function of rotationally induced Sagnac phase difference, showing the operating range for the rotation sensor of FIG. FIG. 11 is a diagram of a preferred embodiment of a gated closed loop rotation sensor with extended dynamic range. FIG. 12 is a diagram of the overall frequency shift resulting from the lower frequency phase modulation and the bias phase modulation for a constant bias resulting from the Sagnac effect. FIG. 13 is a diagram of the overall phase shift for lower frequency phase modulation for a constant bias resulting from the Sagnac effect and the optical output signal resulting from gating. FIG. 14 is a graph of the magnification of the rotation sensor shown in FIG. FIG. 15 is a circuit diagram of the error correction modulator. FIG. 16 is a diagram of the response of the modulator of FIG. 15 to a sample error signal. FIG. 17 is a diagram of a preferred error correction modulator. FIG. 18 is a schematic diagram of the whole sensor using the error correction modulator of FIG. FIG. 19 is a diagram of another error correction modulator that may be used in the embodiment of FIG. FIG. 20 is a diagram of a preferred embodiment of an output circuit for a rotation sensor for converting the amplitude of a lower frequency drive signal into a rotation speed. FIG. 21 is a diagram of an output display circuit that can be used in the linear region of magnification. FIG. 22 is a diagram of another output display circuit that can be used in the linear region of magnification. FIG. 23 is a graph showing the relative phase difference between interfering waves modulated by the ramp waveform. FIG. 24 shows the relative phase between the interfering waves modulated by the sawtooth waveform and the phase difference between these interfering waves. FIG. 25 is a schematic illustration of one method of forming a sawtooth wave, and further showing the relative phase between the interfering waves modulated by the sawtooth waveform and showing between these interfering waves. Indicates the phase difference. FIG. 26 is a diagram of the overall phase shift resulting from bias phase modulation and the lower frequency sawtooth phase modulation for constant bias resulting from the Sagnac effect and the output signal resulting from gating. FIG. 27 is a diagram of a preferred embodiment of a gated closed loop rotation sensor with expanded dynamic range and substantially linearized scaling. FIG. 28 is a diagram of the magnification of the sensor shown in FIG. FIG. 29 is a diagram of another preferred embodiment of a gated, closed loop rotation sensor with extended dynamic range and a substantially linear transfer function. In the figure, 10 is a light source, 12 is an optical fiber, 14 is a detection loop, 24 is a polarization controller, 26 is a directional coupler, 30 is a photodetector, 32 is a polarizer, 36 is a polarization controller, and 38 is a phase. A modulator, 46 is an amplifier, and 47 is a display panel.

 ─────────────────────────────────────────────────── --Continued from the front page (56) References JP-A-55-93010 (JP, A) JP-A-58-501597 (JP, A)

Claims (14)

[Claims] 1. A device for detecting and measuring a physical parameter using an optical loop, comprising a light source for generating a light wave propagating in the optical loop in the opposite direction, and a light wave propagating in the opposite direction. A detector responsive to the phase difference of the counter-propagating light waves is shifted according to the physical parameter, and the light waves are phase modulated at a first frequency and combined to form an output signal, at least A gate circuit for blanking the components of the output signal at a selected time to produce a gated signal, and biasing the phase difference of the light waves propagating in the opposite direction at a second substantially constant frequency. A differential phase modulator for generating a periodic bias at the second frequency of the differential phase modulator in response to the output of the detector, the physical bias being generated by the physical parameter. Sensing a physical parameter, comprising a control element for adjusting the component in the gated signal to zero, and a circuit for measuring the amount of the periodic bias to determine the magnitude of the physical parameter. Device for measuring. 2. An apparatus for detecting and measuring physical parameters according to claim 1, wherein said gating circuit blanks said second frequency component. 3. An apparatus for detecting and measuring a physical parameter according to claim 1, wherein the second frequency is lower than the first frequency. 4. A first signal generator for generating a modulated signal of the second frequency, the first signal generator driving the phase difference modulator via the control element. An apparatus for detecting and measuring a physical parameter according to any one of claims 1 to 3. 5. The gate circuit comprises: a gate element connected to the detector and for erasing a selected portion of the light wave propagating in the opposite direction in response to the first signal generator; Connecting a gate element to the control element, comparing the output of the gate element with a reference signal, and driving the control element to change the amplitude of the modulation of the first frequency to the gated signal according to the physical parameter. A feedback circuit for adjusting so as to cancel the generated components,
An apparatus for detecting and measuring physical parameters according to claim 4. 6. The apparatus for detecting and measuring physical parameters of claim 1 wherein said phase difference modulator imparts a substantially DC phase bias to said counter-propagating light waves. 7. The method for detecting and measuring a physical parameter according to claim 6, further comprising a bias circuit for applying a phase ramp modulation of a second frequency to the light wave propagating in the opposite direction. apparatus. 8. The bias circuit includes a first signal generator for generating a first modulated signal of the second frequency, and a resultant modulated signal in response to the first modulated signal. An electronic circuit for modifying the first modulated signal so that at least a portion of the resulting modulated signal defines a ramp waveform, and further the bias circuit electrically conducts the first signal generation. An error correction modulator for adjusting the amplitude of the resultant modulated signal in response to the physical parameter, the phase difference modulator providing the resultant modulated signal to the light wave propagating in the opposite direction. , Whereby substantially the DC component of the phase difference modulation produced by the phase ramp modulation during the ramp waveform portion of the resultant modulation signal causes the phase shift produced by the physical parameter at the phase difference to be zero. Apparatus for measuring and detecting a physical parameter in the range 7 claim of claims. 9. A gate element electrically connected to said detector for erasing a selected portion of said phase difference signal in response to said first signal generator; And a non-erased portion of the phase difference signal is detected and compared with a reference signal, and a feedback error signal is generated according to the comparison result, the error correction modulator outputs the amplitude of the resultant modulation signal to the feedback signal. An apparatus for detecting and measuring a physical parameter according to claim 8, further comprising a feedback circuit for controlling the amplitude of the error signal to be adjusted so as to be small. 10. The electronic circuit includes a frequency multiplier electrically coupled to the first signal generator to generate a second modulated signal that is a harmonic of the first modulated signal, An amplitude adjusting circuit for adjusting the amplitude of the harmonic so as to correspond to the amplitude of the first modulation signal, and a part of the sum of the waveforms of the first and second modulation signals defines a ramp waveform. Phase shift means for shifting the phase of the harmonic with respect to the first modulation signal, and the first and second modulations provided between the first signal generator and the phase shift means. Electrical connections for combining signals to form the resulting modulated signal,
An apparatus for detecting and measuring a physical parameter according to claim 8. 11. A method for determining a physical parameter utilizing an optical loop for propagating counter-propagating light waves phase modulated at a first frequency and combined to form an output signal therein. A method of blanking at least a component of the output signal at a selected time to produce a gated signal, the phase of a light wave propagating in the opposite direction to a second substantially fixed frequency. Biasing the physical parameter to substantially zero the component in the gated signal caused by the physical parameter; and measuring the amount of the bias to determine the physical parameter. How to decide. 12. Biasing the phase of said counter-propagating light wave in phase modulating said counter-propagating light wave at a second substantially fixed frequency, and responsive to said output signal. Generating a feedback signal, the feedback signal providing a measure of the amount of phase shift caused by the physical parameter in the phase difference of the oppositely propagating light waves that were not zeroed by the phase modulation, Adjusting the amplitude of the phase modulation of the second frequency in response to the feedback signal. 13. A method for determining a physical parameter according to claim 11, wherein the step of measuring the amount of bias comprises the step of monitoring the amplitude of the phase modulation of the second frequency. 14. The step of biasing comprises the step of:
Of the gated signal by applying a phase ramp modulation to the counterpropagating light wave for at least a portion of a period corresponding to the frequency of 12. The method of claim 11, wherein the component caused by the physical parameter is substantially zero.