JPH06169568A - Power supply device for controlling plurality of outputs and switching regulator - Google Patents

Abstract

PURPOSE:To provide a composite output controller simple in constitution and efficient and a switching regulator by which maximum output voltage not dependent on power voltage can be gotten. CONSTITUTION:FET.Tr1 on the primary side of a transformer is PWM- controlled by the output of a PWM control circuit 1. A synchronizing signal is taken out of a circuit 1 by a synchronous detection circuit 3, and the data of a D flip flop is modified, according to this synchronizing signal, and for the transistor Tr2 connected to the rectified output end of a secondary winding N3, the changeover of conductivity and nonconductivity is performed. This way, a composite output power supply device simple in constitution and efficient can be gotten.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a multi-output control power supply device and a switching regulator suitable for a power supply of a copying machine, a printer or the like.

[0002]

2. Description of the Related Art In a multi-output (hereinafter referred to as multi-output) control power supply device as described above, this output is fed back to stabilize a specific one output, and a PWM control circuit is used to control the switching element on the primary side. The conduction period is determined, and the other outputs are output to the secondary winding output by configuring or controlling some control circuit or uncontrolled. For example, if the other output is a small power output, the output can be stabilized easily by series control of a 3-terminal regulator, etc. However, if it is a large power output, the series control causes a large loss and the desired voltage value is usually obtained by the chopper control. Is getting A mag-amplifier is also conceivable as another output control device.

Further, the conventional voltage resonance type switching regulator is widely used as a device for generating a high voltage such as a charger of a copying machine or a horizontal deflection voltage of a CRT. Most of these types of devices are configured to intermittently apply power to the primary winding of the converter transformer by switching means such as transistors and FETs to generate a voltage according to the winding ratio on the secondary side. .

[0004]

[Problems to be Solved by the Invention]

(A) However, in the multiple output control power supply device, when the DC is once converted and the chopper type DC / DC converter is formed in the subsequent stage when the chopper control is performed, the number of parts increases and the efficiency also decreases. Also, when the secondary winding is directly chopped, the switching loss is large.

(B) In the mag-amplifier, since the saturable reactor is magnetized to the saturation region, the core loss of the core material is large.
Moreover, in general, the saturation magnetic flux density of a magnetic material has a negative temperature characteristic, and when there is no load at high temperature (in the corner case of core temperature increase), iron loss increases → temperature increases → saturation magnetic flux density decreases →
To avoid the case where the voltage cannot be controlled, a very large core is required. Furthermore, if the switching frequency is increased to reduce the size, the dead angle due to the uncontrollable magnetic flux of the saturable reactor and the inductance of the winding (the phenomenon that the output voltage drops because it blocks the first fixed time of the pulse) The effect of is relatively greater than at low frequencies. To avoid this, the number of turns of the saturable reactor must be reduced,
In that case, a slightly larger core is required to obtain the same saturation magnetic flux, and the merit of increasing the frequency decreases. The accuracy of general magnetic materials (eg saturation magnetic flux density) is ± 2 at most
It is 0%, and a large core must be used considering the worst condition.

(C) Further, in the above-mentioned voltage resonance type switching regulator, the PWM control circuit forms a triangular wave based on the synchronization signal from the synchronization detection circuit, and compares it with the output of the error amplifier by the comparator to perform PWM.
Forming a signal.

Therefore, the maximum duty of the PWM signal is constant irrespective of the power supply voltage. Therefore, when the power supply voltage increases, the maximum value of the output voltage on the secondary winding side increases, and when the power supply voltage decreases, the maximum output voltage increases. The maximum voltage becomes smaller. Therefore, in order to secure the output voltage, the maximum duty of the PWM signal is determined in accordance with the minimum power supply voltage. Therefore, when the power supply voltage is maximum, the maximum value of the output voltage becomes too large and the output is too high. , The power supply, the load, the switching element of the device, etc. may be damaged. This is particularly problematic because switching regulators are generally designed to handle power supplies in a wide voltage range.

The present invention has been made under the circumstances as described in (a) and (b) above, and it is an object of the present invention to provide a multiple output control power supply device and a switching regulator which are simple in structure and have high efficiency. It is intended.

The present invention has been made to solve the above-mentioned problem (c), and it is an object of the present invention to provide a switching regulator and a multiple output control power supply device which can obtain a maximum output voltage independent of the power supply voltage. It is one purpose.

[0010]

A multi-output control power supply device is constructed as in the following (1) to (8) and (13) to (15), and a switching regulator is installed in the following (9) to (12).
Configure as follows.

(1) A transformer having a primary winding, a first secondary winding and a second secondary winding, a first switching element for switching the primary winding, and the first switching element. Feedback control means for controlling the switching operation of the switching element in accordance with the output of the first secondary winding, a second switching element for switching the output of the second secondary winding, and the second And a synchronizing means for synchronizing the driving of the switching element with the driving of the first switching element.

(2) The multiple output control power supply device according to (1), wherein the synchronizing means causes the second switching element to transition during the OFF period of the first switching element.

(3) Primary winding, first secondary winding and second
Having a secondary winding, a first switching element for switching the primary winding, and a switching operation of the first switching element controlled according to the output of the first secondary winding. Feedback control means for
A second switching element that is provided on the low voltage side of the output of the second secondary winding and switches the output;
And a drive means for switching the switching element at a frequency lower than that of the first switching element.

(4) The multiple output control power supply device according to (3), wherein the driving means makes a transition of the second switching element during an off period of the first switching element.

(5) A converter transformer having a primary winding, a first secondary winding, and a second secondary winding, and a first switching means for connecting and disconnecting the primary winding and its power source. A PWM control of the rectifying / smoothing means for rectifying / smoothing the output of the first secondary winding and supplying the rectified / smoothed output to the first output terminal, and the first switching means in accordance with the voltage of the first output terminal. P
WM control means, rectification means for rectifying the output of the second secondary winding, second switching means for intermittently supplying the output of the rectification means to a second output end, and the PWM control means Synchronization detection means for extracting a signal synchronized with the pulse from the output of the comparator, comparison means for comparing the voltage at the second output end with a reference value, and the output of the comparison means is updated according to the output of the synchronization detection means. And a holding means for holding the same and supplying it as a control signal to the second switching means.

(6) The multiple output control power supply device according to (5), further comprising delay means for delaying a control signal supplied from the holding means to the second switching means by a required time.

(7) The circuit of the PWM control means, the synchronization detecting means, and the holding means is formed on the same chip together with a digital circuit such as CPU, ROM, RAM and an analog circuit such as a DA converter. Multiple output control power supply.

(8) A converter transformer having a primary winding, a first secondary winding, and a second secondary winding, and a first switching means for connecting and disconnecting the primary winding and its power source. A PWM control of the rectifying / smoothing means for rectifying / smoothing the output of the first secondary winding and supplying the rectified / smoothed output to the first output terminal, and the first switching means in accordance with the voltage of the first output terminal. P
WM control means, rectifying means for rectifying the output of the second secondary winding and supplying it to the second output end, and second switching for connecting and disconnecting the low-voltage side of the output of the second secondary winding. Means and
A synchronization detecting means for extracting a signal synchronized with the pulse from the output of the PWM control means, a comparing means for comparing the voltage at the second output end with a reference value, and an output of the comparing means for outputting the synchronization detecting means. A multi-output control power supply device comprising: holding means for updating and holding according to, and supplying as a control signal to the second switching means.

(9) Converter A switching regulator for switching the primary side of the transformer and supplying an output from the on-on winding of the secondary side thereof, comprising switching means connected in series to the on-on winding. A power supply voltage detecting means for detecting a power supply voltage of the converter transformer,
A switching regulator comprising: a control unit that outputs a control signal whose maximum duty changes in accordance with the output of the power supply voltage detection unit and controls ON / OFF of the switch unit.

(10) The control means comprises a comparator for comparing the output of the error amplifier with a triangular wave synchronized with the switching of the primary side of the converter transformer, and the output level of the error amplifier is changed by the output of the power supply voltage detecting means. The switching regulator according to (9) above.

(11) The control means is provided with a comparator for comparing the output of the error amplifier and the triangular wave synchronized with the switching of the primary side of the converter transformer, and the level of the triangular wave is changed by the output of the power supply voltage detecting means. The switching regulator according to (9) above.

(12) The one-chip I is provided with a digital circuit such as a CPU, a ROM, a RAM, or an analog circuit such as a DA converter necessary for controlling the switching regulator.
The switching regulator according to (9) above, which has been converted to C.

(13) Primary winding, first secondary winding and second
Having a secondary winding, a first switching element for switching the primary winding, and a switching operation of the first switching element controlled according to the output of the first secondary winding. Feedback control means for
A second switching element for switching the output of the second secondary winding, a power supply voltage detecting means for detecting an input voltage to the primary winding, and the first switching element according to the output of the power supply voltage detecting means. And a control means for controlling the maximum duty of the second switching element.

(14) Primary winding, first secondary winding and second
A converter transformer having a secondary winding, and first switching means for connecting and disconnecting the primary winding and its power source,
Rectifying / smoothing means for rectifying / smoothing the output of the first secondary winding and supplying it to the first output terminal, and PWM controlling the first switching means according to the voltage of the first output terminal. PWM control means, rectification means connected to one end of the second secondary winding, tapped or two-winding choke coil for supplying the output of this rectification means to a second output end, and this choke coil Second switching means for connecting and disconnecting between one end of the tap or two windings and the other end of the second secondary winding, and the voltage at the second output terminal is compared with a reference value, and the comparison output Comparing means for supplying a control signal to the second switching means, the other end of the two windings, and the second
And a flywheel diode connected between the other ends of the secondary windings of FIG.

(15) Primary winding, first secondary winding and second
A converter transformer having a secondary winding, and first switching means for connecting and disconnecting the primary winding and its power source,
Rectifying / smoothing means for rectifying / smoothing the output of the first secondary winding and supplying it to the first output terminal, and PWM controlling the first switching means according to the voltage of the first output terminal. PWM control means, rectifying means connected to one end of the second secondary winding, two-winding choke coil for supplying the output of the rectifying means to the second output end, and two-winding choke Second switching means for connecting and disconnecting a common connection point between the coil and the rectifying means and the other end of the second secondary winding; and an end opposite to the rectifying means connecting side of the second winding and the second winding. A flywheel diode connected between the other ends of the secondary windings,
A multiple output control power supply device comprising: a comparison unit that compares the voltage at the second output terminal with a reference value and supplies the comparison output to the second switching unit as a control signal.

[0026]

According to the configurations (1) to (8), the second switching means is turned on and off during the zero voltage period. According to the configurations (9) to (13), the maximum output value is not influenced by the power supply voltage. (14), (15)
According to this configuration, the loss of the control circuit of the second switching means is small, and the power supply for the control circuit is unnecessary.

[0027]

EXAMPLES The present invention will be described in detail below with reference to examples.

(A) Embodiments of the invention described in claims 1 to 8.

(Embodiment 1) FIG. 1 is a circuit diagram of a "multiple output control power supply device" which is Embodiment 1. In FIG. In the figure, the + output end that rectified and smoothed the AC line input is 1 of the transformer T1.
It is connected to one end of the N1 winding on the next side. The other end of the N1 winding is a switching transistor (FET in this embodiment).
It is connected to the drain of Tr1 and the source of FET • Tr1 is connected to the rectifying / smoothing-output terminal. FET ・ T
A resonant capacitor C1 is connected in parallel between the drain and source of r1. This is to resonate with the inductance of the N1 winding and efficiently transmit the power to the secondary side of the transformer T1. The pulse signal for driving the FET • Tr1 is generated by the PWM control circuit 1 and drives the gate of the FET • Tr1 via the drive circuit 2. In this embodiment, the PWM control circuit 1 is the transformer T1 2
The drive circuit 2 includes an insulating means such as a transformer and a photocoupler to be placed on the next side. N on the secondary side of the transformer T1
The two windings are rectified and smoothed by the rectifier diode D3 and the capacitor C3 and output as V2, and at the same time the PWM
The pulse width is input to the control circuit 1 to determine the pulse width. Further, the rectifying diode D3 of the N2 winding is connected so as to be non-conductive when the FET Tr1 is on and conductive when the FET Tr1 is off. That is, they constitute a flyback converter that drives the FET Tr1 so that the N2 winding output has a predetermined value.

On the other hand, a voltage according to the winding ratio of N1 and N3, the input AC line voltage, and the PWM duty ratio is also generated in the N3 winding on the secondary side of the transformer T1. The N3 winding is connected so that the rectifier diode D1 is conducted when the FET Tr1 is on. N
The 3-winding output is rectified by the rectifying diode D1 and supplied to the emitter of the switching transistor Tr2. The other end of the N3 winding is ground. The collector of the transistor Tr2 is connected to the cathode of the flywheel diode D2 and one end of the choke coil L1.
The other end of is connected to one end of the capacitor C2 and, at the same time, is output as V1 to the outside. The output V1 is a resistor R
The voltage is divided by 3 and R4 and input to the minus terminal of the comparator Q1. The + terminal of the comparator Q1 has a known voltage Vc
The divided voltage value by the resistors R5 and R6 of c is input. The output of the comparator Q1 is a D flip-flop (hereinafter referred to as D-F).
/ F) ・ It is input to Q2. The output of D-F / F-Q2 is transmitted through transistor Tr3 to transistor Tr2.
Turn on / off. Further, the PWM pulse signal of the PWM control circuit 1 is input to the clock of DF / F · Q2 via the synchronization detection circuit 3. The anode of the diode D2, the other end of the capacitor C2, the emitter of the transistor Tr3, etc. are at the ground level.

In the present embodiment, power is supplied to each control circuit on the secondary side from another auxiliary power supply (not shown).

Next, the operation of this embodiment will be described. FIG. 2 is a timing chart of this embodiment. In the figure,
(A) to (F) show waveforms at points (A) to (F) in FIG. That is, (A) is the output of the PWM control circuit 1, and is F when the N2 winding side output V2 is lower than a predetermined value.
The ON period of ET · Tr1 (the period of “L” in this figure) is lengthened, and when it is higher than a predetermined value, the ON period of FET · Tr1 is shortened. While the FET Tr1 is on, a voltage is generated at the non-ground end of the N2 winding, the rectifying diode D3 is non-conducting, no energy is released through the N2 winding, and the primary inductance is electromagnetic. Store energy. When the FET Tr1 is turned off, a flyback pulse is generated in the N1 winding, and at the same time, the voltage waveform of the N2 winding is inverted and the rectifying diode D
The energy stored in the primary inductance is released to the secondary side when 3 becomes conductive. The flyback pulse has a pulse width determined by the primary inductance, the resonance capacitor C1, and the winding capacitance, and eliminates switching loss by zero voltage switching that turns on the FET Tr1 when the voltage waveform is at zero level. The PWM control circuit 1 operates so that the flyback pulse width is constant during the OFF period of Tr1. That is, the duty control operation is performed in which the off period is constant and the pulse width of the on period is variable.

FIG. 2B shows the drain voltage waveform of the FET Tr1. Further, (C) shows the waveform of the N3 winding. If the rectifier diode D1 is turned on while the transistor Tr2 is on, energy is supplied to the load via the choke coil L1 and at the same time the capacitor C2 is supplied.
Is charged and the choke coil L1 is excited. Next, when the rectifier diode D1 becomes non-conductive, the flywheel diode D2 becomes conductive, and the excitation energy accumulated in the choke coil L1 and the charging energy of the capacitor C2 are supplied to the load. The above operation is repeated according to ON / OFF of the FET Tr1 to perform a normal forward conversion operation.

When the transistor Tr2 is off, the transformer T is irrespective of the voltage waveform generated in the N3 winding.
Since there is no energy supply from 1, the energy accumulated in the choke coil L1 and the capacitor C2 is continuously supplied to the load. The output voltage V1 on the N3 winding side is divided by the resistors R3 and R4 and connected to the minus input terminal of the comparator Q1. Further, the comparator Q1 divides the known voltage Vcc by the resistors R5 and R6 and compares the reference voltage supplied to the + input terminal with the divided voltage of V1. The output terminal of the comparator Q1 is connected to the D input terminal of the D-F / F · Q2, and the D-
The Q output terminal of the F / F · Q2 is connected to the base of the transistor Tr3, and the collector of the transistor Tr3 is connected to the base of the transistor Tr2.
The transistor Tr2 is turned on / off by the output of F · Q2. The PWM control circuit 1 is used for the D-F / F-Q2 clock.
Is output via the synchronization detection circuit 3.

For example, a case where the rising edge shown in FIG. 2A is detected and used as a clock will be described. As mentioned above, PW
While the output of the M control circuit 1 is "L", the FET Tr1 is on and the + voltage is generated in the winding N3. Therefore, if the detection result of V1 is lower than a predetermined value, the comparator Q1
Outputs "H". This comparison result is latched in DF / F · Q2 by the rise of (A) (see FIG. 2).
Medium [I] equivalent). By this latch result, FET ・ Tr
2 is turned on, but at this time, the FET Tr1 has already been turned off, and the rectifier diode D1 is in the non-conducting state because the winding N3 is generating a negative voltage, and the transistor Tr2 is turned on.
No current flows through. That is, the transistor Tr
The transition of 2 is basically performed when no voltage is applied to the emitter of the transistor Tr2. Then, when the FET Tr1 is turned on again and the N3 winding generates a + voltage, a current flows through the FET Tr2 to supply the current to the load and at the same time excite the choke coil L1 to charge the capacitor C2. Further, when the FET Tr1 is turned off, the voltage waveform of the N3 winding is inverted and the rectifying diode D1
Becomes non-conductive, and as a result, the current supply through the transistor Tr2 is lost, and the flywheel diode D2
Is conducted, the excitation energy accumulated in the choke coil L1 and the charging energy of the capacitor C2 are supplied to the load. The forward conversion operation as described above is repeated while the detection result of the output voltage V1 on the N3 winding side is lower than the predetermined value.

When the detection result of the output voltage V1 reaches a predetermined value, the output of the comparator Q1 is inverted. As a result, the output of the PWM control circuit 1 is delayed until the output rises, and D-
It is latched by F / F · Q2 (corresponding to [II] in FIG. 2).
The transistor Tr2 is turned off according to the latch result, but at this time, the FET Tr1 is already turned off, and N3
The winding is a transistor Tr2 while generating a voltage.
Turn off. That is, when the transistor Tr2 makes a transition, no switching loss occurs because no voltage is applied to the emitter. As a result, a highly efficient regulator can be constructed. When the transistor Tr2 is turned off, the flywheel diode D2 is turned on and the excitation energy accumulated in the choke coil L1 and the charging energy of the capacitor C2 are supplied to the load as in the forward operation described above. When all the excitation energy of the choke coil L1 is released, the voltage at the point (F) becomes the same potential as the output voltage V1, the flywheel diode D2 is turned off (corresponding to [III] in FIG. 2), and the capacitor C2 is turned on. Only the charging energy is released to the load, and the output voltage gradually decreases. When it falls below the set value of the comparator Q1, the output of the comparator Q1 is inverted and the transistor Tr2 is turned on again, and the above-described operation is repeated.

FIG. 10 shows a conventional method. Since the transistor Tr2 is immediately switched by the comparison result, as shown in the timing chart, in the switching transistor Tr2, the on / off Z,
Switching loss occurs in Z '. In particular, in the case of a large current switching transistor, many elements have a slow switching time, resulting in a large loss. On the other hand, according to the present embodiment, the efficiency is improved with the simple configuration as described above.

Although the switching transistor Tr2 is described as a bipolar transistor in the present embodiment, it may be any element capable of performing a switching operation, and may be constituted by an FET, for example.

(Embodiment 2) FIG. 3 shows a circuit of a main portion of Embodiment 2. In the first embodiment, synchronous detection is performed from the output of the PWM control circuit 1 and the clock of D-F / F.Q2 is used, whereas in the present embodiment, the delay circuit 4 is provided after the synchronous detection circuit 3 as illustrated. , Through this circuit DF / F ・ Q2
Is supplying the clock.

The output of the PWM control circuit 1 drives the transistor on the primary side through the driver 2 and further drives the transformer as shown in the first embodiment, and as a result, an output waveform is generated in the winding on the secondary side. Therefore, there may be a considerable delay between the output waveform of the PWM control circuit 1 and the output waveform of the secondary winding N2. Furthermore, the switching transistor Tr
Depending on the switching time of 2, P as in the first embodiment
If the output of the WM control circuit 1 is directly used as the clock of DF / F · Q2, the switching transistor Tr2 may be switched during the period when the + voltage is generated in the N3 winding. Therefore, the delay circuit 4 sets an appropriate delay P
By providing the output of the WM control circuit 1 with the clock of DF / F · Q2, it is possible to realize a configuration in which no switching loss occurs even in such a case.

(Embodiment 3) FIG. 4 shows a circuit diagram of essential parts of Embodiment 3. In this embodiment, the output current is detected by the resistor Rx, and when the voltage drop exceeds the Vbe of the transistor Tr4, the transistor Tr4 is turned on and the clear terminal of DF / F.Q2 is forced to "L". The transistor Tr2 is turned off and the energy supply from the N3 winding to the load is stopped. As a result, overcurrent protection can be realized. Further, the idle period is set by the time constant of the resistor R7 and the capacitor C3.

(Fourth Embodiment) FIG. 5 is a block diagram of the essential parts of the fourth embodiment. As shown, CPU6, ROM7, R
Along with the digital circuit such as AM8 and timer and the analog circuit such as A / D converter, the above-mentioned control circuits 1, 3, 4, Q
1, Q2 etc. are integrated in one chip. By doing this,
For example, while performing sequence control of a copying machine, a printer, etc., power supply control adapted to the state can be performed. For example, if the delay circuit 4 is composed of a programmable counter,
Even when the delay in which the voltage waveform is generated in the emitter of the transistor Tr2 changes after the PWM control circuit 1 outputs a pulse in the load state, the CPU 6 keeps track of the load state and sets the optimum value in the programmable counter. Therefore, the loss of the transistor Tr2 can be minimized. Further, low power consumption is possible by switching the comparison (reference) voltage of the comparator Q1 between the steady state and the stationary state. Furthermore, the overcurrent protection is controlled by the CPU via the set / reset register 9.
With the configuration in which the CPU 6 takes in and resets from the CPU 6, the CPU 6 can arbitrarily set the rest period. By integrating them on the same chip in this way, more versatile and more intelligent power control becomes possible.

(Embodiment 5) Embodiments 1 to 4 are intended to reduce the switching loss by turning on and off the switching transistor Tr2 at zero voltage.
However, since the base current of the switching transistor Tr2 is obtained from the output of the chopping N3 winding, the zero voltage cannot be turned on in principle. Examples 5 to 7 described below solve this problem.

FIG. 6 is a circuit diagram of a "multiple output control power supply device" which is a fifth embodiment.

The + output obtained by rectifying and smoothing the AC line input is supplied to one end of the primary side N1 winding of the transformer T1.
The other end of the N1 winding has a switching transistor Q11.
Of the transistor Q11, and the emitter of the transistor Q11 is connected to the rectified / smoothed-output. A resonance capacitor C1 is provided between the collector and the emitter of the transistor Q11.
Is inserted. This is to resonate with the inductance of the N1 winding and efficiently transmit the power to the secondary side of the transformer T1. The pulse signal for driving the transistor Q11 is generated by the PWM control circuit 1 and drives the base of the transistor Q11 via the drive circuit 2. In this embodiment, since the PWM control circuit 1 is placed on the secondary side, the drive circuit 2 includes an insulating means. The N2 winding on the secondary side of the transformer T1 is rectified and smoothed by the diode D3 and the capacitor C3, output as V2, and at the same time input to the PWM control circuit 1 to determine the pulse width.
In addition, the rectifier diode D3 of the N2 winding is the transistor Q1.
The flyback converter is connected so that when 1 is on, it is non-conductive, and when it is off, it is conductive, that is, it drives the transistor Q11 so that the N2 output has a predetermined value.

On the other hand, the N3 winding on the secondary side of the transformer T1 also has a winding ratio of N1 and N3, an input AC line voltage, and P.
A voltage corresponding to the duty ratio of WM is generated. Also, N3
The winding is connected so that the rectifier diode D1 is conductive when the transistor Q11 is on. The N3 winding voltage is rectified by the rectifying diode D1. The cathode of the diode D1 is connected to the cathode of the flywheel diode D2 and one end of the choke coil L1.
The other end of 1 is connected to one end of the capacitor C2 and, at the same time, is output to the outside as V1. This output is a resistor R3
And R4, and the voltage is input to the + terminal of the comparator Q1. The negative terminal of the comparator Q1 has a known voltage Vre.
f is input. The output of the comparator Q1 is DF / F
・ It is input to Q2. The output of the D-F / F-Q2 is transmitted through the transistors Q14 and Q13 to the transistor Q described later.
Turn 12 on / off. In addition, the PWM of the PWM control circuit 1
The pulse signal is input to the clock of D-F / F · Q2 via the synchronization detection circuit 3.

The other end of the N3 winding has a transistor Q
12 is connected to the collector of the transistor Q12, the emitter of the transistor Q12, the anode of the diode D2, the other end of the capacitor C2, etc. are at the ground level. In addition, the transistor Q
The emitter and the like of 13 are connected to the negative power source -Vss. The secondary side control circuits 1, 2, 3, Q1, Q2, etc. are supplied with power from another auxiliary power supply (not shown).

Next, the operation of this embodiment will be described. FIG. 7 shows a timing chart of this embodiment. In FIG.
(A) is the output of the PWM control circuit 1. When the output V2 on the N2 winding side is lower than a predetermined value, the ON period of the transistor Q11 (the "L" period in this figure) is lengthened and is higher than the predetermined value. In this case, the transistor Q11 operates so as to shorten the ON period. Transistor Q11 is on,
A voltage is generated in the N2 winding, and the rectifier diode D3
Is non-conductive, no energy is released through the N2 winding, and electromagnetic energy is stored in the primary inductance. Then, when the transistor Q11 is turned off, a flyback pulse is generated in the N1 winding, the voltage waveform of the N2 winding is inverted at the same time, and the energy stored in the primary inductance is transferred to the secondary side by conducting the rectifying diode D3. discharge. The flyback pulse has a pulse width determined by the primary inductance, the resonance capacitor C1, and the winding capacitance, and when the voltage waveform is zero level, the transistor Q11
In order to eliminate the switching loss by the zero voltage switching for turning on, the PWM control circuit 1 operates so that the flyback pulse width is constant during the off period of the transistor Q11. That is, the control operation is performed in which the pulse width of the on period is variable while the off period is constant.

FIG. 7B shows the collector waveform of the transistor Q11. Further, (c) shows the waveform of the N3 winding. If the rectifier diode D1 conducts while the transistor Q12 is turned on, energy is supplied to the load via the choke coil L1 and at the same time the capacitor C2 is supplied.
Is charged and the choke coil L1 is excited. Next, when the rectifier diode D1 becomes non-conductive, the flywheel diode D2 becomes conductive, and the excitation energy accumulated in the choke coil L1 and the charging energy of the capacitor C2 are supplied to the load. The above-described operation is repeated in accordance with ON / OFF of the transistor Q11 to perform a normal forward conversion operation. When the transistor Q12 is off, energy is not supplied from the transformer T1 regardless of the voltage waveform generated in the winding N3, so that the energy accumulated in the choke coil L1 and the capacitor C2 is continuously supplied to the load. The output voltage V1 is divided by the resistors R3 and R4 and supplied to the + input of the comparator Q1. Also, the comparator Q1
Compares the reference voltage Vref supplied to the-input with said voltage. The output of the comparator Q1 is DF / F · Q2
Is supplied to the D input of the transistor Q14, and the output of the D-F / F · Q2 is supplied to the base of the transistor Q14.
The emitter of is connected to the positive power supply, and transistor Q14
The collector of is connected to the base of transistor Q13,
Further, the collector of the transistor Q13 is the transistor Q
It is connected to the base of 12 and turns on / off the transistor Q12 by the output of DF / F · Q2. DF / F
The output of the PWM control circuit 1 is input to the clock of Q2 via the synchronization detection circuit 3.

For example, a case where the rising edge of FIG. 7A is detected and used as a clock will be described. As mentioned above, PWM
The transistor Q11 is on while the output of the control circuit 1 is "L", and the + voltage is generated in the N3 winding. Therefore, when the detection result of the output voltage V1 is lower than the predetermined value, the comparator Q1 outputs "L". This comparison result is latched in DF / FQ2 at the rising edge of (a) (corresponding to [I] in the figure). The latching result turns on the switching transistor Q12, but at this time the transistor Q11 is already turned off, and the rectifier diode D1 is in the non-conduction state because the winding N3 is generating a negative voltage. Current does not flow. That is, the transition of the transistor Q12 is basically performed when no voltage is applied between the emitter and collector of the transistor Q12. Then, when the transistor Q11 is turned on again and the N3 winding generates a + voltage, a current flows through the transistor Q12 as shown in (g) to supply the current to the load and at the same time to supply the current to the choke coil L.
1 is excited and the capacitor C2 is charged. Further, when the transistor Q11 is turned off, the voltage waveform of the N3 winding is inverted, the rectifying diode D1 becomes non-conducting, and as a result, the current supply through the transistor Q12 disappears, and the flywheel diode D2 becomes conducting, thereby making the choke coil. The excitation energy stored in L1 and the charging energy of the capacitor C2 are supplied to the load. The forward conversion operation as described above is repeated while the detection result of the output voltage V1 on the N3 winding side is lower than the predetermined value.

When the detection result of the output voltage V1 reaches a predetermined value, the output of the comparator Q1 is inverted, but this result is delayed until the rise of the PWM control circuit 1 output,
It is latched by DF / F · Q2 (corresponding to [II] in FIG. 7). The latching result turns off the switching transistor Q12.
Is off, and the N3 winding turns off the transistor Q12 while generating a-voltage. That is, when the transistor Q12 makes a transition, no switching loss occurs because no voltage is applied between its emitter and collector. As a result, a highly efficient regulator can be constructed. When the transistor Q12 is turned off, the flywheel diode D2 is turned on to supply the excitation energy accumulated in the choke coil L1 and the charging energy of the capacitor C2 to the load, as in the forward operation described above. When all the excitation energy of the choke coil L1 is released, the voltage at the point (X) becomes the same potential as the output voltage, the flywheel diode D2 is turned off (corresponding to [III] in FIG. 7), and the capacitor C2 is turned on. Only the charging energy is released to the load, and the output voltage gradually decreases. When it falls below the set value of the comparator Q1, the output of the comparator Q1 is inverted and the transistor Q12 is turned on again, and the operation described above is repeated.

In this embodiment, the base current of the switching transistor Q12 is obtained from the auxiliary power source -Vss,
Complete zero voltage on can be realized. Further, since the switching transistor is provided on the low voltage side, the loss of the base current limiting resistor can be small. In addition, in this embodiment,
Although the switching transistor Q12 has been described as a bipolar transistor, any element capable of performing a switching operation may be used, and for example, an FET may be used.

(Sixth Embodiment) FIG. 8 shows a sixth embodiment. In the fifth embodiment, the switching element Q12 is realized by the PNP type transistor, but the PNP type is now more than the NPN type.
High price and poor availability. Therefore, the photo coupler PH1
Is used to obtain a function equivalent to that of the fifth embodiment with an NPN transistor.

(Seventh Embodiment) FIG. 9 shows a seventh embodiment. While the fifth embodiment requires an external auxiliary power supply, by adding it to the diode D4, a certain output voltage V1 can be output even when the transistor Q12 is off at the time of rising. Therefore, this V1 can be used as a substitute for the auxiliary power supply, and the cost can be reduced.

As described above, Examples 1 to 7
According to this, the transistor Tr2 can be switched between conducting and non-conducting in the zero voltage period, and the efficiency is simple and the structure is good. (B) An example of the invention according to claims 9 to 13.

In the first to seventh embodiments, when power is supplied from the forward winding (see N3 in FIG. 11) to a load such as a motor to a certain extent or more for the reason described later, the falling side of the resonance voltage waveform (See period A in FIG. 12)
It has the drawback that it will not zero-cross. As a countermeasure against this, the configuration shown in FIG. 11 can be considered. That is, Examples 1 to 7 are examples that are applied when medium power is required, and Examples 8 to 10 below are examples that are suitable when higher power is required.

In the voltage resonance type switching regulator of FIG. 11, in order to cope with a high power pulse load such as a motor, a switching element Q4 is provided on the secondary side, and
Pulse width control is performed at the same frequency in synchronization with the secondary switching element Q1. In addition, a configuration is used in which the timing of turning on the switching element Q4 is prohibited during the period in which the flyback waveform of the switching element Q1 is generated.

The configuration of this voltage resonance type switching regulator will be described below with reference to FIG. In the figure, T
1 is a converter transformer, and this converter transformer T
The voltage Vin is supplied to one end of the primary winding N1 of No. 1.
This "Vin" is the power supply voltage of the converter transformer T1 and also the input voltage of the converter transformer T1 when the switching element Q1 is on. However, when expressed as "input voltage", it is confused with the input voltage of the converter transformer T1 when the switching element Q1 is off. On the other hand, since this Vin is also the power supply voltage of the switching regulator, "Vin" is hereinafter referred to as the power supply voltage. The other end of the primary winding N1 is the FE which is a switching element.
It is connected to the drain of TQ1. The source of this switching element FET Q1 is grounded. A voltage according to the winding ratio is generated in the secondary windings N2 and N3 by the switching of the switching element FET.Q1. Two
One end of the next winding N3 is connected to the anode of the rectifier diode D1, and the rectifier diode cathode is connected to the PWM control circuit 1
And a choke coil L1 of the flywheel diode D2 via a switching element Q4 controlled by
Connected to one end of. The other end of the secondary winding N3 is connected to the COM potential. The other end of the choke coil L1 is connected to one end of the output capacitor C2. The other ends of the flywheel diode D2 and the output capacitor C2 are connected to the COM potential.

Further, N2 is a main winding on the secondary side, and the PWM control circuit 1 is used to change the switching duty of the switching element Q1 in order to make the voltage obtained by rectifying and smoothing the output thereof constant. . With this configuration,
The capacity of this type of voltage resonance type switching regulator has been increased. FIG. 12 shows the waveform of each part in this switching regulator. The details will be described later.

The circuit operation will be described below with reference to FIGS. 11 and 12. For convenience of description, first, the operation when the switch element Q4 is not provided will be described.

When the switch element Q4 does not exist,
The resonance voltage Vds (voltage between the drain and source of the switching element Q1) does not cross zero as the load on the secondary winding N3 side increases, and power cannot be transmitted. The reason is that the forward side diode D1 of the on-on side output is turned on during the period when the resonance voltage Vds is equal to or lower than the power supply voltage Vin, and there are the following two periods.

The switching element Q1 is off, the resonance voltage Vds <the power supply voltage Vin, and the resonance current is flowing from the capacitor C1 to the power supply Vin (period A in FIG. 12). During this period, the resonance current I (RES) is flowing from the capacitor C1 to the power supply Vin. On the other hand, the load current Ip (on / on) of the on-on side winding N3 tends to flow from the power supply Vin to the capacitor C1 with a magnitude of Io * (Ns / Np) when viewed from the primary side. Therefore, the resonance current (direction flowing from the capacitor C1 to the power supply Vin) is apparently small, the voltage drop is slowed down, and the zero crossing becomes difficult.

The switching element Q1 is off, the resonance voltage Vds <the power supply voltage Vin, and the resonance current is the power supply Vin.
Period from time to capacitor C1 (period B in FIG. 12). During this period, the resonance current I (RES) and the load current Ip (on / on) of the on-on side winding N3 are both strengthened in the direction of flowing into the capacitor C1 and the rising slope of Vds becomes sharp.

As described above, the resonance voltage Vds deviates from the sine wave as the load current is taken from the on-on side winding N3, and finally does not cross zero.

As measures against this, there are the following methods a and b.

A. Increase the resonance current I (RES),
The on-on side winding current Ipon / on) is relatively reduced. Specifically, the resonance capacitor C1 is increased or the inductance of the converter transformer T1 is decreased. However, in this case, the resonance frequency decreases, the exciting current of the converter transformer T1 increases, and the load on the switching element Q1 and the like increases, and the converter transformer T1 increases.
However, magnetic saturation is likely to occur, which leads to a larger transformer and is not the optimum method.

B. The ON-ON side winding current Ip (on / on) in the period A or the period B or both is stopped.

The circuit of FIG. 11 adopts the method of b, that is, it is turned on during the period A by using the switch element Q4.
The on-side winding current Ip (on / on) is prevented from flowing.

Next, the operation when the switch element Q4 is provided will be described with reference to FIG. In FIG. 12, Vx is the voltage of the on-on winding N3, and a substantially isosceles trapezoidal voltage appears on the positive side. Therefore, when the PWM control circuit 4 appropriately controls the ON period of the switch element Q4, the output voltage Vy of the switch element Q4 has a waveform blocking the Y region as shown in the figure, and the current I (N3 of the ON-ON winding N3 is generated. ) That is, the current flowing through the rectifying diode D1 becomes 0 in the period A. Therefore, as described above, the on-on winding Ip (on / on) and the resonance current I (RES) do not interfere with each other in the period A. As a result, the Vds waveform approaches a sine wave (strictly, it is necessary to block the period B as well), but it is possible to secure the zero-cross condition even if the load current is several times as large as before. That is, a large capacity can be realized.

However, in the above-mentioned voltage resonance type switching regulator, the PWM control circuit 4 forms a triangular wave based on the synchronization signal from the synchronization detection circuit 3 and compares it with the output of the error amplifier by the comparator to output the PWM signal. Is formed. Therefore, since the maximum duty of the PWM signal is constant regardless of the power supply voltage Vin, when the power supply voltage Vin increases, the secondary winding N
The maximum value of the output voltage V1 on the 3rd side increases and the power supply voltage V1
As in becomes smaller, the maximum value of output voltage V1 becomes smaller. Therefore, in order to secure the output voltage V1, the power supply voltage Vi
Since the maximum duty of the PWM signal is determined according to the case where n is the minimum, the maximum value of the output voltage V1 becomes too large when the power supply voltage Vin is the maximum, and the output becomes too high, and the power supply, the load, and the device concerned. There is a risk of damage to the switching elements of. This is particularly problematic because switching regulators are generally designed to handle power supplies in a wide voltage range.

An embodiment for solving this problem will be described in detail below.

(Embodiment 8) FIG. 13 is a circuit diagram of a "voltage resonance type switching regulator" according to an embodiment 8.
In the figure, T1 is a converter transformer. The power supply voltage Vin is supplied to one end of the primary winding N1 of the converter transformer T1. The other end of the primary winding N1 is connected to the drain of the FET Q1 which is a switching element. The source of this switching element FET Q1 is grounded. A voltage according to the winding ratio is generated in the secondary windings N2 and N3 by the switching of the switching element FET.Q1. One end of the secondary winding N3 is connected to the anode of the rectifying diode D1 and the other end is connected to the COM potential. The cathode of the rectifier diode D1 is passed through the switch element Q4 driven by the PWM control circuit 14 and the pulse width limiting circuit 17 to the flywheel diode D2.
Is connected to one end of the choke coil L1.
The other end of the choke coil L1 is connected to one end of the output capacitor C2. The other ends of the flywheel diode D2 and the output capacitor C2 are connected to the COM potential. The pulse width limiting circuit 17 limits the maximum duty of the switch element Q4 according to the power supply voltage Vin detected by the power supply voltage detection circuit 16. The PWM control circuit 14 is
PWM synchronized with the drive circuit 2 by the synchronization detection circuit 3
It controls.

The secondary winding N2 is the main winding on the secondary side, and the PWM control circuit 1 is used to set the switching duty of the switching element Q1 in order to make the output voltage rectified and smoothed. It is changing.

As described above, the circuit of this embodiment is different from the circuit of FIG. 11 in that the power supply voltage detection circuit 16 and the pulse width limiting circuit 17 are provided. Therefore, the difference will be described with reference to the detailed circuit diagram shown in FIG. As shown in the figure, a synchronization signal is received from the synchronization detection circuit 3, a triangular wave is created thereby, and the output of the error amplifier AMP18 is compared by a comparator COMP19 to perform PWM control of the switch element Q4. Here, a method of determining the maximum duty of the comparator COMP19 by the resistors R5 and R6 is generally performed. However, in this method, since the maximum duty is constant, the maximum value of the output voltage V1 increases when the power supply voltage Vin of the converter transformer increases, and conversely, the maximum value of the output voltage V1 decreases when the power supply voltage Vin decreases. . Therefore, in order to secure the output voltage V1, when the values of the resistors R5 and R6 are determined in accordance with the minimum power supply voltage Vin, when the power supply voltage Vin is maximum, the maximum value of the output voltage V1 becomes too large and the power supply, Obviously, this is not desirable considering the protection of the load and the switching elements of the switching regulator.

Therefore, in this embodiment, the power supply voltage detection circuit 1
The power supply voltage Vin is detected by 6 and the maximum duty of the switch element Q4 is made to depend on the power supply voltage Vin by this voltage, and the maximum value of the output voltage V1 is made constant regardless of the change of the power supply voltage Vin. That is, FIG.
4, the elements N4, D4, C4 and R8 make the power supply voltage Vi
n is detected and this voltage is supplied to the error amplifier A through the resistor R7.
When the voltage is supplied to the output side of MP18 and the power supply voltage Vin increases, the voltage at the negative input terminal of the comparator COMP19 increases, and the duty of the switch element Q4 decreases so that the maximum duty is dependent on the power supply voltage Vin. As a result, the maximum value of the output voltage V1 becomes constant regardless of the change in the power supply voltage Vin.

In this way, in this embodiment, regardless of the change in the power supply voltage Vin of the converter transformer T1,
Since the maximum value of the output voltage V1 is constant, the power source, the load, the switching element Q1, etc. are not overloaded.

Further, since the maximum output power does not depend on the power supply voltage, the switching regulator can be miniaturized and reduced in cost in terms of thermal design.

(Embodiment 9) FIG. 15 is a detailed circuit diagram of this embodiment. As shown in the figure, in this embodiment, the voltage amplitude of the triangular wave is changed by the transistor Q5 controlled by the output of the power supply voltage detection circuit 16, and the switching element Q5 is changed.
The maximum duty in No. 4 has power supply voltage dependency so that the output voltage V1
The maximum value of is kept constant. In this way, the same effect as that of the eighth embodiment can be obtained in the present embodiment.

(Embodiment 10) FIG. 16 is an explanatory diagram of this embodiment. The PWM control circuits 1 and 4, the synchronization detection circuit 3, the pulse width limiting circuit 17, and the power supply voltage detection circuit 1 within the dotted line in the figure
6 and the like can be made into a one-chip IC, and a part of the circuit can be made soft, so that it can be downsized and generalized.

Although the above eighth to tenth embodiments are voltage resonance type switching regulators, the present invention is not limited to them and can be implemented in other types of switching regulators. Further, in each embodiment, the comparator compares the voltage levels, but the present invention is not limited to this, and the current levels may be compared.
As described above, according to the eighth to tenth embodiments, it is possible to provide a multiple output control power supply having a simple structure and high efficiency. Further, the power supply, the load, and the switching element are not overloaded, and the reliability of the device is improved. Further, since the upper limit of pulse width * voltage can be limited without depending on the power supply voltage, it becomes possible to cope with the delay of control response to the pulse load. Further, since the maximum output voltage does not depend on the power supply voltage, the size and cost can be reduced in terms of thermal design.

(C) An embodiment of the invention according to claims 14 and 15.

In Examples 1 to 4, since the switching element is provided on the high voltage side, the loss of the base current limiting resistance is large. In Examples 5 to 7, since the switching element is provided on the low voltage side, the loss of the base current limiting resistance is small, but a power source for the base drive is required. Embodiments 11 and 12 described below solve this problem.

(Embodiment 11) FIG. 17 is a circuit diagram of the "multiple output control power supply device" according to Embodiment 11. In FIG. In the figure, T1 is a converter transformer. The power supply voltage Vin is supplied to one end of the primary winding N1 of the converter transformer T1. The other end of the winding N1 is connected to the collector of the transistor Q21 which is a switching element. The emitter of this transistor Q21 is grounded. The secondary winding N is switched by the switching of the transistor Q21.
At 3, a desired voltage is generated according to the winding ratio. Winding N3
Has one end connected to the anode of the rectifying diode D1 and the other end connected to the COM (common) potential point. Rectifier diode D
The cathode of No. 1 is connected to the cathode of the flywheel diode D2 and one end of the choke coil L21 with a tap. The other end of the choke coil L21 has an output capacitor C2.
Connected to 1. The anode of the flywheel diode D2 and the other end of the capacitor C21 are connected to the COM potential point.

The cathode of the diode D23 is connected to the tap of the choke coil L21, the emitter of the switching transistor Q22 is connected to the anode thereof in series, and the collector thereof is connected to the COM potential point. Further transistor Q
A drive circuit 22 is composed of a transistor Q23, a comparator COMP1, and an error amplifier AMP1.

The winding N2 is the main winding on the secondary side, and has a CTL for rectifying and smoothing the output of the winding N2 to make it a constant voltage.
The (control) circuit 21 is used to change the switching duty of the transistor Q21.

The circuit operation will be described below. If the transistor Q22 is always off, the circuit of the present embodiment becomes the average value rectification as in the usual on / on converter, and the output voltage V1 is as follows.

V1 = Vin * {Ton1 / (Ton1 + Toff1)} * x ... If the transistor Q22 is always on, the circuit shown in FIG.
Equivalent to 8, V1 = Vin * {Ton1 / (Ton1 + a * Toff1)} * x, where a = n4 / n5 x = n3 / n1 Ton1: ON time of switch 1 (Q21) Toff1: of switch 1 Off-time n □: Number of turns of N □ winding Here, the case of a = 1 is a normal on / on converter (see the following for proof).

Next, the operation of this embodiment will be described with reference to the circuit of FIG. 19 which is a generalized main part of this embodiment and FIG. 20 which is a timing chart thereof. First, in the circuit of FIG. 19, while the switch SW1 is off, the switch S
Consider the case where W2 turns on / off.

However, Ton2: Effective ON time of the switch SW2 (t1 to t2 in FIG. 20) Toff2: Effective OFF time of the switch SW2 (t2 to t3 in FIG. 20) Ton2 + Toff2 = Toff1 i1: Current flowing through the rectifying diode D1 (I1 in FIG. 20
I2: current flowing through flywheel diode D2 (see i2 in FIG. 20) i3: current flowing through sub-flywheel diode D2 (see i3 in FIG. 20) L4: inductance of n4 winding L5: inductance of n5 winding The forward voltage Vf of the diode is neglected for simplicity.

One cycle will be described by dividing it into the following three periods (1), (2) and (3).

Here, L4> L5 is required, which is a step-down type.

(1) t1 to t1 (SW1: ON) While the switch SW1 is on, i3 = 0 regardless of whether the switch SW2 is on or off. That is, it is the same as N5 does not exist, and the change amount of i1 Δi1
Is Δi1 = [{V (n3) -V2} / L4] * Ton1 However, V (n3) = Vin * (n3 / n1), and Δ12 and Δ13 are clearly Δi2 = 0 Δi3 = It becomes 0.

(2) t1 to t2 (SW1: OFF, S
W2: ON) While the switch SW1 is off and the switch SW2 is on, the flywheel current of the choke coil flows in L5 having a small inductance value and does not flow in L4 having a large inductance value, so Δi1 = 0 Δi2 = 0 Δi3 = − [V1 / L5] * Ton2.

(3) t2 to t3 (SW1: OFF, S
W2: OFF) While the switch SW1 is off and the switch SW2 is off, the flywheel current of the choke coil is switched from i3 to i2. Δi1 = 0 Δi2 =-[V1 / L4] * Toff2 Δ i3 = 0.

Under the condition that the current of the output choke is continuous (more than the critical current), (Δi1 * n4 + Δi2 * n4 + Δi3 * n5) is 1
Considering that the cycle becomes 0, n4 * {V (n3) -V1} / L4 * Ton1-n5 * V1 / L5 * Ton2-n4 * V1 / L4 * Toff2 = 0 L4 / L5 = (n4 / n5) ² ... Solving V1 = Vin * (n3 / n1) * [Ton1 / {Ton1 + (n4 / n5) * Ton2 + Toff2}] Ton2 + Toff2 = Toff1 .. Then, if Ton2 = 0, then the formula becomes, and Toff2 = 0 If it does, it becomes a formula.

That is, by controlling the on / off timing of the switch SW2, if n4> n5, V1 (max) = Vin * x * {Ton1 / (Ton1 + Toff1)} V2 (min) = Vin * x * {Ton1 / (Ton1 + (n4 / n5) Toff1)} However, the output voltage V1 can be controlled to a constant voltage in the range of x = n3 / n1.

As can be seen from FIG. 21, the switch S
If W2 is turned off to an impedance of (n4 / n5), it commutates to the flywheel diode D2 (i3 commutates to i2), so that the switching loss is small.

FIG. 21 is a principle view of the eleventh embodiment, which is a step-down type using a choke coil with a tap as described above.

As described above, according to the present embodiment, it is possible to obtain a multiple output control power supply device having a relatively simple structure that does not require synchronization and has a low loss and does not require a power supply for a base drive.

(Embodiment 12) The same operation as that of Embodiment 11 can be realized by a boost type as shown in FIG. This will be described as Example 12. In this embodiment, L4 <L5 is required.

(1) t0 to t1 (SW1: ON) Similar to the step-down type, the variation Δi1 of i1 is Δi1 = [{V (n3) −V1} / L4] * Ton1 where V (n3 ) = Vin * (n3 / n1), and Δi2 and Δi3 are obviously Δi2 = 0 and Δi3 = 0.

(2) t1 to t2 (SW1: OFF
SW2: ON) While the switch SW1 is off and the switch SW2 is on, the flywheel current of the choke coil flows in L4 having a small inductance value and does not flow in L5 having a large inductance value, so Δi1 = 0 Δi2 = − [V1 / L4] * Ton2 Δi3 = 0.

(3) t2 to t3 (SW1: OFF
SW2: OFF) While the switch SW1 is off and the switch SW2 is off, the flywheel current of the choke coil is switched from i2 to i3. Δi1 = 0 Δi2 = 0 Δi3 =-[V1 / L5] * Toff2.

Under the condition that the current of the output choke is continuous (above the critical current), (Δi1 * n4 + Δi2
Considering that * n4 + Δi3 * n5) becomes 0 in one cycle, n4 * {V (n3) -V1} / L4 * Ton1-n4 * V1 / L4 * Ton2-n5 * V1 / L5 * Toff2 = 0 L4 / L5 = (n4 / n5) ² ... Solving V1 = Vin * (n3 / n1) * [Ton1 / {Ton1 + Ton2 + (n4 / n5) * Toff2}] Ton2 + Toff2 = Toff1 ... In other words, the switch SW2 By controlling the timing of turning on / off of, if n4> n5, V1 (min) = Vin * x * {Ton1 / (Ton1 + Toff1)} V2 (max) = Vin * x * {Ton1 / (Ton1 + (n4 / n5) Tff1)} However, V1 can be controlled to a constant voltage in the range of x = n3 / n1.

This embodiment can be modified as shown in FIG.

In the eleventh and twelfth embodiments,
Switching frequency of switch SW2 << switch SW
The same switching frequency can be used. Further, as shown in FIG. 24, a half bridge circuit can also be used. In the eleventh and twelfth embodiments, the point is to switch the inductance of the choke coil, and it does not depend on the rectifying means on the secondary side. Therefore, the same operation is performed in a half bridge circuit other than the forward converter.

[0107]

As described above, according to the inventions of claims 1 to 8, claim 14 and claim 15, a multiple output control power supply device having a simple structure and high efficiency can be obtained. According to the inventions of claims 9 to 13, it is possible to obtain a switching regulator having a maximum output voltage and a multiple output control power supply device having a simple structure, high efficiency, and no power supply voltage dependency.

[Brief description of drawings]

FIG. 1 is a circuit diagram of a first embodiment.

FIG. 2 is a timing chart of the first embodiment.

FIG. 3 is a circuit diagram of a main part of the second embodiment.

FIG. 4 is a circuit diagram of a main part of the third embodiment.

FIG. 5 is a circuit diagram of a main part of the fourth embodiment.

FIG. 6 is a circuit diagram of a fifth embodiment.

FIG. 7 is a timing chart of the fifth embodiment.

FIG. 8 is a circuit diagram of a sixth embodiment.

FIG. 9 is a circuit diagram of Example 7.

FIG. 10 is an explanatory diagram of a conventional method.

FIG. 11 is a circuit diagram for explaining Examples 8 to 10.

FIG. 12 is a diagram showing waveforms at various parts in the circuit of FIG.

FIG. 13 is a circuit diagram of Example 8.

FIG. 14 is a detailed circuit diagram of Example 8.

FIG. 15 is a detailed circuit diagram of the ninth embodiment.

16 is an explanatory diagram of Example 10. FIG.

FIG. 17 is a circuit diagram of the eleventh embodiment.

FIG. 18 is a schematic diagram when the Q22 is on in the eleventh embodiment.

FIG. 19 is a generalized principle diagram of the eleventh embodiment.

FIG. 20 is a timing chart of the circuit of FIG.

FIG. 21 is a principle diagram of Example 11

22 is a principle diagram of Example 12. FIG.

FIG. 23 is a principle diagram of a modification of the twelfth embodiment.

FIG. 24 is an explanatory diagram of a modification of the eleventh and twelfth embodiments.

[Explanation of symbols]

 1 PWM control circuit 3 Synchronization detection circuit N1 Primary side winding N2 Secondary side winding N3 Secondary side winding T1 Transformer Tr1 FET Tr2 Transistor Q1 Comparator Q2 D flip-flop

─────────────────────────────────────────────────── ─── Continuation of the front page (51) Int.Cl. ⁵ Identification code Office reference number FI technical display location B41J 29/46 J 9113-2C // G03G 15/00 102

Claims (15)

[Claims] 1. A transformer having a primary winding, a first secondary winding, and a second secondary winding, a first switching element for switching the primary winding, and the first switching element. Feedback control means for controlling the switching operation of the switching element according to the output of the first secondary winding, a second switching element for switching the output of the second secondary winding, and the second switching element. A multiple output control power supply device, comprising: a synchronizing means for synchronizing the driving of the switching element with the driving of the first switching element. 2. The multiple output control power supply device according to claim 1, wherein the synchronization means transitions the second switching element during an off period of the first switching element. 3. A primary winding, a first secondary winding and a second secondary winding.
A transformer having a secondary winding, a first switching element for switching the primary winding, and a feedback for controlling the switching operation of the first switching element according to the output of the first secondary winding. Control means, a second switching element provided on the low-voltage side of the output of the second secondary winding and switching the output, and the second switching element is switched at a frequency lower than that of the first switching element. A multiple output control power supply device, comprising: 4. The multiple output control power supply device according to claim 3, wherein the driving means makes a transition of the second switching element during an off period of the first switching element. 5. A converter transformer having a primary winding, a first secondary winding, and a second secondary winding, and first switching means for connecting and disconnecting the primary winding and its power supply. Rectifying / smoothing means for rectifying / smoothing the output of the first secondary winding and supplying it to the first output terminal, and PWM controlling the first switching means according to the voltage of the first output terminal. PWM
Control means, rectifying means for rectifying the output of the second secondary winding, second switching means for intermittently supplying the output of the rectifying means to the second output terminal, and the PWM control means. A synchronization detecting means for extracting a signal synchronized with the pulse from the output, a comparing means for comparing the voltage at the second output terminal with a reference value, and an output of the comparing means are updated according to the output of the synchronizing detecting means. Holding means for holding and supplying the control signal to the second switching means as a control signal. 6. The multiple output control power supply device according to claim 5, further comprising delay means for delaying a control signal supplied from the holding means to the second switching means by a required time. 7. A circuit for PWM control means, synchronization detection means, and holding means is formed on the same chip together with digital circuits such as CPU, ROM, RAM and analog circuits such as DA converters. Item 5. The multiple output control power supply device according to Item 5. 8. A converter transformer having a primary winding, a first secondary winding, and a second secondary winding, and first switching means for connecting and disconnecting the primary winding and its power supply. Rectifying / smoothing means for rectifying / smoothing the output of the first secondary winding and supplying it to the first output terminal, and PWM controlling the first switching means according to the voltage of the first output terminal. PWM
Control means, rectifying means for rectifying the output of the second secondary winding and supplying the rectified output to the second output end, and second switching means for connecting and disconnecting the low voltage side of the output of the second secondary winding. A synchronization detection means for extracting a signal synchronized with the pulse from the output of the PWM control means, a comparison means for comparing the voltage at the second output end with a reference value, and an output of the comparison means for the synchronization detection means. Is updated and held according to the output of the second
And a holding means for supplying the switching means as a control signal. 9. A switching regulator for switching the primary side of a converter transformer and supplying an output from an on-on winding of its secondary side, comprising switching means connected in series to said on-on winding. Power supply voltage detection means for detecting the power supply voltage of the converter transformer, and control means for outputting a control signal whose maximum duty changes in accordance with the output of the power supply voltage detection means and controlling the switching means on / off. A switching regulator characterized in that 10. The control means comprises a comparator for comparing the output of the error amplifier with a triangular wave synchronized with the switching of the primary side of the converter transformer, and the output level of the error amplifier is changed by the output of the power supply voltage detecting means. The switching regulator according to claim 9, which is a thing. 11. The control means comprises a comparator for comparing the output of the error amplifier and a triangular wave synchronized with the switching of the primary side of the converter transformer, and the level of the triangular wave is changed by the output of the power supply voltage detecting means. The switching regulator according to claim 9, wherein: 12. The switching regulator according to claim 9, wherein a digital circuit such as a CPU, a ROM, a RAM, or an analog circuit such as a DA converter necessary for controlling the switching regulator is integrated into a one-chip IC. . 13. A primary winding, a first secondary winding, and a second 2
A transformer having a secondary winding, a first switching element for switching the primary winding, and a feedback for controlling the switching operation of the first switching element according to the output of the first secondary winding. Control means, a second switching element for switching the output of the second secondary winding, a power supply voltage detecting means for detecting an input voltage to the primary winding, and an output of the power supply voltage detecting means. And a control means for controlling the maximum duty of the second switching element accordingly. 14. A primary winding, a first secondary winding, and a second two.
A converter transformer having a secondary winding, a first switching means for connecting and disconnecting the primary winding and its power source, and an output of the first secondary winding is rectified and smoothed and supplied to a first output end. Rectifying / smoothing means for performing PWM control of the first switching means according to the voltage of the first output end.
M control means, rectifying means connected to one end of the second secondary winding, choked coil with taps or two windings for supplying the output of the rectifying means to the second output end, and this choke coil Second switching means for connecting and disconnecting between one end of the tap or two windings and the other end of the second secondary winding, and the voltage at the second output terminal is compared with a reference value, and the comparison output Is supplied as a control signal to the second switching means, the other end of the two windings and the second
A multiple output control power supply device comprising: a flywheel diode connected between the other ends of the following windings. 15. A primary winding, a first secondary winding, and a second two.
A converter transformer having a secondary winding, a first switching means for connecting and disconnecting the primary winding and its power source, and an output of the first secondary winding is rectified and smoothed and supplied to a first output end. Rectifying / smoothing means for performing PWM control of the first switching means according to the voltage of the first output end.
M control means, rectifying means connected to one end of the second secondary winding, two-winding choke coil for supplying the output of the rectifying means to the second output end, and two-winding choke Second switching means for connecting and disconnecting a common connection point between the coil and the rectifying means and the other end of the second secondary winding;
The flywheel diode connected between the end of the winding opposite to the rectifying means connection side and the other end of the second secondary winding, and the voltage at the second output end are compared with a reference value, and the comparison output is compared. A multiple output control power supply device, comprising: a comparison means for supplying a control signal to the second switching means.