JPH0568955B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to a drive circuit for a brushless DC motor that eliminates the need for a rotor position detection element.

[Background technology and its problems]

Conventionally, brushed DC motors have commutated current using a mechanical switch using brushes and a commutator, and the spark discharge noise generated by the intermittent switching of this mechanical switch can adversely affect the surrounding electronic circuits. It was hot. Moreover, this repetition of spark discharge causes wear of the brushes and damage to the commutator, which shortens the life of the motor.

On the other hand, a brushless DC motor replaces the mechanical switch with a semiconductor switch such as a transistor, and eliminates the drawbacks of a brushed DC motor.

In this brushless DC motor, it is necessary to detect the position of the rotor, which is the field magnet, and sequentially switch the mode of energization of the armature coils that make up the stator. For example, a Hall element or the like is used. Generally, it is necessary to provide the same number of Hall elements as the number of phases of the armature coil, which are arranged within the magnetic field of the rotor.

By the way, using a position detection element such as a Hall element in a brushless DC motor in this way,
This has the disadvantage of increasing costs and increasing the number of assembly and wiring steps. Further, by disposing the position detection element, the coil volume of the armature may be limited, and the motor cannot be miniaturized.

Therefore, attempts have been made to reduce or eliminate the number of position detection elements such as Hall elements arranged to detect the position of the rotor. For example, in a three-phase brushless DC motor, two position detection elements are used and the third phase is synthesized by the sum of the outputs of these detection elements. Furthermore, a sensorless method for three-phase unidirectional energization is known in which the back electromotive force induced in two resting coils is detected and the next energization is determined.

However, unidirectional energization has the disadvantage that the output ratio to the external shape of the motor is small and the torque ripple is large. Furthermore, the above-described control using two position detection elements has drawbacks such as the need to match the sensitivities of the two detection elements and restrictions on the shape of the detected magnetic flux.

[Problem that the invention seeks to solve]

As described above, conventional brushless DC motors have problems such as the inability to downsize the motor, increased costs, and increased man-hours for assembly and wiring due to the use of rotor position detection elements. . In addition, a brushless DC motor without a position detection element is limited to one-way energization, and has a problem in that the output ratio relative to the size of the motor shape is small.

Therefore, the present invention was proposed to solve these problems. Even in bidirectional energization, there is no need for a rotor position detection element, and the number of assembly and wiring steps can be reduced, making it effective. To provide a drive circuit for a brushless DC motor, which does not limit the coil volume of an armature due to improved space, has a large output ratio to the external shape size, can be made ultra-miniaturized, and can reduce costs. With the goal.

Furthermore, the present inventor previously proposed in Japanese Patent Application No. 186372/1987 a drive circuit for a brushless DC motor that performs control based on an ultra-small phase shift in the current flowing through an armature coil. The present invention seeks to provide a novel drive circuit that does not rely on current control.

[Means for solving problems]

In order to achieve this object, the brushless DC motor drive circuit of the present invention counts the oscillation output of the oscillation circuit with a counter, and generates an energization mode switching signal for switching energization to each phase from the counter output. The maximum value of the drive voltage applied to the armature coil of each phase is detected, and the oscillation circuit It is characterized by controlling the phase of the oscillation output.

[Effect]

Therefore, according to the present invention, there is no need for a detection element to detect the position of the rotor even in bidirectional energization, and the number of assembly and wiring steps can be reduced. It becomes possible to provide a drive circuit for a brushless DC motor that is free from volume limitations, has a large output ratio to external shape size, can be ultra-miniaturized, and can reduce costs.

〔Example〕

Hereinafter, embodiments of the present invention will be described in detail based on the drawings.

First, the basic idea of the present invention will be described. In a brushless DC motor, electricity is supplied to the armature coils of each phase forming the stator by switching switching transistors and the like.
If the switching phase is correct and the current is constant,
Due to the induced voltage (back electromotive force) induced in the armature coil by the rotation of the rotor, which is the field magnet,
The drive voltage applied to the armature coil exhibits the highest voltage value at the center between the switching points. Further, if there is an advance or a lag in the phase of the switching, a shift will occur in the phase of the maximum value of the voltage waveform of the drive voltage. Therefore, from the output of a counter that counts the output of the oscillator, an energization mode switching signal for switching the energization mode for each phase is generated, and the above voltage is applied to a drive circuit that supplies this energization mode switching signal to the switching transistor. By detecting the phase shift of the maximum value of the waveform and using this detection signal to control the phase of the oscillation output of the oscillator, the motor can be driven without providing the position detection element described above, and the switching The phase of the motor can be corrected correctly, and the maximum torque can always be obtained regardless of load fluctuations.

FIG. 1 shows a drive circuit for a brushless DC motor constructed according to the present invention.
The brushless DC motor shown in FIG. 1 is, for example, a three-phase Y-connected bidirectionally energized brushless DC motor, with armature coils 1, composing three phases,
It has 2 and 3. Furthermore, the switching transistors 4 and 5 that supply power to the armature coil 1 and switch the energization mode are connected to the other end of the armature coil 1, with the emitter of the transistor 4 connected to the collector of the transistor 5, and this connection point being Y-connected. It is connected to the. Furthermore, in the switching transistors 6 and 7 that supply power to the armature coil 2 and switch the energization mode, the emitter of the transistor 6 is connected to the collector of the transistor 7, and this connection point is connected to the other end of the armature coil 2. . Further, in the switching transistors 8 and 9 that supply power to the armature coil 3 and switch the energization mode, the emitter of the transistor 8 is connected to the collector of the transistor 9, and this connection point is connected to the other end of the armature coil 3. . In addition, the transistors 4, 6, 8
The collectors of the two are commonly connected, and a DC power source Is, which is a constant current source, is supplied to this connection point. moreover,
The emitters of the transistors 5, 7, and 9 are each grounded.

Further, a voltage controlled oscillation circuit VCO 10 is connected to a hexadecimal counter 11, and the oscillation output of the VCO 10 shown in FIG. 2E is counted by the counter 11 at the falling edge of a pulse. Further, the hexadecimal counter 11 is connected to a three-phase logic circuit 12, and in this logic circuit 12, an energization mode switching signal for switching the energization mode to each phase of the armature coils 1, 2, and 3 is sent to the counter 11. is generated based on the counter output of The output of this three-phase logic circuit 12 is connected to the bases of the switching transistors 4, 5, 6, 7, 8, and 9, and these transistors 4, 5, 6, 7, 8,
9 are sequentially supplied with the energization mode switching signals.
As a result, switching transistors 4, 5,
The on state and off state of coils 6, 7, 8, and 9 are sequentially switched, and the energization mode is switched in which the armature coils 1, 2, and 3 are sequentially energized in both directions. A driving force is generated and the rotor rotates.

Further, the drive voltage applied to the armature coils 1, 2, and 3 is detected from a supply point 13 of the DC power supply Is, which is a common connection point of the collectors of the switching transistors 4, 6, and 8. The detected driving voltage waveform is shown in FIG. 2B. Here the second
Figure A shows a switching pulse corresponding to the timing of switching, which is switching of the energized phase. The horizontal axis in Figure 2 shows the passage of time; in Figure 2, period T ₃ is a period when the switching phase is correct, period T ₂ is a period when this phase is slightly advanced, and period T ₁ is a period when this phase is correct. indicates a more advanced period. Further, period T ₄ indicates a period in which the switching phase is slightly delayed, and period T ₅ indicates a period in which this phase is further delayed. This figure 2B
As shown in the voltage waveform above, if the switching phase is correct, each armature coil 1,
The phase of the maximum value of the drive voltages applied to 2 and 3 is located at the center of the switching interval, that is, at the center between the switching pulses. Furthermore, the phase of the maximum value of the voltage is shifted in accordance with the phase shift of the switching.

By the way, the supply point 13 is connected to a differentiating circuit 14, and the voltage waveform is differentiated in this differentiating circuit 14. When the voltage waveform shown in FIG. 2B is differentiated by the differentiating circuit 14, a differential output shown in FIG. 2C is obtained. As shown in FIG. 2C, this differential output crosses zero at the local maximum point and switching point of the voltage waveform. Further, the output of the differentiating circuit 14 is connected to a non-inverting input terminal of a comparator circuit 15 whose inverting input terminal is grounded. Therefore, the comparison circuit 15
As shown in FIG. 2D, the comparison circuit 15 provides a comparison output (pulse waveform) having a pulse width between the local maximum point and the zero crossing point of the voltage. Further, the output of the comparator circuit 15 is supplied to one input terminal of a phase comparator circuit 16, and the oscillation output of the VCO 10 is supplied to the other input terminal of the phase comparator circuit 16.
Here, the VCO 10, the phase comparison circuit 16, and the low-pass filter LPF provided at the next stage are
17 is PLL (Phase Locked Loop)
It constitutes a circuit. By the way, in the phase comparator circuit 16, the comparison output of the comparator circuit 15 and the VCO1
By comparing the phase with the oscillation output of 0 at the rising edge of the pulse, the phase shift of the maximum value of the voltage waveform shown in FIG. 2B, that is, the phase shift of the switching is detected. This detected phase shift is output from the phase comparator circuit 16 as two outputs corresponding to the lead and lag of the phase shift. These two outputs are shown in Figures 2F and 2G. Figure 2F shows the output when the switching phase is advanced, and Figure 2G shows the output when the switching phase is delayed. ing. Furthermore, the output of the phase comparator circuit 16 is passed through the LPF 17 and converted into a DC voltage proportional to the phase shift. The output of this LPF 17 is shown in FIG. 2H. Further, the output voltage of the LPF 17 is added to the bias voltage V in an adder circuit 18, and then supplied to the VCO 10. Therefore, in the VCO 10, when there is a phase advance in the switching, the oscillation frequency temporarily decreases,
Further, when there is a delay in this phase, the oscillation frequency is temporarily increased, thereby controlling the phase of the oscillation output of the VCO 10. As a result, the phase of the maximum value of the voltage waveform shown in FIG. 2B is always controlled to be in the center between the switching pulses,
The above switching phase shift is corrected. In this way, the neutral point shift due to armature reaction is automatically corrected, so output at high loads can be improved, and the motor can always be driven with maximum torque regardless of load fluctuations. can. Furthermore, by correctly correcting the switching phase shift, so-called spike noise can be suppressed, and for example, the influence of noise that passes through the power supply line and affects peripheral electronic circuits can be reduced.

Here, the bias voltage V is supplied to the VCO 10 at the time of starting when the output voltage of the LPF 17 is zero, and the motor is started. Furthermore, at this startup, the bias voltage V is set so that the oscillation frequency of the VCO 10 becomes low. In addition, by gradually increasing the bias voltage,
It may be started by gradually increasing the oscillation frequency of the VCO 10.

By the way, in the above embodiment, the phase comparator circuit 1
Consists of 6, LPF17, and VCO10
The PLL circuit is used to control the phase of the oscillation output of VCO10.
Regardless of the circuit configuration, the comparison output of the comparison circuit 15 is passed through the low-pass filter LPF 19, and this LPF
19 may be integrated and then input to the VCO 10. FIG. 3 shows the essential parts of another embodiment constructed in this manner, and only the modified portions of the above-described embodiment of FIG. 1 are shown in a block diagram. In FIG. 3, the comparison output of the comparison circuit 15 is passed through a low pass filter LPF 19 to obtain an average value. The output of this LPF 19 is shown in FIG. 4F. In this FIG. 4, FIGS. 4A to 4E correspond to FIGS. 2A to 2E, and periods T ₁ to T ₅ shown in FIG.
This corresponds to periods T ₁ to T ₅ in the figure. by the way,
The output of the LPF 19 is supplied to an integrating circuit 20. This integrating circuit 20 is provided to eliminate frequency deviation and control the phase of the oscillation output of the VCO 10. The output of this integrating circuit 20 is supplied to the VCO 10 after passing through the adding circuit 18. As a result, LPF 19 for period T ₃ in Figure 4
, that is, the output of LPF19 when the above switching phase is correct, VCO1
The phase of the 0 oscillation output is controlled according to the switching phase shift, and the switching phase shift is correctly corrected.

Next, still another embodiment will be described with reference to the block diagram of FIG. 5 and the waveform diagram of FIG. 6. This FIG. 5 shows only the modifications of the embodiment of FIG. 1. In addition, in the waveform diagram of Fig. 6,
6 A to E correspond to FIG. 2 A to E, and the periods T ₁ to T ₅ shown in FIG. 6 are the periods of FIG. 2.
It corresponds to T ₁ to T ₅ .

In FIG. 5, the comparison output of the comparison circuit 15 is input to a sample pulse generation circuit 21. In FIG. This sample pulse generation circuit 21 generates a sample pulse signal shown in FIG. In this triangular wave generation circuit 22, a triangular wave signal shown in FIG. 6G synchronized with the sample pulse signal is generated,
This triangular wave signal is input to the peak hold circuit 23. This peak hold circuit 23 holds the peak value of the triangular wave signal based on the sample pulse signal.
The peak hold signal shown in Figure I is created.

Further, the comparison output of the comparison circuit 15 is input to the sample pulse generation circuit 24. This sample pulse generation circuit 24 generates a sample pulse signal shown in FIG. The above-mentioned triangular wave signal is also input to this sample-hold circuit 25, and in this sample-hold circuit 25, the above-mentioned triangular wave signal is sampled and held based on the input sample pulse signal.
FIG. 6 J of the output of this sample and hold circuit 25
The sample hold signal shown in is input to the adder circuit 26.

By the way, the peak hold signal that is the output of the peak hold circuit 23 is transmitted to the 1/2 multiplier circuit 2.
After being multiplied by 1/2 in step 7, the voltage is inverted and input to the adder circuit 26. Therefore, this adder circuit 26 calculates 1/2 from the sample hold signal.
The multiplied peak hold signal is now subtracted.

The output signal of this adder circuit 26 is shown in FIG. 6K, and this output signal is supplied to the VCO 10 via the above-mentioned adder circuit 18.

As described above, in this embodiment, digital differentiation and digital time counting are performed using means such as a sample hold and a counter. The switching phase shift from 2 is extracted as an error voltage signal. Here, if the switching phase is correct, the phase of the maximum value of the drive voltage becomes the center between the switching pulses (1/2 of the switching period) as described above.

Therefore, based on the output of the adder circuit 26 during period _T3 in FIG. 6, that is, the output of the adder circuit 26 when the switching phase is correct,
The phase of the oscillation output of the VCO 10 is controlled according to the switching phase shift, and the switching phase shift is correctly corrected.

By the way, the reason why 1/2 of the switching period is used as a reference is to obtain a reference voltage that is not related to changes in the rotational speed of the motor.

As described above, according to the present invention, the oscillation output of the VCO 10 is counted by the hexadecimal counter 11, and the 3-phase logic circuit 12 switches the energization mode of each phase based on the output of the hexadecimal counter 11. In the drive circuit that generates the signal,
The maximum value of the drive voltage applied to the armature coils 1, 2, and 3 is detected, and based on the phase shift between the maximum value of the detected voltage waveform and the phase of 1/2 of the switching period, the voltage of the VCO 10 is determined. Controls the phase of the oscillation output. Therefore, it is possible to drive the motor without disposing the above-mentioned rotor position detection element, and it is also possible to always correct the phase shift in switching for switching the energized phase to the correct position. As described above, the present invention eliminates the need for the position detection element, thereby reducing costs and reducing the number of assembly and wiring steps required for the position detection element. Moreover, the effective space inside the motor is improved, the coil volume of the armature is not limited, and the coil can be wound effectively. Furthermore, since no position detection element is required, the motor can be made ultra-small. In addition, no position detection element is required even in bidirectional energization.
The output can be improved relative to the size of the external shape of the motor, and torque ripple can be reduced. Furthermore, since the switching phase shift is always correctly corrected, maximum torque can always be generated regardless of load fluctuations, and noise can be prevented from occurring. Further, it can be easily added externally to an existing drive circuit.

Incidentally, in the above three embodiments, an example of a Y-connected brushless DC motor with 3-phase bidirectional energization is cited, but the number of phases is not limited to 3 and may be any number of phases, and it is also possible to use a unidirectional energized motor. It's okay. Further, the present invention can also be applied to a brushless DC motor with Δ (delta) connection.

〔Effect of the invention〕

As is clear from the above description, according to the present invention, in a drive circuit that counts the output of an oscillation circuit with a counter and generates an energization mode switching signal for switching the energized phase based on the counter output, the armature coil of each phase is The maximum value of the applied driving voltage is detected, and the phase of the oscillation output of the oscillation circuit is controlled based on the phase shift between the maximum value of the detected voltage waveform and the phase of 1/2 of the switching period. As a result, the switching phase for switching the energized phase can always be correctly corrected.

Therefore, the brushless DC motor can be driven without providing a rotor position detection element. Since this position detection element is not required,
Costs can be reduced, the number of man-hours for assembly and wiring are reduced, and the effective space inside the motor is improved, so that the coil volume of the armature is no longer limited. In addition, it becomes possible to make the motor extremely compact. Moreover, since a position detection element is not required even in bidirectional energization, the output ratio to the external shape of the motor can be improved and torque ripple can be reduced.

In addition, since neutral point shift due to armature reaction is automatically corrected, the motor can always be driven with the correct switching phase, and the motor can always be driven with maximum torque regardless of changes in motor load. At the same time, generation of noise can be prevented.

Further, it can be easily added externally to an existing drive circuit.

[Brief explanation of the drawing]

FIG. 1 is a block diagram of a drive circuit for a brushless DC motor according to the present invention, FIG. 2 is a waveform diagram explaining the operation of the drive circuit, and FIG. 3 is a drive circuit for a brushless DC motor showing another embodiment of the present invention. A block diagram showing the main parts of the circuit, FIG. 4 is a waveform diagram explaining the operation of the drive circuit of FIG. 3, and FIG. 5 is a main part of the drive circuit of a brushless DC motor showing still another embodiment of the present invention. FIG. 6 is a waveform diagram illustrating the operation of the drive circuit shown in FIG. 5. 1, 2, 3...armature coil, 4, 5, 6,
7, 8, 9...Switching transistor, 10
...VCO, 11...Hex counter, 12...3
Phase logic circuit, 14...Differential circuit, 15...Comparison circuit, 16...Phase comparison circuit, 17...LPF,
18...Addition circuit, 19...LPF, 20...Integrator circuit, 21, 24...Sample pulse generation circuit, 22...Triangle wave generation circuit, 23...Peak hold circuit, 25...Sample hold circuit, 2
7...1/2 multiplier circuit, 26...addition circuit.

Claims (1)

[Scope of Claims] 1. In a drive circuit that counts the oscillation output of an oscillation circuit by a counter and generates an energization mode switching signal for switching energization to each phase from the counter output, a drive applied to the armature coil of each phase. It is characterized by detecting the maximum value of the voltage and controlling the phase of the oscillation output of the oscillation circuit based on the phase shift between the phase of the detected maximum value of the voltage waveform and the phase of 1/2 of the switching period. A drive circuit for a brushless DC motor.