JPH05126636A - Infrared spectroscope - Google Patents

Abstract

PURPOSE:To provide an infrared spectroscope having very low noises with a simple structure. CONSTITUTION:An infrared spectroscope 10 making spectral diffraction on the infrared light generated by an infrared light source S, an infrared detector receiving the infrared light to generate the electric signal, and an AC amplifier AMP 1 connected to the infrared detector are provided. The amplifier AMP 1 has multiple feedback amplifying circuits connected in parallel, it has the first amplification section synthesizing the outputs of these amplifying circuits via resistors and the second amplification section connected to the first amplification section, and the impedance of the second amplification section is set to the parallel resistance value or above of the resistors of the first amplification section.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an infrared spectroscopic device, and more particularly to a highly sensitive infrared spectroscopic device suitable for time-resolved infrared spectroscopy.

[0002]

2. Description of the Related Art Time-resolved spectroscopy has been used effectively for observing spectra of short-lived molecular species.
Conventionally, it has been performed mainly using light in the ultraviolet to visible range. However, in recent years, an infrared light source with high brightness and an infrared light detector with high sensitivity and excellent time response have been developed, and therefore infrared light is often used. This is because time-resolved infrared spectroscopy has the following advantages.

(1) Vibration (rotation) obtained by infrared spectroscopy
The spectrum is composed of many bands, and is very sensitive to changes in the molecular structure, electronic structure, and existing state of the molecular species that give it. Therefore, it is possible to easily distinguish between molecular species that are difficult to distinguish by ultraviolet to visible spectroscopy, and it is possible to extract a wealth of information regarding changes in molecular structure, electronic structure, existing state, and the like.

(2) The dynamic process of generation / relaxation can be traced for each vibration (rotation) level.

(3) Unlike the absorption and emission spectra in the ultraviolet to visible region, almost all molecular species have infrared absorption intensities. Therefore, whether neutral molecules or ions, ground state or excited state,
Alternatively, it can be applied to almost all molecular species regardless of solute molecules or solvent molecules.

When observing the spectrum of molecular species by time-resolved infrared spectroscopy, a glow bar or a laser is usually used as an infrared light source, and an MCT (mercury-cadmium-tellurium) is used as an infrared light detector. ) Detector and I
An nSb (Indium-Antimony) detector is used. A suitable DC bias voltage is applied to the infrared light detector. The electrical signals obtained by these infrared photodetectors are
After being amplified by the amplifier, it is input to an electrical gate circuit, and the time-resolved signal is extracted there. A boxcar integrator, a digital oscilloscope (DSA) or the like is usually used as the electric gate circuit.

In this kind of electronic measurement, the signal from the detector is often weak, and it is required to amplify this signal with low noise and high stability. 9 and 10
Figure 1 shows a conventional amplifier used for this type of measurement.

FIG. 9 is a circuit diagram showing an example of the circuit configuration of a conventional amplifier used for this type of measurement. The conventional amplifier 51 is a two-stage amplifier including a first amplification section and a second amplification section.

The first amplification section is composed of n same feedback amplification circuits connected in parallel. A1 to An are operational amplifiers,
RF1 to RFn and RS1 to RSn are feedback resistors, and R1 to Rn are resistors for combining the outputs of the operational amplifiers A1 to An.
The input of each amplifier circuit is common, and the input signal is the operational amplifier A1.
~ An is input from its + terminal. Operational amplifier A1 to A
The-terminal of n is grounded via the feedback resistors RS1 to RSn. A part of the outputs of the operational amplifiers A1 to An is connected to the feedback resistor R
It is fed back to the-terminals of the operational amplifiers A1 to An via F1 to RFn. The outputs of the amplifier circuits are taken out via the resistors R1 to Rn, combined, and then sent to the second amplifier section.

The second amplifier section is composed of one feedback amplifier circuit having an operational amplifier Aa and a feedback resistor RFa. The output obtained by combining the outputs of the n amplifier circuits of the first amplifier section is input to the operational amplifier Aa from its negative terminal. The + terminal of the operational amplifier Aa is grounded. Operational amplifier Aa
A part of the output of the operational amplifier Aa is fed back via the feedback resistor RFa.
Is fed back to the-terminal.

In the conventional amplifier 51, when a signal is input from the input terminal, the signal is input to the n feedback amplification circuits of the first amplification section and amplified. The outputs of the amplifier circuits are taken out through the resistors R1 to Rn, respectively, combined, and input to the operational amplifier Aa of the second amplifier section. The signal input to the operational amplifier Aa is amplified and sent to the output terminal.

In the first amplifying section, the signal component of the combined output of each amplifying circuit has the same magnitude as that of each amplifying circuit alone, but the noise component contained in the combined output is the noise of each amplifying circuit. It becomes 1 / n ^1/2 . Therefore, in the above conventional amplifier 51, the noise of the first amplifying section is reduced to 1 / n ^1/2 and the S / N is improved.

FIG. 10 is a circuit diagram showing an example of the circuit configuration of another conventional amplifier used for this type of measurement.

The conventional amplifier 61 shown in FIG. 10 has a two-stage structure using two transistors Q1 and Q2 whose emitters are grounded. The signal is input to the base of the transistor Q1 of the first stage via the capacitor C1, and the output is taken out from the collector of the transistor Q2 of the second stage via the capacitor C2. In order to increase the stability of the operation, the first resistor from the emitter of the second-stage transistor Q2 is fed through the feedback resistor RF.
Negative feedback is applied to the base of the transistor Q1 of the stage.
RC is the collector resistance of the transistor Q1, RE is the emitter resistance of the transistor Q2, and R1 is the collector resistance of the transistor Q2.

Generally, in order to match the input impedance of the amplifier with the signal source resistance (for example, 50Ω), a matching resistor having a resistance value equal to the signal source resistance may be connected in parallel to the input terminal of the amplifier. However, this increases the noise because the matching resistor becomes a new noise source.
Therefore, it is desired to match the input impedance without using the matching resistor.

The conventional amplifier 61 equivalently realizes the matching of the input impedance by the negative feedback without using the matching resistor as the noise source. By doing so, it is not necessary to connect a resistor to the input terminal, and it is possible to perform amplification with low noise.

The collector resistance RC is, for example, 2.2 k.
Ω, the emitter resistance RE is 200Ω, the collector resistance R1 is 360Ω, and the feedback resistance RF is 2.2k, for example.
Ω, the capacitor C1 is, for example, 2000 pF, and the capacitor C2 is, for example, 10,000 pF.

[0018]

The conventional amplifier 51 described above is used.
In the above, the feedback resistances RF1 to RFn and RS1 to R directly affect the noise reduction of each amplifier circuit of the first amplifier.
It is Sn, and it is preferable to set their resistance values as small as possible. However, in the above conventional amplifier 51, the negative terminal of the second amplifying section is regarded as virtual ground, so that the operational amplifiers A1 to An have feedback resistors RF1 to RFn and RS1 to RSn.
Not only this, it is necessary to drive the resistors R1 to Rn. Therefore, it is difficult to set the resistance values of the feedback resistors RF1 to RFn and RS1 to RSn to be low, and there is a problem in that noise reduction is limited.

On the other hand, in the above conventional amplifier 61, the gain and the input impedance fluctuate when the ambient temperature fluctuates due to the temperature dependence of the transistors Q1 and Q2, which makes it difficult to perform highly stable amplification. is there.

Therefore, the conventional amplifiers 51 and 61 described above are used.
Even if infrared spectroscopy is carried out using, the detection sensitivity is insufficient, so that there is a problem that, for example, a time-resolved infrared spectrum or a temporal change in infrared absorption intensity cannot be obtained with a good S / N ratio.

Therefore, an object of the present invention is to provide an infrared spectroscope having a simple structure and extremely low noise.

Another object of the present invention is to provide an infrared spectroscope whose operation is extremely stable against changes in ambient temperature.

[0023]

A first infrared spectroscopic device of the present invention is an infrared light source, an infrared spectroscopic means for separating infrared light generated by the infrared light source, and the infrared light source. Infrared light detecting means for receiving the infrared light generated by the infrared light detecting means and generating an electric signal in accordance therewith; and amplification for amplifying the electric signal coupled to the infrared light detecting means and generated by the infrared light detecting means. A first amplifying unit that includes a plurality of feedback amplifying circuits connected in parallel with each other, and that combines the outputs of the amplifying circuits via resistors.
A second amplifying section coupled to the first amplifying section, and the input impedance of the second amplifying section is equal to or more than the parallel resistance value of the resistance of the first amplifying section.

A second infrared spectroscopic device of the present invention is an infrared light source, an infrared spectroscopic means for separating infrared light generated by the infrared light source, and an infrared light generated by the infrared light source. Infrared light detection means for receiving light and generating an electric signal corresponding thereto, and amplification means coupled to the infrared light detection means and amplifying the electric signal generated by the infrared light detection means The amplifying means realizes the matching of the input impedance with the resistance of the signal source by the negative feedback, and controls the voltage across the load to be equal to the reference voltage, and the reference voltage depends on the ambient temperature. And a negative voltage feedback means for changing the gain of the amplifying means so as to cancel the gain change, negative AC feedback means for negative AC feedback, and negative DC feedback means for negative DC feedback. Means and DC negative feedback There characterized in that it has a temperature coefficient for canceling the change in the gain of the amplifying means changes according to ambient temperature.

[0025]

In the first infrared spectroscopic device of the present invention, the first amplifying section of the amplifying means has a plurality of feedback amplifying circuits connected in parallel with each other, and combines the outputs of these amplifying circuits via resistors. Because of the configuration, the noise component included in the combined output is smaller than that of each feedback amplifier circuit.

Moreover, since the input impedance of the second amplifying section looks as many times as the number of the first amplifying sections connected in parallel,
The current flowing through the second amplification unit through the resistance is small, and therefore, the resistance can be ignored for the signal component, so that the resistance value of the feedback resistance of each feedback amplification circuit of the first amplification unit is low. Can be set. Therefore, the noise generated by each feedback amplifier circuit is further reduced.

Therefore, it is possible to further improve the low noise amplification.

In the second infrared spectroscopic device of the present invention, since the amplifying means realizes the matching of the input impedance with the signal source resistance by negative feedback, this occurs as compared with the case of connecting the impedance matching resistance. There is little noise.

Moreover, when the ambient temperature changes, the reference voltage of the voltage control means changes correspondingly so as to cancel the change of the gain of the amplification means, and the AC negative feedback means and the DC negative feedback means operate as the amplification means. Since it changes so as to cancel the change in gain, it is very stable against changes in ambient temperature.

[0030]

Embodiments of the present invention will be described below with reference to the accompanying drawings.

(Overall Structure) FIG. 1 is a functional block diagram showing the system structure when performing time-resolved infrared spectroscopy using an embodiment of the infrared spectroscopy apparatus of the present invention.

The infrared spectroscopic device of the present invention is a dispersion type infrared spectroscope 10 having an infrared light source S and an optical system for guiding infrared light, and an infrared spectroscope incorporated in the infrared spectroscope 10 and used for infrared detection. Bowl D
And an amplifier AMP1 for amplifying an electric signal emitted from the infrared detector D. The infrared detector unit 1 has a bias power supply VB for supplying a bias to the infrared detector D.

Around the infrared spectroscopic device of the present invention, a pulse generator 11, a laser device 12 which is a light source for exciting the sample SAM, and an optical system for guiding the laser light emitted from the laser device 12 to the sample SAM, An amplifier AMP2 for further amplifying the electric signal amplified by the amplifier AMP1,
A boxcar integrator 16 and a digital oscilloscope (DSA) 17 for extracting a time-resolved signal from an electric signal amplified by the amplifier AMP2, and a computer 18 for processing the data obtained by the boxcar integrator 16 and the digital oscilloscope 17 are provided. It is provided.

(Infrared spectroscope) The infrared spectroscope 10 introduces the infrared light emitted from the infrared light source S into the infrared photodetector unit 1 by using a plurality of mirrors, and is provided there. The infrared light detector D is irradiated. The infrared light optical path from the infrared light source S to the infrared light detector D is as follows.

The infrared light emitted from the infrared light source S is first reflected by the convex mirror M2. Next, the light is reflected by the plane mirror M3 and is irradiated onto the sample SAM through the optical diaphragm AP2. The infrared light transmitted through the sample SAM is reflected by the plane mirrors M4, M.
5, M6, M7, and M8 are sequentially reflected, and after passing through the optical slit SL1, they are reflected by the concave mirror M9 and enter the diffraction grating G. The infrared light incident on the diffraction grating G is reflected by the diffraction grating G, and the wavelength (wave number) is selected. The infrared light whose wavelength has been selected is reflected again by the concave mirror 9 and passes through the optical slit SL2. Then, after passing through the optical filter F for removing high-order light, reflected by the plane mirror M10, further reflected by the concave mirror M11, the infrared light detector D is irradiated.

The usable wave number range of the infrared spectroscope 10 is about 40.
It is preferably from 00 to 650 cm ^-1 .

A chopper CH is provided between the plane mirror M3 and the diaphragm AP2 so that it can be inserted into and removed from the optical path. This is used to measure the absolute value of the intensity of the infrared light by performing the measurement with modulation of about 2 to 3 kHz by the chopper CH. That is, in the infrared spectrometer of the present invention,
This is because when the amplifier AMP1 has the circuit configuration of FIG. 6, AC amplification is performed, and thus the infrared light intensity cannot be directly obtained. When measuring the change in infrared light intensity, chopper C
H is taken out of the optical path.

The not-shown cam for driving the diffraction grating, the interference filter for removing higher-order light, the slit, the shutter, the sector mirror, the diffraction grating switching mirror, etc., inside the infrared spectroscope 10 are
It is driven by a stepping motor controlled by a one-board computer (not shown) mounted inside the spectrophotometer. Data communication between this one-board computer and the external computer 18 for data processing is
Performed by serial line (RS-232C).

As the infrared light source S, a glow bar used as a standard in a spectrophotometer is used. However, it is preferable that a ceramic light-emitting body (a platinum resistance wire covered with a rare earth oxide) is provided together with the glow bar, and both are switched and used according to the operating frequency.

Since the laser light emitted from the laser device 12 is condensed on the sample SAM to some extent, as will be described later, a pair of sample SAMs is formed in order to collect the monitor infrared light to the same extent. A beam condenser (not shown) having a structure sandwiched between off-axis ellipsoidal mirrors is provided.
With this beam condenser, the size of the infrared light beam at the sample SAM position is about 1 mm in diameter.

Although the dispersion type infrared spectroscope 10 is used here, an infrared spectroscope other than the dispersion type can also be used.

The pulse generator 11 synchronizes the oscillation of the laser device 12 which is the excitation light source of the sample SAM, the modulation of the chopper CH, and the trigger of the boxcar integrator 16 and the digital oscilloscope 17,
It works to operate them in synchronization.

The laser device 12 is a light source for photoexcitation of the sample, and two kinds of laser devices having a time width of nanosecond and picosecond are switched and used.

The nanosecond laser device is a CW Q-switched Nd: YAG laser whose output is converted into a double wave by a KTP crystal (wavelength 532 nm, repetition frequency 50).
Hz to 2 kHz, energy 300 μJ / pulse, pulse width 100 ns).

The picosecond laser device uses the output of the CW mode-locked Nd: YAG laser as the seed light, and outputs the output of the CW regenerative amplifier with the BBO crystal.
Converted to double wave (wavelength 532nm or 266n
m, repetition frequency 2 kHz, energy 200-40
0 μJ / pulse, pulse width 65 ns or less).

The laser light emitted from the laser device 12 is first introduced into the prism P through the convex lens L1, the second harmonic generator SHG and the convex lens L2. Therefore, after the wavelength is selected, it is reflected by the plane mirror M1 through the optical aperture AP1 and is incident on the sample SAM. Since the laser light is condensed on the sample SAM by the lenses L1 and L2, the photoexcitation density of the sample SAM can be increased.

With the laser device 12 having the above-described structure, it is possible to improve the S / N ratio of the measurement by performing the repeated averaging while obtaining the excitation speed sufficient for measuring the transient phenomenon.

(Infrared Light Detector Unit) FIG. 2 is an explanatory view showing an embodiment of the infrared light detector unit of the infrared spectroscopic apparatus of the present invention and its peripheral equipment, and FIG. 3 is its circuit diagram and FIG. 4 is a perspective view of a head to which an infrared light detector is attached.

As shown in FIG. 2, the infrared light detector unit 1 includes an infrared light detector D, a cryogenic container (Dewar bottle) 2 accommodating the infrared light detector D, and a detector D. And a lead wire LD for taking out an electric signal to the outside of the cryocontainer 2. A bias power supply VB for applying a bias voltage to the detector D and an amplifier AMP1 for amplifying an electric signal generated by the detector D are connected to the lead wires LD outside the cryogenic container 2.

The infrared photodetector D is composed of a photoconductive MCT crystal, and when it is irradiated with infrared light, it generates an electric signal corresponding to its intensity. In this embodiment, as shown in FIG. 4, the detector D is attached to a metal head 5 provided inside a cryogenic container (Dewar bottle) 2. Infrared light is introduced through a window 4 provided in the infrared light introducing section 3 of the cryogenic container 2 and is irradiated on the detector D. The head 5 is adjacent to the chamber 2a,
The liquid nitrogen contained in the chamber 2a is cooled to 77 ° K. By cooling, the thermal noise generated by the MCT crystal can be reduced. The detector D is cooled by liquid nitrogen via the head 5. The electric signal generated by the detector D is taken out of the container 2 via the lead wire LD.

A current is supplied to the detector D from the bias power source VB through the bias resistor RB (see FIG. 3). When the detector D is irradiated with infrared light, the resistance RD of the detector D slightly decreases in proportion to the light intensity, and as a result, the potential of the connection point A between the detector D and the bias resistor RB changes. To do. Therefore, if this potential change is taken out as a signal, the absolute value of the intensity of the irradiated infrared light can be measured.

In the detector D, when the composition ratio of the MCT crystal used is changed to extend the cutoff frequency to the low wave number side, the specific detection capability D ^* is lowered (S / N ratio is lowered). Here, considering the balance between the cutoff wave number and the specific detection capability D ^* , the sensitivity is 800 to 700 cm ^-1 , and Sp
ectral D ^* (peak, 10kHz, 1Hz) is 3-5 × 1
It is assumed to be about 0 ¹⁰ . The rise of the response to the pulsed light is 20 ns, and the decay of the response is about 1 μs.

Here, a photoconductive MCT crystal is used as the infrared light detector D, but a photovoltaic MCT crystal may be used as long as it detects infrared light and generates an electric signal. For example, other known infrared light detectors can be used.

Generally, when the signal intensity generated by the photoconductive detector is SMCT, the noise density is NMCT, and the input conversion noise density of the amplifier is Namp, the total input conversion noise density Nt after amplification is Nt.
Since otal is the root mean square of the noise density NMCT of the photoconductive detector and the input equivalent noise density Namp of the amplifier, the S / N ratio Rtotal in consideration of the amplification system is Rtotal = SMCT / Ntotal = SMCT / (NMCT ² + Namp ² ) ^1/2 .

Since the signal intensity SMCT and the noise density NMCT of the photoconductive detector are both substantially proportional to the bias current Ib, the bias current Ib can be ignored in order to neglect the influence of the input conversion noise density Namp of the amplifier. Is selected as large as possible, and the noise density NMCT is the input equivalent noise density Na of the amplifier.
It must be at least 3 times the mp.

The input-equivalent noise density Namp of the conventional wideband low-noise preamplifier of 1 MHz or more is 2 to 2 even if it is low.
It is usually about 4 nV (Hz) ^1/2 . In order to make this negligible, the bias current Ib needs to be considerably large. For example, the bias current Ib needs to be about 6 to 12 times larger than the recommended value.

However, if the bias current Ib is increased in this way, there is a possibility that the thermal breakdown strength of the MCT crystal may be exceeded. Also, the signal strength SMCT becomes too large, and when measuring the monitor strength using a chopper,
There is a great risk of exceeding the output clip level of the amplifier. Therefore, to increase the dynamic range of the measurement, it is necessary to use an even lower noise amplifier.

(Bias Power Supply) The bias power supply VB is connected to the detector D via a lead wire LD and supplies a predetermined DC voltage to the detector D. As the bias power source VB, it is preferable to use one having as low noise as possible.

(Amplifier) The amplifier AMP1 can have any of the circuit configurations shown in FIGS. However, both circuit configurations may be incorporated into one amplifier, and in that case, it is preferable to switch between the two in accordance with the frequency of the signal to be amplified.

(Amplifier) FIG. 6 is a circuit diagram showing an example of the circuit configuration of the amplifier AMP1. This amplifier AMP1
Is connected to the infrared photodetector D via the capacitor C1, so that the DC signal and the low frequency signal are not amplified. High frequency signal (for example, 10 kHz to 100 MHz)
It is suitable for amplification of.

This amplifier AMP1 has a grounded emitter 2
It has a two-stage configuration using individual transistors Q1 and Q2. Further, it has an error amplifier Q3 and an inverting amplifier Q4 for stabilizing the bias voltage supplied to the transistors Q1 and Q2.

Vcc is the bias power supply for the amplifier AMP1, V
REF is a reference power source for generating the reference voltage Vstd of the error amplifier Q3, RC is a collector resistance of the transistor Q1, and RE is an emitter resistance of the transistor Q2.

The input signal is fed through the capacitor C1 to the first
It is input to the base of the transistor Q1 of the stage. The output signal is taken out from the emitter of the second stage transistor Q2. The operation of the transistors Q1 and Q2 is the same as that of the conventional amplifier 61 of FIG. This amplifier AMP1
However, unlike the conventional amplifier 61, negative feedback is applied to DC and AC separately. That is, AC negative feedback is applied to the base of the transistor Q1 of the first stage from the emitter of the transistor Q2 of the second stage via the AC feedback resistor RFac and the capacitor C4. Furthermore, the error amplifier Q3
Is sent to the base of the transistor Q1 of the first stage through the inverting amplifier Q4, the DC feedback resistor RFdc and the resistor R3, so that the DC negative feedback is applied. The DC feedback circuit is grounded between the DC feedback resistor RFdc and the resistor R3 via the resistor R4 and the capacitor C5. The minus terminal of the error amplifier Q3 is connected to the collector of the transistor Q1 via the resistor R2, and the plus terminal thereof is the reference power source VRE.
It is connected to one end of the collector resistance RC of the transistor Q1 via F. A part of the output of the error amplifier Q3 is fed back to the-terminal via the capacitor C3.

The error amplifier Q3 is a differential amplifier, and when the voltage across the collector resistor RC fluctuates and an error occurs between the error and the reference voltage Vstd given by the reference power supply VREF, the error amplifier Q3 is amplified to amplify it. Inject the correction signal into the base of Q1. In this way, the error amplifier Q3 detects the voltage across the collector resistance RC of the transistor Q1 and operates so that the voltage is always equal to the reference voltage Vstd.
As a result, assuming that the current flowing through the collector resistance RC becomes constant and the base current flowing into the transistor Q2 can be ignored, the current flowing through the transistor Q1 becomes constant. At this time, if the bias power supply Vcc is constant, the collector voltage of the transistor Q1 is also constant, and the gain of the first-stage amplifier is stable. The transistor Q2 is an emitter follower having a voltage of about 1 time and its gain is basically stable.

The reference power supply VREF is designed so that the reference voltage Vstd changes in response to changes in ambient temperature, thereby canceling changes in the gain of the transistor Q1 caused by changes in ambient temperature. The reference power supply VREF having such a function can be realized by using a resistor whose resistance value changes at a desired rate of change in proportion to the ambient temperature. Alternatively, it can be realized by using a known IC capable of taking out a voltage or current proportional to the ambient temperature.

Since the inverting amplifier Q4 has the same polarity,
It is connected to the output of the error amplifier Q3.

Error amplifier Q3, inverter Q4, resistor R
2, R3, R4, DC feedback resistor RFdc, reference power supply VREF, and capacitors C3, C5 constitute voltage control means.

Since the amplification factor of the amplifier AMP1 of FIG. 6 is determined by the physical constants of the transistors, it is basically constant regardless of variations and types of the transistors Q1 and Q2. Therefore, the reproducibility is excellent, the temperature can be corrected well, and the gain stability of, for example, 100 to 200 ppm / ° C is not difficult. Further, since the bias voltage and the bias current are kept constant and the negative feedback is applied, the amplifying action of the transistors Q1 and Q2 is also stabilized. Furthermore, since the matching of the input impedance is equivalently realized by the negative feedback without using the matching resistor as the noise source, the noise is low.

Further, in this amplifier AMP1, since the negative feedback circuit is divided into the AC feedback circuit and the DC feedback circuit, the AC feedback amount and the DC feedback amount can be set independently, and as a result, the operating point can be easily set. This increases the degree of freedom in design. For example, it is possible to set the DC negative feedback at the first stage at the point of minimum noise, and further set the AC negative feedback so that the input impedance is matched.

Next, the conditions under which the gain and the input impedance of the transistors Q1 and Q2 are kept constant even if the ambient temperature changes will be described.

Considering one transistor, the voltage gain Gv in the case of grounded emitter is expressed as follows.

First, assuming that the feedback resistance of the transistor is Rc, the emitter current ie is: ie = VREF / Rc Further, the transconductance gm is gm = q.ie / kT where q is the electron charge and k is the Boltzmann. The constant T is an absolute temperature.

Therefore, the voltage gain Gv of the transistor is
Gv = gm.Rc = (q / kT) .VREF ........ (1)

From the equation (1), it can be seen that the voltage gain Gv changes in inverse proportion to the temperature T. Therefore, the reference voltage V
If REF is changed in proportion to the temperature T, the voltage gain Gv does not change with the temperature T.

That is, if the reference voltage VREF is changed so that (VREF / T) = α (constant), Gv = (q / k)  (VREF / T) = (q / k) α ... (2) The voltage gain Gv can be kept constant even if the ambient temperature changes.

As will be described below, the input impedance Zi of the transistor also changes with changes in ambient temperature.

The input impedance Zi of the grounded-emitter transistor Q is Zi = rb + (hfe / gm) = rb + (qαhfe / kRc) where rb is the base resistance of the transistor and hfe is the current. The amplification factor, Rc, is a feedback resistor.

In a high frequency transistor, rb is usually several Ω
Since it can be ignored, Zi ≈ (q · α / k · Rc) · hfe (3).

The current amplification factor hfe is usually +7000 ppm
Since the temperature coefficient is about / ° C, it can be seen that the input impedance Zi changes with the ambient temperature.

Therefore, in this amplifier AMP, if a temperature coefficient is given to the resistance value of the AC feedback resistor RFac so as to satisfy the relation of the above expression (3), the temperature change of the input impedance of the amplifier AMP1 can be compensated. As a result, it is possible to keep the input impedance and the voltage gain of the amplifier AMP1 constant over a wide temperature range.

For example, in an amplifier having a voltage gain of 40 dB and an input impedance of 50Ω, according to a simulation, if a temperature coefficient of −800 to −850 ppm / ° C is given to the feedback resistor, the change of the input impedance due to the temperature change can be canceled. The input impedance can be kept constant over a wide temperature range.

The collector resistance RC is, for example, 750.
Ω, emitter resistance RE is, for example, 1 kΩ, AC feedback resistance RF
ac is, for example, 11.5 kΩ, DC feedback resistor RFdc is, for example, 20 kΩ, resistor R2 is, for example, 1 MΩ, and resistor R3 is, for example, 1
kΩ, the resistance R4 is 1 kΩ, the capacitor C1 is 5 μF, the capacitor C3 is 0.33 μF, the capacitor C4 is 0.33 μF, and the capacitor C5 is 0.33 μF.

The voltage amplifying means does not have to have the above-described structure, but may have another structure as long as it operates in the same manner as described above. Further, as the amplifying element, transistors Q1 and Q2 are used.
Other known amplifying elements (for example, FET and HEMT) can be used.

Although the amplifier AMP1 has a circuit configuration of two-stage amplification here, it may be one-stage amplification or three or more stages.

As described above, the amplifier AM having the circuit configuration shown in FIG.
The reason for using P1 is as follows.

In the infrared photodetector D (MCT crystal),
When irradiated with infrared light, its resistance R is proportional to its light intensity.
D is slightly reduced. Therefore, when a current is supplied to the infrared light detector D by the bias power supply VB through the bias resistor RB, the potential at the connection point A between the infrared light detector D and the bias resistor RB changes, and this is used as a signal. Take out and measure the absolute value of infrared light intensity. The frequency characteristic of this signal is almost flat from the DC region to about MHz,
If this is directly amplified, the S / N will decrease. The reason is as follows.

(1) In the DC region, there is an offset corresponding to the bias voltage applied to the MCT crystal, but the intensity of the signal proportional to the infrared light intensity is considerably smaller than that.

(2) Since the intensity of the signal obtained from the transient change of infrared absorption is smaller than that of the infrared light intensity, it is smaller than the signal proportional to the infrared light intensity. There will be fluctuations.

(3) The 1 / f noise generated by the MCT crystal itself rises at 400 Hz or less.

Considering these points, this amplifier AMP1
Then, the coupling capacitor C1 removes the DC region and the low frequency region of the signal to perform AC amplification. An active filter may be inserted after the amplifier AMP2 for the same purpose.

By the way, if AC amplification is performed, the infrared light intensity I for monitoring cannot be obtained directly. Sample SAM
Change in infrared light intensity due to transient change in absorbance ΔA of ΔI
Is proportional to the infrared light intensity I, the infrared light intensity I can be calculated from the change A in the monitor light intensity ΔI to obtain the absorbance A of the sample.
It is necessary to obtain the absolute value of. Therefore, a chopper CH is placed in the optical path and a measurement with a modulation of several kHz is separately performed to obtain this.

(Amplifier) FIG. 5 is a circuit diagram showing another example of the circuit configuration of the amplifier AMP1. This amplifier AMP
Since No. 1 is directly connected to the detector D without using the coupling capacitor C1, it is suitable for amplifying a DC signal and a low frequency signal (for example, DC to 200 kHz) with low noise. Here, a two-stage configuration including a first amplification section and a second amplification section is provided.

The first amplification section is composed of four identical feedback amplification circuits connected in parallel. These feedback amplification circuits have the same configuration as each feedback amplification circuit of the first amplification section of the conventional amplifier 51 of FIG. That is, each feedback amplifier circuit
Inputs are common and all input signals are operational amplifiers A1 to A4
Is input to the + terminal. Of operational amplifiers A1 to A4
The terminals are grounded via feedback resistors RS1 to RS4, respectively. Some of the outputs of the operational amplifiers A1 to A4n are fed back to the operational amplifiers A1 to A4 via feedback resistors RF1 to RF4, respectively.
It is returned to the terminal. On the output side of the operational amplifiers A1 to A4,
Resistors R1 to R4 are connected to each other, and their outputs are taken out via the resistors R1 to R4, combined, and then sent to the second amplification section.

The operational amplifiers A1 to A4 are all of the same type, and the feedback resistors RF1 to RF4 have the same resistance value. The feedback resistors RS1 to RS4 have the same resistance value, and the output resistors R1 to R4 also have the same resistance value.

The second amplifier section is composed of one feedback amplifier circuit having an operational amplifier Aa and a feedback resistor RFa. The output obtained by combining the outputs of the four feedback amplifier circuits of the first amplifier section is input to the operational amplifier Aa from its + terminal. The-terminal of the operational amplifier Aa is grounded via the feedback resistor RSa. A part of the output of the operational amplifier Aa is a feedback resistor RF.
It is fed back to the-terminal of the operational amplifier Aa via a. The output of the operational amplifier Aa is sent from the output terminal.

The input impedance of the operational amplifier Aa of the second amplifying section may be at least the parallel resistance value of the resistors R1 to R4. This is because if it is smaller than the parallel resistance value, a sufficient noise reduction effect cannot be obtained.

The input impedance of the second amplification section is preferably 10 times or more, and more preferably 100 times or more the parallel resistance value of the resistors R1 to R4.

In this amplifier AMP1, since the input impedance of the second amplification section is set to 100 times or more the parallel resistance value of the resistances R1 to R4 of the first amplification section, the resistances R1 to R4 are set.
The current flowing through the second amplification unit is very small. Therefore, the resistors R1 to R4 can be ignored for the signal component, and each amplifier circuit of the first amplifier section does not need to drive the resistors R1 to R4.

Therefore, according to the resistance values of the resistors R1 to R4,
Since the resistance values of the feedback resistors RS1 to RS4 and RF1 to RF4 of the first amplification unit can be set low, the combined resistance value of the feedback resistors RS1 to RS4 and RF1 to RF4 becomes low, and the feedback amplification circuits of the first amplification unit are reduced. The generated noise is reduced as compared with the conventional one.

In this amplifier AMP1, the noise of each feedback amplifying circuit of the first amplifying section reduced as described above becomes 1/4 ^1/2 = ^1/2, so that the noise of the entire first amplifying section is increased. Is significantly reduced. As a result, much better low noise amplification than before is possible. The gain of the first amplification unit is the same as the gain of each amplification circuit of the first amplification unit.

In this embodiment, the four feedback amplification circuits of the first amplification section have exactly the same configuration, but if the feedback amplification circuits have the same gain, some components may be different. The same component may have different numerical values or the like. Further, as long as it is a feedback amplification circuit, it may be another type of amplification circuit that does not use the operational amplifiers A1 to A4.

In order to confirm the effect of the amplifier AMP1 having the circuit configuration shown in FIG. 5, this amplifier AMP1 and the conventional amplifier 51 are used.
Was manufactured and tested under the following test conditions. Amplifier A
The MP1 has the circuit configuration shown in FIG. 5, and the conventional amplifier 51 has the circuit configuration shown in FIG.

(Test conditions) Amplifier AMP1 of the present invention First amplifying section Operational amplifiers A1 to A4: LT1028 Feedback resistors RF1 to RF4: 760Ω Feedback resistors RS1 to RS4: 10Ω Resistors R1 to R4: 1 kΩ Gain: 77 times second amplification Operational amplifier Aa: AD829 Feedback resistance RFa: 300Ω Feedback resistance RSa: 1kΩ Gain: 1.3 times Gain of the entire amplifier: 100 times Conventional amplifier 51 1st amplification section Operational amplifiers A1 to A4: LT1028 Feedback resistances RF1 to RF4: 1.2kΩ Feedback resistance RS1 to RS4: 50Ω Combined resistance R1 to R4: 2kΩ gain: 25 times 2nd amplification section Operational amplifier Aa: AD829 Feedback resistance RFa: 2kΩ gain: 4 times Gain of entire amplifier: 100 times (test result) The obtained results show that the preamplifier PA of the present invention: noise level O. 47n
V / (Hz) ^1/2 Conventional amplifier 51: Noise level 0.64n
V / (Hz) ^1/2 , the amplifier AMP1 of the present invention is the same as the conventional amplifier 5
It was confirmed that the noise was significantly lower than that of 1.

In this embodiment, the number of feedback amplifier circuits in the first amplifier section is four, but it may be set to any number other than four. Further, the second amplification section may be replaced with an amplification circuit and used as a buffer. Further, the combined output of the first amplifying unit may be directly input to the load without providing the second amplifying unit, or another circuit such as an amplifying circuit may be further provided before the first amplifying unit.

In addition, the number of feedback amplifier circuits is appropriately set to
The amplification unit 10 may be configured as a unit, a plurality of these units may be connected, and the number of units used may be changed by a switch as necessary. In this way, there is an advantage that the first amplification section 10 having a desired noise level can be easily obtained while keeping the gain constant.

The signal amplified by the amplifier AMP1 is further amplified by the amplifier AMP2 and then input to the boxcar integrator 16 or the digital oscilloscope 17,
Therefore, time-resolved measurement is performed. The wave number resolution at the time of resolution is preferably about 4 to 16 cm ^{−1 in} consideration of the S / N ratio.

In this infrared spectroscopy apparatus, when the circuit configuration of FIG. 6 is used for the amplifier AMP1, the amplifier AMP1 carries out AC amplification, so in order to obtain the amount of change in the signal due to the excitation of the sample SAM, the signal before the trigger point must be obtained. It is necessary to subtract the signal level that is not affected by the excitation. Therefore, the digital oscilloscope 17 also acquires the data before the trigger point and numerically subtracts the average value from the entire waveform.

The laser device 12 is usually several kHz.
When the phenomenon to be observed can sufficiently follow it, the S / N ratio of the measurement can be improved by repeating the averaging. Since the integration in the boxcar integrator 16 is performed in an analog manner, it is possible to sufficiently follow such a phenomenon of high repetition. Therefore, when it is desired to obtain the transient infrared spectrum at a certain time after excitation, the boxcar integrator 1
6 is very effective.

(Time-Resolved Measurement) In the system of FIG. 1 using the infrared spectroscopic device of the present invention, time-resolved infrared spectroscopic measurement is performed as follows.

First, the intensity I of infrared light (monitor light) incident on the sample SAM without irradiating the sample SAM with laser light.
0 and the intensity I of infrared light transmitted through the sample SAM are measured.
These are sample SAMs while operating the chopper CH.
Can be obtained by taking it in and out of the infrared light optical path.

Next, a time-resolved measurement is performed while irradiating the sample SAM with laser light to excite it, and a transient change amount ΔI of the infrared light intensity transmitted through the sample SAM is obtained.

The transient absorbance change of the sample SAM caused by exciting with laser light can be obtained as follows.

When the original absorbance of the sample SAM is A (a value including the absorbance of the solvent and other components) and the transient absorbance change due to excitation is ΔA, the following equation is established.

[0114]

[Equation 1]

The change amount ΔI of the infrared light intensity is proportional to the intensity I. Since the ratio (ΔI / I) of the two is usually much smaller than the intensity I (<0.001), the following approximate expression holds.

ΔA≈−O. 43429 × (ΔI / I) From this equation, the change amount ΔI of the infrared light intensity can be calculated.

If the ratio r of the parent molecular species (absorbance AP) is changed to a transient molecular species (absorbance AT) by excitation, ΔA = rAT−rAP AT = Δ (A + rAP) / r The difference spectrum between the solution and the solvent is taken. If AP can be obtained by the method described above, the ratio r can be appropriately selected by the above equation, and the negative peak component in the ΔA spectrum can be removed to obtain A
You can ask for T.

The α-type metal-free phthalocyanine (α-obtained by the system of FIG. 1 using the infrared spectrometer of the present invention).
The time-resolved infrared absorption difference spectrum of H ₂ P _C) thin film shown in FIG. The upper spectrum is a transient absorption difference spectrum between 50 and 122 μs after photoexcitation, the middle spectrum is a transient absorption difference spectrum between 0 and 30 μs after photoexcitation, and the lower spectrum is a ground state absorption spectrum.

For the sample SAM, α-H was formed on the Nacl substrate.
_{Using 2} P _C vapor-deposited about 1.1 μm, for photoexcitation,
The second harmonic of a CW Q-switched Nd: YAG laser was used. The laser light intensity on the sample SAM is 19 μJ / pulse, the wave number resolution is 1240 cm ⁻¹ and 15 cm ⁻¹ , the sweep speed is 1.8 cm ⁻¹ / min, and the laser light repetition frequency is 2 kHz.

[0120] Looking at the vertical axis of FIG. 8, in the absorbance change ΔA is able to detect up to 3 × 10 ^over 7 (300 ppb), it can be seen very sensitive.

It was confirmed that this system is a sub-microsecond time-resolved infrared spectroscopy system having the following excellent performance.

Measurement wave number range 4000 to 700c
m ⁻¹ Maximum time resolution 100 nanoseconds or less Detection sensitivity ΔA Approximately 3 × 10 ⁻⁷ (300 ppb, maximum value) In the above embodiment, the laser device 12 was used as the excitation light source, but other known light sources ( A flash lamp, for example) can also be used.

[0123]

As described above, the first infrared spectroscopic device of the present invention has a simple structure and extremely low noise.

The second infrared spectroscopic device of the present invention is extremely stable in operation with respect to changes in ambient temperature.

[Brief description of drawings]

FIG. 1 is a functional block diagram showing a system configuration when performing time-resolved infrared spectroscopy using an embodiment of an infrared spectroscopy device of the present invention.

FIG. 2 is a schematic configuration diagram showing a configuration of an infrared light detector unit and its peripheral devices in the system of FIG.

FIG. 3 is a circuit diagram of the infrared light detector unit of FIG. 2 and its peripheral devices.

FIG. 4 is a perspective view of a head to which an infrared light detector is attached.

FIG. 5 is a circuit diagram showing an example of a circuit configuration of an amplifier.

FIG. 6 is a circuit diagram showing another example of the circuit configuration of the amplifier.

FIG. 7 is a conceptual diagram of a feedback amplifier.

8 is a time-resolved infrared absorption difference spectrum of the α-type metal-free phthalocyanine thin film measured by the system of FIG.

FIG. 9 is a circuit diagram showing an example of a conventional amplifier.

FIG. 10 is a circuit diagram showing another example of a conventional amplifier.

[Explanation of symbols]

10 Infrared spectroscope S Infrared light source AP1, AP2 Optical diaphragm L1, L2 Convex lens SHG Second harmonic generator P Prism M1, M3, M4, M5, M6, M7, M8, M10
Plane mirror M2 Convex mirror M9, M11 Concave mirror G Diffraction grating SL1, SL2 Optical slit F Filter CH Chopper SAM Sample 11 Pulse generator 12 Laser device 16 Boxcar integrator 17 Digital oscilloscope 18 Data processing computer 1 Infrared detector unit 2 Low-temperature container 2a chamber 3 Light introducing part of container 4 Window of light introducing part 5 Detector mounting head D Infrared photodetector LD Lead wire RB Bias resistance RD Infrared photodetector resistance VB Bias power supply TR transistor DA Differential Amplifier VS Reference power supply CS Capacitor R11, R12 Resistance V power supply AA AC amplifier C1 Coupling capacitor A1 ~ A4 Operational amplifier RS1 ~ RS4, RF1 ~ RF4 Feedback resistance R1 ~ R4 Resistance Aa Operational amplifier RFa Feedback resistance RSa Feedback resistance PA Preamplifier Q1 , Q2 Transistor Q3 Error amplifier Q4 Inverter RC Collector resistance RE Emitter resistance RFac AC feedback resistance RFdc DC feedback resistance R2, R3, R4 resistance C3, C4, C5 capacitor VREF Reference power supply VCC Power supply voltage

Claims (2)

[Claims] 1. An infrared light source, an infrared spectroscopic unit that disperses infrared light generated by the infrared light source, and an infrared signal generated by the infrared light source and an electric signal corresponding to the infrared light. Infrared light detecting means for generating, and an amplifying means coupled to the infrared light detecting means and amplifying an electric signal generated by the infrared light detecting means, wherein the amplifying means are connected in parallel with each other. A plurality of feedback amplifier circuits that are coupled to each other, combine a output of the amplifier circuits via a resistor, and a second amplifier portion that is coupled to the first amplifier portion. 2. The infrared spectroscopic device, wherein the input impedance of the second amplifier is equal to or more than the parallel resistance value of the resistors of the first amplifier. 2. An infrared light source, an infrared spectroscopic unit that disperses the infrared light generated by the infrared light source, and an electric signal corresponding to the infrared light generated by the infrared light source. Infrared light detecting means for generating, and comprising an amplifying means coupled to the infrared light detecting means and amplifying an electric signal generated by the infrared light detecting means, wherein the amplifying means is a negative feedback signal. The input impedance is matched with the source resistance, and the voltage across the load is controlled to be equal to the reference voltage, and the reference voltage cancels the change in the gain of the amplifying means according to the ambient temperature. Thus, the voltage control means, the AC negative feedback means for applying the AC negative feedback, and the DC negative feedback means for applying the DC negative feedback are provided, and the AC negative feedback means and the DC negative feedback means are set to the ambient temperature. Change accordingly Infrared spectrometer, characterized in that it has a temperature coefficient for canceling the change in the gain of the amplification means.