JPH0467775A - Switching power unit - Google Patents

Abstract

PURPOSE:To reduce loss and control a spike current by connecting the primary winding of a transformer with an inductance element and a first switching element in series, and connecting auxiliary winding with a rectifying means, a current limiting means, and a second switching element, and further connecting an output terminal to the secondary winding through a rectifying/smoothing circuit. CONSTITUTION:(b) shows the state of a current I'D, where the exciting energy accumulated in a transformer 3 is consumed by a reset circuit and the magnetic flux of the transformer 3 is reset, and a current IQ, where the flyback voltage induced in the tertiary winding 3c by the turn-on of an auxiliary switching means 6 is applied to a current limiting means 9 and is limited, being circulating. Next, for (c), since the said circulating current is broken by an auxiliary switch 9 being turned off, the energy accumulated in an inductance element 16 lets an exciting current I'Q flow to a DC power source 1, so that it may regenerate energy, through the parasitic capacity 18 of a switching means 4 turned off and the distributed capacity 17 of the transformer 3, so the voltage across the switching means 4 becomes zero.

Description

DETAILED DESCRIPTION OF THE INVENTION Field of the Invention The present invention relates to a switching power supply device that supplies DC stabilized voltage to industrial and consumer electronic equipment.

BACKGROUND OF THE INVENTION In recent years, as electronic devices have become cheaper, smaller, more sophisticated, and more energy efficient, there has been a strong demand for switching power supplies that are smaller, have more stable output, and are more efficient.

A conventional switching power supply device will be explained below.

FIG. 4 shows the circuit configuration of a conventional switching power supply, which is a so-called rectangular wave type power supply. In Figure 4,
1 is by rectifying and smoothing the commercial AC voltage using a DC power supply.
Or it is composed of a battery, etc., and the input terminal 2
, 2', the positive terminal is connected to the input terminal 2', and the negative terminal is connected to the input terminal 2'. 3 is a transformer, one end of the primary winding 3a is connected to the input terminal 2, the other end is connected to the input terminal 2' via the switch means 4, and both ends of the secondary winding 3b are connected to the rectifying and smoothing circuit 12. Connected. 4
is a switch means, which is turned on and off by an on/off signal from the control circuit 19 applied to a control terminal to apply or cut off the input voltage to the primary winding 3a. A rectifying and smoothing circuit 12 converts the AC voltage induced across the secondary winding 3b into a DC voltage and supplies the output voltage to the output terminals 14 and 14'. Reference numeral 13 denotes an output detection circuit, which detects the output voltage of the output terminals 14 and 14', compares it with an internal reference voltage, and transmits the amplified signal to the control circuit 19. 14,
14" is an output terminal, which supplies the output voltage supplied from the rectifier and smoothing circuit 12 to a load 15. 15 is a load, which flows an output current 1 OUT and consumes power. 1
Reference numeral 9 denotes a control circuit which changes the pulse width of the on/off signal applied to the control terminal of the switch means 4 in accordance with the signal transmitted from the output detection circuit 13, so as to control the output voltage to be constantly constant. A diode 20 has an anode connected to the connection point between the primary winding 3a and the switch means 4, and a cathode connected to the primary winding 3a and the input terminal 2 through a parallel circuit of a resistor 22 and a capacitor 21. These diodes 20. capacitor 2
1. A reset circuit constituted by a resistor 22 maintains the flyback voltage induced across the primary winding 3a, resets the magnetic flux of the transformer 3, and absorbs the spike voltage.

The operation of the switching power supply device configured as described above will be explained in detail with reference to FIGS. 5 and 6. 5(a) to (e) show the operating waveforms of each part of the switch means 4, (a) is the waveform of the switching voltage VDS applied to both ends of the switch means 4, and (b) is the waveform of the switching voltage VDS applied to both ends of the switch means 4.
is the waveform of the switching current ID flowing through the switch means 4, and (C) shows the on/off signal VGI of the control circuit 19 applied to the control terminal of the switch means 4, and in order to show the time change of each waveform state, t1 The signal from t4 to t4 is shown in the waveform. It can be seen that in the switching current waveform ID flowing through the switch means 4, a large spike current IDP is generated when the switch means 4 is turned on (time indicated by t3). This is due to the charging and discharging currents to distributed capacitances such as line capacitance and interlayer capacitance generated in each winding of the transformer 3, and the charging and discharging currents to parasitic capacitances related to the switch means 4. FIGS. 6(a) and 6(b) are explanatory diagrams equivalently showing how the spike current rop is generated,
In FIG. 6, (a) shows the distributed capacitance 17 of the transformer 3.
and the parasitic capacitance 18 of the switch means 4 are shown in equivalent form, and the flow of the spike current rop when the switch means 4 is turned on is shown. FIG. 6(b) shows the use of an MO8FET transistor in the switch means 4 The drain-source capacitance C0 corresponding to the parasitic capacitance 18 when
5S shows the flow of charging and discharging current, and the loss PON generated in the switch means 4 due to these capacitances is PON-
(approximately represented by (CI7+ CHB> , and f is the switching frequency.Thus, the distributed capacitance 17 and the parasitic capacitance ji18 cause an increase in noise, a decrease in reliability, and an increase in loss due to the spike current when the switch means 4 is turned on. On the other hand, the excitation energy of the primary winding 3a of the transformer 3 generated when the switch means 4 is turned off (time indicated by t4) absorbs the spike voltage generated by the excitation energy of the case inductance, and the steep switching voltage waveform VOS is absorbed. It acts to prevent rising, reduces the noise and loss caused by the spike voltage when the switch means 4 is turned off, and improves reliability.In FIG. However, similar operations occur in other configurations such as the so-called flyback configuration and feedforward configuration.

FIG. 7 shows another circuit diagram of a conventional switching power supply, which is a so-called voltage resonance type power supply. In FIG. 7, the same parts as in FIG. 4 are denoted by the same reference numerals, and the description thereof will be omitted. In Fig. 7, 1 is a DC power supply, 2 and 2' are input terminals, 3 is a transformer, 4 is a switch means, 12 is a rectifier and smoothing circuit, and 13 is an output detection circuit. 14 and 14' are output terminals, 15 is a load, and 19 is a control circuit. 5 is a diode whose anode is connected to the input terminal 2' and whose cathode is connected to the connection point between the switch means 4 and the primary winding 3a, so that even when the switch means 4 is off, the power is regenerated to the DC power supply 1. Allows current to flow. 24 is a leakage inductance or inductance element (hereinafter simply referred to as leakage inductance) between the primary winding 3a and the secondary winding MSb of the transformer 3, and is connected in series between the input terminal 2 and the primary winding 33. be done. 23 is a capacitor;
One end is connected to the connection point between the input terminal 2 and the leakage inductance 24, the other end is connected to the connection point between the primary winding 3a and the switch means 4, and the capacitor 23 and the leakage inductance 24 or the primary winding 3a are connected to each other. constitutes a resonant circuit.

The operation of the switching power supply device configured as described above will be explained in detail with reference to FIG. 8 as well. Figure 8 (a
) to (C) show the operating waveforms of each part of the switch means 4, (a) is the waveform of the switching voltage VDS applied to both ends of the switch means 4, and (b) is the waveform of the switching voltage VDS applied to both ends of the switch means 4 and the diode 5. (C) shows the on/off signal VGI of the control circuit 19 applied to the control terminal of the switch means 4;
Symbols t1 to t4 are written on the waveforms to indicate temporal changes in each waveform state, and the solid line of each waveform indicates a high load when the current consumption of the load 15 is large, and the dotted line indicates a light load. Excitation energy is accumulated in the transformer 3 and leakage inductance 24 during the on period (ta to t4 period) of the switch means 4, and the accumulated excitation energy of the leakage inductance 24 or the transformer 3 is released to the capacitor 23 during the off period (1, to t2). A voltage that oscillates sinusoidally at a resonance frequency determined by the inductance value L24 of the leakage inductance 24 or the inductance value LP of the primary winding 3a and the capacitance value C23 of the capacitor 23 is applied to both ends of the capacitor 6 and the switch means 4. occurs at both ends. Further, since the sinusoidally oscillating voltage oscillates around the input voltage of the DC power supply 1 or the sum of the input voltage and the flyback voltage deficiency generated in the primary winding 3a, the capacitor 23
If the amplitude of the voltage across the switch means 4 is sufficiently large, the period during which the voltage across the switch means 4 is zero and current flows through the diode 5 (
t2-t3 period), and by turning on the switch means 4 by one control circuit in synchronization with this period, it becomes possible to perform zero-cross switching. In such a voltage resonance type power supply, the voltage waveform applied when the switch means 4 is turned on and turned off changes gradually in a sinusoidal manner regardless of the response speed of the switch means 4, so that the current waveform becomes steep. Even if the voltage changes, switching loss is small, and since the voltage waveform is a sine wave, switching noise is also very low. However, in such a voltage resonance type power supply, unless the switch means 4 is turned on at the zero cross, the accumulated charge in the capacitor 23 will be short-circuited by the switch means 4, as shown by the dotted line in FIG. The switching means 4 may be destroyed, switching loss may increase rapidly, and switching noise may increase. It is necessary to constantly secure the excitation energy, that is, resonance energy, of the front inductance 24 to ensure the voltage amplitude that oscillates in a sinusoidal waveform generated in the resonant circuit. It is very difficult to secure the excitation energy in a wide control range due to a decrease in the excitation energy of the J-cage inductance 24 due to a decrease in the switching current value due to a decrease in the switching current value due to a decrease in the switching current value due to a decrease in power consumption of the load 15, etc. If the amplitude is set large in advance and secured, the switch means 4 will need to have a high withstand voltage, so no effective solution has been found to date.

In FIG. 7, the polarity of the secondary winding 3b and the rectifying and smoothing circuit 12 operate in the same manner regardless of the so-called flyback configuration or feedforward configuration.

FIG. 9 shows another circuit configuration of a conventional switching power supply, which is a so-called current resonance type power supply. In FIG. 9, the same parts as in FIGS. 4 and 7 are denoted by the same reference numerals, and explanations thereof will be omitted. In Figure 9, 1 is a DC power supply, 2 and 2' are input terminals, 3 is a transformer, and 4
is a switch means, 5 is a diode, 12 is a rectifier and smoothing circuit, 13 is an output detection circuit, and 14
゜14' is an output terminal, 19 is a control circuit, and 2
0 is a diode, 21 is a capacitor, 22
is resistance. 25 is a secondary leakage inductance or inductance element (hereinafter simply referred to as leakage inductance) between the primary winding 3a and the secondary winding 3b of the transformer 3, and the secondary winding 3b and the rectifying and smoothing circuit 12 connected in series between. 26 is a capacitor;
It is connected to both ends of the secondary winding 3b via the secondary leakage inductance 25, and the secondary leakage inductance 25 and the capacitor 26 form a resonant circuit.

The operation of the switching power supply device configured as described above will be explained in detail with reference to FIG. 10 as well. 10(a) to (C) show operating waveforms of each part of the switch means 4, (a) shows the waveform of the switching voltage VDS applied across the switch means 40, and (b) shows the waveform of the switching voltage VDS applied to both ends of the switch means 40. Switching current ID flowing through 4 and diode 5
(C) shows the on/off signal VGI of the control circuit 19 applied to the control terminal of the switch means 4, and the signals from t1 to t4 are written in the waveform to show the time change of each waveform state. There is. The on period of the switch means 4 (1+
~t4 period), the resonance frequency fc=' determined by the inductance value L25 of the secondary side leakage inductance 25 and the capacitance value C26 of the capacitor 26 that constitute a resonance circuit by the voltage induced across the secondary winding 3b.
-r-100'', a sinusoidal oscillating current flows at π 2π.As a result, the switching current ID flowing through the switch means 4 also oscillates sinusoidally, and especially at the point where the sinusoidal current rises from zero and becomes negative. Zero-cross switching can be achieved by turning off the switch means 4.With such a current resonant power supply, the waveform of the current flowing when the switch means 4 is turned on and off changes gently with a sinusoidal slope. Therefore, even if the voltage waveform changes sharply, the switching loss is relatively small (and since the current waveform is a sine wave, the switching noise is also very small. However, in such a current resonant power supply, the switching means 4 is turned on. As explained in detail in FIGS. 4, 5, and 6, at time t3, a spike current TDP occurs due to charging and discharging of the parasitic capacitance generated in connection with the distributed capacitance of the transformer 3 and the switch means 4. Therefore, it is inferior to the voltage resonance type power source in terms of noise and loss, and the limit for increasing the frequency is said to be 1 to 2 MHz.

Problems to be Solved by the Invention However, in the conventional configuration described above, in the case of the rectangular wave power supply shown in FIG. 4, both the switching current and the voltage waveform are rectangular waves with steep rises and falls, and even larger switching noise is generated. and switching loss increases, making it impossible to increase the frequency. In the case of the voltage resonance type power supply shown in FIG. 7, it is difficult to ensure zero-cross switching and, furthermore, an excessive voltage is applied to the switch means 4, so that stable control over a wide range cannot be performed. In the case of the voltage resonant power supply shown in FIG. 9, there is a limit to how high the frequency can be increased due to switching loss and switching noise due to the generation of large spike currents. This limitation of increasing the frequency hinders the miniaturization of the switching power supply device, and the switching means 4
Excessive voltages and spike currents applied to the switch deteriorate reliability, and difficulty in control deteriorates output stability, making it impossible to meet the technological trends of switching power supplies.

The present invention solves the above-mentioned conventional problems, and aims to provide a switching power supply device that realizes stable zero-cross switching to reduce loss, suppress spike current, and reduce noise, and is capable of operating at high frequencies. purpose.

Means for Solving the Problem In order to solve this object, a switching element of the present invention, an inductance element, and a first switching element that repeats on/off are connected in series, and a connection point between the primary winding and the inductance element is connected in series. An auxiliary winding having one end connected to the auxiliary winding, a rectifier at both ends of the auxiliary winding, a current limiting means, a second switching element that alternately turns on and off with the first switching element, and the inductance element form a series circuit. The output terminal is connected to the secondary winding of the transformer via a rectifying and smoothing means.

Operation With this configuration, when the first switching element is turned off, the auxiliary winding of the transformer, the rectifying means, the current limiting means, and the second
Circulating current flows through the switching element and inductance element, and energy is stored in the inductance element. When the second switching element turns off, the stored energy causes current to flow through the distributed capacitance of the second winding of the capacitor and the parasitic capacitance of the switching element, increasing the distributed capacitance.

Zero cross switching can be achieved by turning on and off the switching elements. Further, since the circulating current is stably obtained against changes in the repeated on/off state of the first switching means by controlling the output voltage, stable zero-cross switching can always be maintained.

EXAMPLE An example of the present invention will be described below with reference to the drawings.

FIG. 1 shows the configuration of a switching power supply device in an embodiment of the present invention. In FIG. 1, the same parts as in FIG. 4 are denoted by the same symbols, and explanations thereof will be omitted. In Fig. 1, 1 is a DC power supply, 2゜2゛ is an input terminal, and 3 is a transformer with a primary winding 3a.

It has a secondary winding 3b and a tertiary winding 3C, 4 is a switch means, 12 is a rectifying and smoothing means, 13 is an output detection circuit, 14 and 14' are output terminals, and 15 is a It's a load. 16 is an inductance element, and the DC power supply 1. The primary winding 3a, the inductance element 16. The switch means 4 are connected to form a series circuit and store energy for accumulated charges such as the distributed capacitance of the transformer 3 and the parasitic capacitance of the switch means 4. 5 is a diode, whose anode is connected to the input terminal 2'' and whose cathode is connected to the connection point between the switch means 4 and the inductance element 16, so that even when the switch means 4 is off, the current regenerated to the DC power supply 1 is Reference numeral 6 denotes an auxiliary switch means, one end of which is connected to the connection point between the inductance element 16 and the switch means 4, and the other end connected to the connection point of the current limiting means 9, the diode 8, and the tertiary winding 3C connected in series. The primary winding 3a and the inductance element 16 through the circuit.
The second on/off signal of the control circuit 11 is applied to the control terminal, which repeatedly turns on and off, thereby turning on and off the current flowing through the tertiary winding 3c, the diode 8° current limiting means 9. A diode 8 has an anode connected to the tertiary winding 3c and a cathode connected to the current limiting means 9, which limits the flyback voltage induced in the tertiary winding 3C. from line 3C to said diode 8
limit the current flowing through the 7 is a capacitor, 10 is a resistor, and capacitor 7 and resistor 10 are connected in parallel,
One end is connected to the connection point between the diode 8 and the current limiting means 9, and the other end is connected to the input terminal 2. It is used to maintain the flyback voltage induced across the transformer 3a and reset the magnetic flux of the transformer 3. Reference numeral 11 denotes a control circuit 11, which applies first and second on/off signals to the control terminals of the switch means 4 and the auxiliary switch means 6, respectively, and receives the signal from the output detection circuit 13 so that the output voltage becomes constant. Controlling the pulse width of the first on/off signal applied to the switch means 4,
A second on/off signal applied to the auxiliary switch means 6 is also controlled so that the auxiliary switch means 6 is turned on only while the switch means 4 is off.

The operation of the switching power supply device configured as described above will be explained in detail with reference to FIGS. 2 and 3. FIGS. The operating waveforms of each part are shown, (a) is the switching voltage waveform VOS applied to both ends of the switch means 4, and (b) is the switching current waveform (D) flowing through the switch means 4 and the diode 5. (C) shows the first on/off signal VGI of the control circuit 11 applied to the control terminal of the switch means 4, (d) shows the waveform of the switching current IQ flowing through the auxiliary switch means 6, and (e) indicates the second on/off signal VG2 of the control circuit 11 applied to the control terminal of the auxiliary switch means 6, and symbols t1 to t4 are written on the waveforms to indicate temporal changes in each waveform state. It can be seen from the switching current waveform ID flowing through the switching means 4 that a negative current flows during the period (t2 to t3 period) immediately before the switching means 4 is turned on, resulting in zero cross switching.Next, referring to FIG. A detailed explanation of the operation resulting in zero cross switching will be given below. Figures 3 (a) to (C) are diagrams of equal ffE circuits in each waveform state. Figure 3 (a')
is the period during which the switch means 4 is on (t, 3 to t4 period)
(b) shows the state during the period (period 1 to t2) when the switch means 4 is turned off and the auxiliary switch means 6 is on, and (c) shows the state when the auxiliary switch is turned on immediately before the switch means 4 is turned on. The period during which the switch means 6 is off (
t2 to t3 period). In Figure 3, 17
18 equivalently represents the distributed capacitance of the transformer 3, and 18 equivalently represents the parasitic capacitance of the switch means 4. Figure 3 (a
) is a state in which a current, in which the excitation current or the output current is also superimposed, flows from the DC power supply 1 to the primary winding 3a. FIG. 3(b) shows that the excitation energy stored in the transformer 3 is passed through the primary winding 3a and the tertiary winding 3c to the diode 8. The current (T
D, and the flyback voltage induced in the tertiary winding 3c by turning on the auxiliary switch means 6 is the current limiting means 9.
A current I applied to the current limiting means 9 and limited by the current limiting means 9
Q is the tertiary winding 3c, the diode 8. Current limiting means 9
゜Auxiliary switch means 6. This figure shows how current circulates through the inductance element 16 (hereinafter referred to as circulating current). This circulating current excites the transformer 3 only slightly and does not affect the reset operation of the transformer 3, and the energy represented by the excitation energy U-L+6IQ2 is accumulated only in the inductance element 16. Here, L, +s
is the inductance value of the inductance element 16,
IQ indicates the circulating current value. Next, as shown in FIG. 3(c), the auxiliary switch means 6 is turned off and the circulating current is cut off, so that the excitation energy accumulated in the inductance element 16 is transferred to the parasitic energy of the switch means 4 which is turned off. The exciting current is set so as to regenerate energy to the DC power source 1 via the capacitor 18 and the distributed capacitor 17 of the transformer 3. , the voltage across the switch means 4 becomes zero, and then the switch means 4 is turned on, resulting in zero cross switching. The conditions under which such zero cross switching is possible are as follows: Since all of this energy is regenerated to the DC voltage source 1, no loss occurs.

Here, C10 is the equivalent capacitance value of the distributed capacitance 17, and C10 is the equivalent capacitance value of the distributed capacitance 17.
10 is the equivalent capacitance value of the parasitic capacitance 18, and VOS is the voltage value applied to both ends of the switch means 4 during the off period. This means that regardless of the on/off time width of the switch means 4 that is varied to control the output voltage and the magnitude of the output current, the circulating current IQ is limited to the induced voltage of the tertiary winding 3c by the current limiting means 9. Since the value is determined, stable zero-cross switching can always be ensured.

Furthermore, if the circulating current IQ is constant as shown in FIG. It can also reduce the losses that occur. Furthermore, there is a slight delay between turning off the auxiliary switch means 6 and turning on the switch means 4 (t2 to t
3 period) is necessary to obtain the charging/discharging time of the distributed capacitance 17 and the parasitic capacitance 18 until the voltage across the switching means 4 becomes zero voltage, and the value is approximately t3 t2'' L+e lc+7+c+sl. and this becomes the inductance element 16.The first
In the figure, the polarity of the secondary winding 3b and the rectifying and smoothing circuit 12 operate in the same manner regardless of the so-called flyback configuration or feedforward configuration.

As described above, according to this embodiment, during the OFF period of the switch means 4, the current limiting means 9. Auxiliary switch means 6. By configuring the circulating current to flow through the tertiary winding 3C, it is possible to prevent the spike current generated when the switch means 4 is turned on, and zero-cross switching can be achieved, and furthermore, the spike current that occurs when the switch means 4 is turned off can be prevented. The excitation energy key of the primary winding 3a of the transformer 3 and the inductance element 1
By actively absorbing the spike voltage generated by 6 into the distributed capacitance 17 and the parasitic capacitance 18, or by connecting a capacitor to the outside and equivalently increasing the distributed capacitance 17 and the parasitic capacitance 18, the spike voltage is absorbed even more. It prevents the steep rise of the switching voltage waveform and allows turn-off to be zero-cross switching without loss, resulting in low noise.

Low loss, high efficiency, and high frequency can be achieved. The losses due to the circulating current include losses due to the winding resistance of the primary winding 3a and tertiary winding 3c, losses in the current limiting means 9, and switching losses in the auxiliary switch means 6. By designing it, there will be no big loss. Furthermore, by using an inductance element as the current limiting means 9, the loss of the current limiting means 9 can be eliminated.

Further, a circuit for resetting the excitation of the transformer 3 during the off-period of the switch means 4 is constructed using a diode 8 and a resistor 10. A similar effect can be obtained by using a power consumption type reset circuit consisting of the capacitor 7 or by using another reset circuit. In particular, by connecting one end of the capacitor 7 to the input terminal 2 and connecting the other end to the connection point between the auxiliary switch means 6 and the current limiting means 9, a lossless type can be achieved. Furthermore, it goes without saying that the present invention can be similarly applied to a switching power supply circuit constituted by the fifth type.

As described above, the present invention provides a tertiary winding connected in parallel to the primary winding of a transformer, a rectifying means, a current limiting means, an auxiliary switch means, and a tertiary winding connected between the switch means and the primary winding. A series circuit of inductance elements is connected to turn on and off the auxiliary switch means complementary to the switch means. When the auxiliary switch means is turned on, a circulating current flows through the tertiary winding, and when the auxiliary switch means is turned off, the accumulated charge of the distributed capacitance of the transformer and the parasitic capacitance of the switch means is discharged, thereby reducing the voltage across the switch means. Achieves constant and stable zero-cross switching, which turns on when the voltage is zero, resulting in low noise and low loss.

This makes it possible to realize an excellent switching power supply device that is highly efficient, highly reliable, and has a wide control range.

[Brief explanation of the drawing]

1 is a circuit configuration diagram of a switching power supply according to an embodiment of the present invention, FIG. 2 is an operating waveform diagram of the circuit configuration diagram of FIG. 1 of the present invention, and FIG. 3 is a circuit diagram of the circuit configuration of FIG. 1 of the present invention. An explanatory diagram of the configuration diagram, FIG. 4 is a circuit configuration diagram of a conventional switching power supply device, FIG. 5 is an operating waveform diagram of the conventional circuit configuration diagram of FIG. 4, and FIG. 6 is a conventional circuit configuration diagram of FIG. 4. 7 is a circuit diagram of another conventional switching power supply device, and FIG. 8 is an explanatory diagram of the diagram.
The figure is an operating waveform diagram of the conventional circuit diagram of FIG. 7, FIG. 9 is a circuit diagram of another conventional switching power supply, and FIG.
This figure is an operation waveform diagram of the conventional circuit diagram of FIG. 9. 1...DC power supply, 2, 2'...Input terminal, 3...Transformer, 4...Switch means, 5, 8.22... ...Diode, 6...
... Auxiliary switch means, 7 ... Capacitor,
9... Current limiting means, 10.24...
Resistor, 11... Control circuit, 12... Rectifying and smoothing means, 13... Output detection circuit, 14.1
4"...Output terminal, 15...Load, 1
6...Inductance element. Name of agent: Patent attorney Shigetaka Awano and one other person Figure C. Figure Figure (σ) / (b)

Claims (1)

[Claims] Connect the primary winding of a transformer, an inductance element, and a switching means that repeats on and off in series to the DC input terminal.
A tertiary winding having one end connected to a connection point between the primary winding and the inductance element, and a rectifier at both ends of the tertiary winding, a current limiting means, and an auxiliary switch means that alternately turns on and off with the switch means. and the inductance element are connected to form a series circuit, and an output terminal is connected to the secondary winding of the transformer via a rectifying and smoothing means.