JPH0398463A - Switching power source device - Google Patents

Abstract

PURPOSE:To realize zero-cross switching, in which the ON/OFF of a first switching means is effected at both terminal voltages of zero, stably at all times by a method wherein a second switching means is turned ON/OFF complementarily to the first switching means while the second switching means is turned ON to conduct a current which circulates between a tertiary winding and a primary winding. CONSTITUTION:A first switching means 4 is turned OFF and a second switching means 6 is put ON whereby circulating current is conducted between tertiary winding 3c and a primary winding 3a. Next, the second switching means 6 is turned OFF whereby the circulating current is intercepted and a counter electromotive voltage is generated in a direction, in which energy is regenerated from the primary winding 3a into a DC input voltage, whereby zero-cross switching, in which the both terminal voltage of the first switching means 4 is nullified before switching it ON, is effected. The circulating current is obtained stably in spite of the change of the repeating condition of the ON/OFF of the first switching means 4.

Description

DETAILED DESCRIPTION OF THE INVENTION Field of the Invention The present invention relates to a switching power supply device that supplies DC stabilized voltage to industrial and consumer electronic equipment.

Conventional technology In recent years, as electronic equipment has become cheaper, more compact, more sophisticated, and more energy efficient, there has been a strong demand for switching power supplies that are smaller, have higher output stability, are more efficient, and have lower noise. There is. A conventional switching power supply device will be explained below.

FIG. 11 shows the circuit configuration of a conventional switching power supply device, which is a so-called rectangular wave type power supply.

In Fig. 11, 1 is a DC power supply that rectifies and smoothes commercial AC voltage, or is configured with a battery, etc., and supplies input voltage to input terminal 2-2°, and outputs a positive voltage to input terminal 2. and a negative voltage is connected to input terminal 2'. A transformer 3 has one end of a primary winding 3a connected to an input terminal 2, the other end connected to an input terminal 2' via a switch means 4, and both ends of a secondary winding 3b connected to a rectifying and smoothing circuit 12.
is connected to. Reference numeral 4 denotes a switch means, which is turned on and off by an on/off signal from the control circuit 30 applied to a control terminal to apply or cut off the input voltage to the primary winding 3a. A rectifying and smoothing circuit 12 converts the alternating current voltage induced across the secondary winding 3b into a direct current voltage, and supplies the output voltage to the output terminals 14-14. Reference numeral 13 denotes an output detection circuit, which detects the output voltage at the output terminals 14-14°, compares and amplifies the signal with an internal reference voltage, and transmits the signal to the control circuit 30. 14-14'' is an output terminal, which supplies the output voltage supplied from the rectifying and smoothing circuit 12 to the load 15. 15 is a load, and the output voltage V O
It consumes UT and causes output current I OUT to flow. Reference numeral 30 denotes a control circuit which changes the pulse width of the on/off signal applied to the control terminal of the switch means 4 in accordance with the signal transmitted from the output detection circuit 13, so as to control the output voltage to be constantly constant. 31 is a diode whose anode is connected to the connection point between the primary winding 3a and the switch means 4, and whose cathode is connected to the primary winding 3a and the input terminal 2 through a parallel connection circuit of a resistor 33 and a capacitor 32. A reset circuit composed of a diode 31, a capacitor 32, and a resistor 33 maintains the flyback voltage induced across the primary winding 3a and resets the magnetic flux of the transformer 3. belongs to.

The operation of the switching power supply device configured as described above will be explained in detail with reference to FIGS. 12 and 13 as well. FIG. 12 (al to fcl show operating waveforms of each part of the switch means 4, (a) is the switching voltage waveform applied to both ends of the switch means 4;
b) is the switching current waveform 1 flowing through the switch means 4.
o, and (C) shows the on/off signal VC+ of the control circuit 30 applied to the control terminal of the switch means 4,
Signals 71-74 are shown in the waveforms to show temporal changes in each waveform state. The switching current waveform 1o flowing through the switch means 4 is when the switch means 4 is turned on (t
It is observed that a large spike current IDP is generated at the time indicated by 3). This is due to charging and discharging currents to distributed capacitances such as line capacitances and interlayer capacitances generated in each winding of the transformer 3, and discharging currents from parasitic capacitances related to the switch means 4. 13(al) and (b) are explanatory diagrams equivalently showing how the spike current IDP is generated. In FIG. 13 (
b) shows the flow of discharge of the drain-source capacitance Coss, which corresponds to the parasitic capacitance 41, when a MOSFET} transistor is used as the switch means 4, and the loss generated in the switch means 4 due to these capacitances. PON is indicated. Here, C4o is the capacitance value of the distributed capacitance 40, C41 is the capacitance value of the parasitic capacitance 41, and V
os is the voltage across the switch means 4 before turning on, and f is the switching frequency.

In this way, the distributed capacitance 40 and the parasitic capacitance 41 cause an increase in noise, a decrease in reliability, and an increase in loss due to the spike current when the switch means 4 is turned on. It absorbs the excitation energy of the primary winding 3a of the transformer 3 and the spike voltage generated by the leakage inductance during the time (time), and prevents the switching voltage waveform VDS from rising sharply. Reduces noise and loss caused by spike voltages and improves reliability. In FIG. 11, the polarity of the secondary winding 3b and the rectifying and smoothing circuit 12 operate in the same manner regardless of the so-called flyback configuration or feedforward configuration.

FIG. 14 shows another circuit diagram of a conventional switching power supply, which is a so-called voltage resonance type power supply. 14th
In the figure, the same parts as in FIG. 11 are denoted by the same reference numerals, and the explanation will be omitted. In FIG. 14, 1 is a DC power supply,
2-2° is an input terminal, 3 is a transformer, 4 is a switch means, 12 is a rectifier and smoothing circuit, and 13
is an output detection circuit, 14-14° is an output terminal, 15 is a load, and 30 is a control circuit. 5 is a diode whose anode is connected to the switch means 4 and the input terminal 2.
The cathode is connected to the connection point between the switch means 4 and the primary winding 3a, so that the regenerated current flows through the DC power supply 1 even when the switch means 4 is off. 19 is a leakage inductance or inductance element between the primary winding 3a and the secondary winding 3b of the transformer 3, and is connected in series between the input terminal 2 and the primary winding 3a. 16 is a capacitor, the end of which is connected to the connection point between input terminal 2 and reincarnation inductance 19;
The other end is connected to the connection point between the primary winding 3a and the switch means 4, and the capacitor 16 and leakage inductance 19 are connected to each other.
Alternatively, it forms a resonant circuit with the primary winding 3a.

The operation of the switching power supply device configured as described above will be explained in detail with reference to FIG. 15 as well. FIG. 15 1al~(Cl indicates the operating waveform of each part of the switch means 4, fa) is the switching voltage waveform Vos applied to both ends of the switch means 4, and (b) shows the operation waveform of the switch means 4 and the diode 5. Switching current waveform 1 flowing through
o, and {C} is the on/off signal V. of the control circuit 30 applied to the control terminal of the switch means 4. The symbols t1 to t4 are written on the waveforms to show the temporal changes in each waveform state, and the solid line of each waveform shows the heavy load when the power consumption of the load 15 is large, and the dotted line shows the light load. It shows the time. On period of switch means 4 (t3 to t4 period)
The excitation energy is accumulated in the transformer 3 and the leakage inductance 19 during the off period (t+=t2), and the accumulated excitation energy of the leakage inductance 19 or the transformer 3 is released to the capacitor 16, and the inductance value LI9 of the leakage inductance 19 or the above 1
Resonant frequency f c =1/2 determined by the inductance value Lp of the next winding and the capacitance value CI6 of the capacitor 16
A voltage that oscillates in a sinusoidal manner at π or fc = 1/2 week is applied to both ends of the capacitor 6 and the switch means 4.
occurs at both ends. Furthermore, since the sinusoidally oscillating voltage oscillates around the input voltage of the DC power supply 1 or the sum of the input voltage and the flyback voltage generated in the primary winding 3a, the voltage across the capacitor 16 is If the amplitude is sufficiently large, the voltage across the switch means 4 is zero and the current is flowing through the diode 5 (period t2 to t3).
Zero-cross switching can be achieved by generating this period and turning on the switch means 4 by the control circuit 30 in synchronization with this period. In such a voltage resonant power supply, the voltage waveform applied when the switch means 4 is turned on and off changes slowly with a sinusoidal slope regardless of the response speed of the switch means 4, so the current waveform does not change sharply. However, switching loss is small, and since the voltage waveform is a sine wave, switching noise is also very low. However, such a voltage resonant type N
If the zero-cross switching is not performed without fail when the switch means 4 is turned on, the stored charge in the capacitor 16 will be short-circuited by the switch means 4 as shown by the dotted line in FIG. 15, and the switch means 4 will be destroyed. , switching loss may increase rapidly, and switching noise may increase. In order to ensure the zero-cross switching, it is necessary to constantly secure the excitation energy, that is, resonance energy, of the transformer 3 and the reincarnation inductance 19 during the ON period of the switch means 4, and to secure the voltage amplitude that oscillates in a sinusoidal waveform generated in the resonance circuit. However, due to a decrease in excitation energy due to a decrease in the on-period for controlling the output voltage and a decrease in excitation energy for the leakage inductance due to a decrease in switching current value due to a decrease in power consumption of the load 15, etc. It is very difficult to secure the excitation energy in a wide control range, and if it were to be secured by setting the voltage amplitude large in advance, the switch means 4 would need a high withstand voltage, so there are currently no effective solutions. has not been discovered.

In FIG. 14, the polarity of the secondary winding 3b and the rectifying and smoothing circuit 12 operate in the same manner regardless of the so-called flypack configuration or feedforward configuration.

FIG. 16 shows another circuit configuration of a conventional switching power supply, which is a so-called current resonance type power supply. In FIG. 16, the same parts as in FIG. 11 and FIG.

In FIG. 16, 1 is a DC power supply, 2-2 is an input terminal, 3 is a transformer, 4 is a switch means, 5 is a diode, 12 is a rectifier and smoothing circuit, and 13 is a rectifier and smoothing circuit. In the output detection circuit, 14-14' is an output terminal, 30 is a control circuit, 31 is a diode, 32 is a capacitor, and 33 is a resistor. 1
Reference numeral 7 denotes a secondary side leakage inductance or inductance element between the primary winding 3 a and the secondary winding 3 b of the transformer 3 , and is connected in series between the secondary winding 3 b and the rectifying and smoothing circuit 12 . 18 is a capacitor, which connects the secondary winding 3b via the secondary side leakage inductance 17.
The secondary side leakage inductance 17 and the capacitor 18 constitute a resonant circuit.

The operation of the switching power supply device configured as described above will be explained in detail with reference to FIG. 17 as well. FIG. 17 fa) to fcl show the operating waveforms of each part of the switch means 4, (al is the switching voltage waveform V.5 applied to both ends of the switch means 40, and {b} is the operation waveform of the switch means 4 and the diode. Switching current waveform I flowing through 5
t, and (C) shows the on/off signal VGI of the control circuit 30 applied to the control terminal of the switch means 4, and symbols t1 to t4 are written on the waveforms to indicate temporal changes in each waveform state. ing. The on period of the switch means 4 (t3
~t4 period), the resonance frequency f c = 1 determined by the inductance value LI7 of the secondary side leakage inductance l7 and the capacitance value Cps of the capacitor 18, which constitute a resonant circuit by the voltage induced across the secondary winding 3b.
/ 2π A sinusoidal oscillating current flows at Ll7CI8. As a result, the switching current waveform 1o flowing through the switch means 4 also oscillates in a sinusoidal manner, and especially when the sinusoidal current rises from zero and becomes negative, the switch means 4
It is possible to perform zero cross switching by turning off. In such a current resonance type power supply, the waveform of the current flowing when the switch means 4 is turned on and off changes slowly with a sinusoidal slope.
Even if the voltage waveform changes sharply, switching loss is relatively small, and since the current waveform is a sine wave, switching noise is also very low. However, in such a current resonance type power supply, when the switch means 4 is turned on for 2 hours (+1 hour), the distributed capacitance of the transformer 3 and the switch means 4 are Since a spike current rDP is generated due to the charging and discharging current of the parasitic capacitance that occurs in connection with
It is said to be MHz.

Problems to be Solved by the Invention However, in the conventional configuration described above, in the case of the rectangular wave type power supply shown in FIG. Since current is also generated, switching noise and switching loss increase, making it impossible to increase the frequency. In the case of the voltage resonance type power supply shown in FIG. 14, the resonance energy changes greatly as the output voltage is controlled, making it difficult to ensure zero-cross switching, and furthermore, an excessive voltage is applied to the switch means 4. Wide range and stable control is not possible. In the case of the voltage resonant power supply shown in FIG. 16, there is a limit to high frequency because of switching loss and switching noise due to the generation of large spike currents. These limits to higher frequencies impede the miniaturization of switching power supplies, excessive voltages and spike currents applied to the switch means 4 deteriorate reliability, and increased switching losses deteriorate efficiency and control. The difficulty was that it could not satisfy the technical trends of switching power supplies, such as deterioration of output stability.

The present invention solves the above-mentioned conventional problems, and aims to provide a switching power supply device that achieves stable zero-cross switching to reduce loss, suppress spike current, and reduce noise, and which enables higher frequencies. purpose.

Means for Solving the Problems In order to solve the problems, the switching power supply device of the present invention includes a transformer having at least a primary winding, a secondary winding, and a tertiary winding, and a transformer having a transformer at one end of the primary winding. A first switch means that is connected and repeats on/off, a DC input voltage that supplies voltage to the primary winding via the first switch means, and a rectifying and smoothing means that is connected to the secondary winding. an output for providing a DC output voltage, said tertiary winding being connected in parallel across said primary winding via a series connection of rectifier means, current limiting means and second switch means; Second
The switch means repeatedly turns on and off in a complementary manner to the first switch means, and the tertiary winding has a larger number of turns than the primary winding, and when the second switch means is turned on, the current is limited. The means is configured such that a difference in induced voltage between the primary winding and the tertiary winding is applied.

Operation With this configuration, the first switch means is turned off and the second switch means is turned off.
When the switch means is turned on, a circulating current is caused to flow between the tertiary winding and the primary winding, and when the second switch means is turned off, the circulating current is interrupted and DC input from the primary winding is caused. Zero-cross switching can be achieved by generating a back electromotive force in the direction in which energy is regenerated in the voltage, and turning on the voltage across the first switching means after the voltage across the first switching means is zero. This is achieved stably regardless of the repeated on/off state changes of the switching means, so that stable zero-cross switching can always be maintained.

EXAMPLE Hereinafter, an example of the present invention will be described with reference to the drawings.

FIG. 1 shows a circuit configuration diagram of a switching power supply device according to a first embodiment of the present invention.

In FIG. 1, the same parts as in FIG. 11 are denoted by the same symbols, and explanations thereof will be omitted. In Figure 1, 1 is a DC power supply, 2-
2° is the input terminal, 3 is the transformer, and the primary windings 3a and 2
It has a secondary winding 3b, and a tertiary winding 3C having more turns than the primary winding 3a, 4 is a switch means, 12 is a rectifying and smoothing circuit, and 13 is an output detection 14-14'' is an output terminal, 15 is a load. 5 is a diode, the anode of which is connected to the connection point between the switch means 4 and the input terminal 2'', and the cathode connected to the connection point between the switch means 4 and the input terminal 2''. It is connected to the connection point of the primary winding 3a, so that the current regenerated to the DC power source 1 flows even when the switch means 4 is off. Reference numeral 6 denotes an auxiliary switch means, one end of which is connected to the primary winding 3a. 3a and the switch means 4, and the other end is connected to the connection point between the input terminal 2 and the primary winding 3a through a series connection circuit of the current limiting means 9, the diode 8, and the tertiary winding 3c. The second on/off signal of the control circuit 11 is applied to the control terminal, which repeats on/off operations to turn on and off the current flowing from the tertiary winding 3C to the primary winding 3a via the diode 8 and current limiting means 9. 8 is a diode, the anode of which is connected to the tertiary winding 3C, the cathode of which is connected to the current limiting means 9, and the diode is connected to allow current to flow when a flyback voltage is induced in the tertiary winding 3c. 9 is a current limiting means,
The current flowing from the tertiary winding 3C to the primary winding 3a is limited. 7 is a capacitor, and 10 is a resistor. The capacitor 7 and the resistor 10 are connected in parallel, and are connected to the connection point between the diode 8 and the current limiting means 9 and both ends of the input terminal 2. A reset circuit composed of a resistor 10 is used to maintain the flyback voltage induced across the tertiary winding 3C and reset the magnetic flux of the transformer 3. Reference numeral 11 denotes a control circuit 11, which applies first and second on/off signals to the control terminals of the switch means 4 and the auxiliary switch means 6, respectively, and receives a signal from the output detection circuit 13 and operates the switch so that the output voltage is constant. The pulse width of the first on/off signal applied to the means 4 is controlled, and the second on/off signal applied to the auxiliary switch means 6 is also controlled so that the auxiliary switch means 6 is turned on only during the off period of the switch means 4. do.

The operation of the switching power supply device configured as described above will be described in detail with reference to FIGS. 2 and 3. FIG. 2 1al to +e) show the operating waveforms of each part of the switch means 4 and the auxiliary switch means 6, and (a)
is the switching voltage waveform VOS applied across the switch means 4, and (b) is the switching voltage waveform VOS applied across the switch means 4 and the diode '
- The switching current waveform Io flowing in the node 5 is (C
l is the control circuit 1 applied to the control terminal of the switch means 4;
1, the first on/off signal V c + of (d
) is the switching current waveform IQ flowing through the auxiliary switch means 6, and (e) shows the second on/off signal VG2 of the control circuit 11 applied to the control terminal of the auxiliary switch means 6, and the time of each waveform state is t1~ to show the change in
The symbol t is written on the waveform. In the switching current waveform Io flowing through the switch means 4, it can be seen that a negative current flows during the period immediately before the switch means 4 is turned on (period t2 to t3), resulting in zero-cross switching. Next, a detailed explanation of the zero-cross switching will be given with reference to FIG. FIG. 3 (al to fcl is an explanatory diagram showing the equivalent circuit and the state of switching voltage and current in each waveform state. ) indicates the status of
(bl is when the switch means 4 is turned off and the auxiliary switch means 6
indicates the state during the on period (1+ to t2 period), and (Cl
shows the state during the period (period t2 to t3) in which the auxiliary switch means 6 is off immediately before the switch means 4 is turned on. In FIG. 3, 19 equivalently represents the leakage inductance between the primary winding 3a and the secondary winding 3b, 40 equivalently represents the distributed capacitance of the transformer 3, and 41 represents the parasitic capacitance of the switch means 4. is shown equivalently. Third
Figure (al) shows a state in which the excitation current or current superimposed on the output current is flowing from the DC power supply 1 to the primary winding 3a. It is consumed by the reset circuit of diode 8, capacitor 7, and resistor 10 from the next winding 3C, and the transformer 3
A current 1'o whose magnetic flux is reset, a flyback voltage value v3c induced in the tertiary winding 3C by turning on the auxiliary switch means 6, and a flyback voltage value v3 induced in the primary winding 3a. ．． The voltage difference V3c-v3s
is applied to the current limiting means 9, and the current Io limited by the current limiting means 9 is applied to the tertiary winding 3c, the diode 8, the current limiting means 9, the auxiliary switch means 6, the primary winding 3a, and the leakage inductance. 19 shows how current is circulating (hereinafter referred to as circulating current).

This circulating current flows through the primary winding 3a of the transformer 3.
The transformer 3
Since the current flows to excite only the leakage inductance 19 without energizing the transformer 3 and without affecting the reset operation of the transformer 3, excitation is accumulated only in the leakage inductance 19. Here, Lσ is the inductance value of the reincarnation inductance 19, and ro is the circulating current value. Next, as shown in FIG. 3 (Cl), when the auxiliary switch 6 is turned off, the circulating current is cut off, so the excitation energy accumulated in the leakage inductance 19 is transferred to the parasitic capacitance 4l of the switch means 4 that is turned off. Since a current flows through the distributed capacitance 40 of the transformer 3 to regenerate energy in the DC power supply 1, the voltage across the switch means 4 becomes zero, and then the switch means 4 is turned on, resulting in zero cross switching. The conditions for such zero-cross switching to be possible include the following relationship between the energy stored in the leakage inductance 19 and the stored energy due to the voltage applied during the off period of the switch means 4 of the distributed capacitance 40 and parasitic capacitance 41. Since all of this energy is regenerated to the DC power supply 1, no loss occurs.

Here, C40 is the equivalent capacitance value of the distributed capacitance 40, and C40 is the equivalent capacitance value of the distributed capacitance 40.
41 is the equivalent capacitance value of the parasitic capacitance 41, and Vos is the voltage value applied to both ends of the switch means 4 during the off period. This means that zero-cross switching is achieved as long as the circulating current 1o has a value that satisfies the above relational expression, regardless of the on/off time width of the switch means 4 that is varied to control the output voltage and the magnitude of the output current. This shows that the circulating current 1o is a value determined only by the current limiting means 9 if there is an induced voltage difference between the primary winding 3a and the tertiary winding 3C, so that stable zero-cross switching is always possible. Can be secured. Furthermore, if the circulating current Io is constant as shown in FIG. It is also possible to reduce the losses that occur. Moreover, the reason why there is a slight delay between when the auxiliary switch means 6 is turned off and when the switch means 4 is turned on (period t2 to t3) is because the switch means 4
A certain amount of time is necessary to obtain the charging/discharging time of the distributed capacitance 40 and the parasitic capacitance 41 until the voltage across the terminal becomes zero voltage, and the value is t3t2#-! ! −Lcr(
Cno+C<+) t) Sa, 2 This includes the re-cage inductance 19 and the above distribution. In FIG. 1, the polarity of the secondary winding 3b and the rectifying and smoothing circuit 12 operate in the same manner regardless of the so-called flyback sliding configuration or feedforward configuration.

As described above, according to this embodiment, the circulating current is passed from the tertiary winding 3C through the diode 8, the current limiting means 9, the auxiliary switch means 6, and the primary winding 3a during the off period of the winch means 4. By configuring the current to flow, it is possible to prevent a spike current when the switch means 4 is turned on, and also to achieve zero-cross switching, and furthermore, the excitation of the primary winding 3a of the transformer 3 that occurs when the switch means 4 is turned off can be prevented. Spike voltage generated by energy and leakage inductance 19 is transferred to the distributed capacitor 4.
0. By actively absorbing it into the parasitic capacitance 41 or by connecting a capacitor externally and equivalently increasing the distributed capacitance 40 and parasitic capacitance 41, it is possible to absorb even more and prevent a steep rise in the switching voltage waveform. Zero-cross switching can also be used for turn-off without loss, resulting in low noise, low loss, high efficiency, and high frequency. Note that the loss due to the circulating current is
These are losses due to the winding resistances of a and the tertiary winding 3C, losses in the current-limiting means 9, and switching losses in the auxiliary switch means 6, but by designing the circulating current to be small, the losses do not become too large.

A second embodiment of the present invention will be described below with reference to the drawings.

FIG. 4 shows a circuit configuration diagram of a switching power supply device showing a second embodiment of the present invention, which is a so-called voltage resonance type power supply. In FIG. 4, the same parts as in FIG. 1 are denoted by the same reference numerals, and the description thereof will be omitted. In Fig. 4, 1 is a DC power supply, 2-2'' is an input terminal, 3 is a transformer, 3a is a primary winding, 3b is a secondary winding, 3
C is a tertiary winding, 4 is a switching means, 5 is a diode, 6 is an auxiliary switch means, 8 is a diode, 9 is a current limiting means, and 11 is a control circuit. , 12 is a rectifying and smoothing circuit, 13 is an output detection circuit, 14-14' is an output terminal, and 15
is the load. 19 is a leakage inductance or inductance element between the primary winding 3a and the secondary winding 3b of the transformer 3, and is connected in series to the connection point between the input terminal 2 and the tertiary winding 3C and one end of the primary winding 3a. connected to. 1
6 is a capacitor, one end of which is connected to the input terminal 2, the other end of which is connected to the connection point of the primary winding 3a, the switch means 4 and the auxiliary switch means 6, and the capacitor 16 and the re-cage inductance 19 or the A resonant circuit is configured with the primary winding 3a.

The operation of the switching power supply device configured as described above will be explained in detail with reference to FIG. 5 as well.

FIG. 5 (al to let shows the operating waveforms of each part of the switch means 4 and the auxiliary switch means 6, (a) is the switching voltage waveform VDS applied to both ends of the switch means 4, and tb) is the switch means 4 and diode 5
(C) shows the first on/off signal vG1 of the control circuit 11 applied to the control terminal of the switch means 4, and {d} shows the switching current waveform Io flowing through the auxiliary switch means 6. is the waveform To, tel indicates the second on/off signal VG2 of the control circuit 11 applied to the control terminal of the auxiliary switch means 6, and the symbol j+4j< is written to indicate the temporal change of each waveform state, Furthermore, the solid line in the waveform of FIG. , it can be seen that a negative current flows during the period immediately before the switch means 4 is turned on (period 12 to t3), resulting in zero-cross switching.During the on period of the switch means 4 (period t3 to t), the transformer 3 and Excitation energy is stored in the cage inductance 19, and during the off period (t+-t2), the stored energy in the leakage inductance 19 or the transformer 3 is released to the capacitor 16, and the inductance value Ll9 of the leakage inductance 19 or the inductance value of the primary winding is The resonant frequency f c = 1/2πV determined by Lp and the capacitance value CI6 of the capacitor 16, or rc = 1/2vτpc16-, is applied to both ends of the capacitor 6 and the switch means 4. Furthermore, the sinusoidally oscillating voltage is
It oscillates around the input voltage of the DC power supply 1 or the sum of the input voltage and the flyback voltage generated in the primary winding 3a. At the same time, the off period of the switching means 4 (
1+ to t2 period), as already explained in detail using FIGS. 1 and 3, due to the voltage difference between the sinusoidal flyback voltages induced in the primary winding 3a and the tertiary winding 3c, From the tertiary winding 3c to the diode 8, the current limiting means 9, and the auxiliary switch means 6 which is turned on. A current determined by the current limiting means 9 flows through the leakage inductance 19 as a circulating current through the primary winding 3a, and excitation energy is accumulated in the leakage inductance 19. Next, the circulating current is cut off by turning off the auxiliary switch 6 (time t2), so that the energy accumulated in the leakage inductance 19 is transferred from the leakage inductance 19 to the capacitor 16 and the primary winding in the direction of discharging the capacitor 16. It flows through the line 3a and works to bring the voltage across the switch means 4 which is off to zero voltage, and when the voltage reaches zero, the remaining energy is regenerated to the DC power supply 1 without loss, and at the same time the switch means 4 is turned off. By turning on (time t3), zero cross switching occurs. Such operation is affected even if the amplitude of the sinusoidal flyback voltage induced in the primary winding 3a is small due to a light load on the output current or a change in the on period of the switch means 4 due to output voltage control. Zero-cross turn-on can be achieved reliably without causing any deer damage, and zero-cross turn-on is possible even when the switching means 4 is turned on at a time when the amplitude of the sinusoidal flypack voltage is not necessarily at its minimum. It can be seen that there is no need to use special synchronization means. In FIG. 4, the secondary volume 1
The polarity of 13b and the rectifying and smoothing circuit 12 operate in the same manner in any configuration such as a so-called flypack configuration or a feedforward configuration.

Furthermore, Fig. 5(d). If the circulating current is constant as shown by +el, the on period of the auxiliary switch means 6 is set to a short time before the switch means 4 is turned on, as shown by the dotted line, thereby reducing the loss occurring in the current limiting means 9. You can also do it. Furthermore, by designing the amplitude of the resonant voltage to be small in advance, it is also possible to lower the withstand voltage of the switch means 4.

As described above, according to this embodiment, during the OFF period of the switch means 4, the circuit circulates from the tertiary winding 3C to the diode 8, the current limiting means 9, the auxiliary switch means 6, the primary winding 3a, and the leakage inductance 19. By configuring it to allow current to flow, zero-cross switching can always be ensured without loss even when the amplitude of the resonant voltage changes due to light loads, input fluctuations, output control, etc., and the output voltage can be controlled over a wide range with low noise. It is possible to create a highly reliable voltage resonant power supply.

A third embodiment of the present invention will be described below with reference to the drawings.

FIG. 6 shows a circuit configuration diagram of a switching power supply device showing a third embodiment of the present invention, which is a so-called current resonance type power supply. In FIG. 6, the same parts as in FIG. 1 are denoted by the same reference numerals, and explanations thereof will be omitted. In FIG. 6, 1 is a DC power supply, 2-2' is an input terminal, 3 is a transformer, 3a is a primary winding, 3b is a secondary winding, 3
c is a tertiary winding, 4 is a switch means, 5 is a diode, 6 is an auxiliary switch means, 7 is a capacitor, 8 is a diode, 9 is a current limiting means, 10 is a resistor, 11 is a control circuit, 12 is a rectifying and smoothing circuit, 13 is an output detection circuit, 14-14' is an output terminal, and 15 is a load. Reference numeral 17 denotes a secondary side leakage inductance or inductance element between the primary winding 3a and the secondary winding 30 of the transformer 3, and is connected in series between the secondary winding 3b and the rectifying and smoothing circuit 12. A capacitor 18 is connected to both ends of the secondary winding 3b via the secondary leakage inductance 17, and the secondary leakage inductance 17 and the capacitor 18 form a resonant circuit.

The operation of the switching power supply device configured as described above will be explained in detail with reference to FIG. 7 as well. Figure 7 (a
)~tel is the switch means 4 and the auxiliary switch means 6
, (al is the switching voltage waveform Vo≦ applied to both ends of the switch means 4, +b) is the switching current waveform 1o flowing through the switch means 4 and the diode 5, and (Cl shows the first on/off signal v6l of the control circuit 11 applied to the control terminal of the switch means 4, (d) shows the switching current waveform 1o flowing through the auxiliary switch means 6, and (e) shows the switching current waveform 1o flowing through the auxiliary switch means 6. Control circuit 1 applied to the control terminal of 6
1, and symbols t1 to t are written on the waveforms to indicate temporal changes in each waveform state. The switching current waveform 1o flowing through the switch means 4 continues during the period (t2) immediately before the switch means 4 is turned on.
It can be seen that a negative current flows during the period t3 to t3, resulting in zero cross switching. The on period of the switch means 4 (
The inductance value L+? of the secondary side leakage inductance 17 that constitutes a resonant circuit due to the voltage induced across the secondary winding 3b during the period t3 to t4)? and capacitor 1
A current that oscillates in a sinusoidal manner flows at a resonance frequency fc=1/2πψ=κ determined by the capacitance value CI8 of 8. As a result, the switching current waveform 1o flowing through the switch means 4 also oscillates in a sinusoidal manner, and especially when the sinusoidal current rises from zero and becomes negative, the switch means 4
It is possible to perform zero cross switching by turning off. Furthermore, the off period (t+-
t2 period), the flypack voltage generated by the excitation energy of the transformer 3 is already applied to the tertiary winding 3C, as explained in detail in FIGS. 1 and 3 (Cl).
At the same time, the magnetic flux consumed by the reset circuit, which is a circuit connected in parallel with the diode 8, the capacitor 7, and the resistor 10, is reset, and at the same time, due to the voltage difference between the flypack voltages induced in the primary winding 3a and the tertiary winding 3c, A current determined by the current limiting means 9 flows as a circulating current from the tertiary winding 3c through the diode 8, the current limiting means 9, the auxiliary switch means 6 which is turned on, and the primary winding 3a, and the current flows through the primary winding. Excitation energy is stored in the reincarnation inductance 3a. Next, the auxiliary switch 6 turns off (
t2 time), the circulating current is cut off, and the energy accumulated in the leakage inductance is transferred to the parasitic capacitance of the switch means 4 which is turned off and the transformer 3.
Since a current is passed through the DC power supply 1 so as to regenerate energy through the distributed capacitance of This makes it possible to prevent zero cross switching. Here, the seventh
If the circulating current Io is constant as shown in the figure, the on period of the auxiliary switch means 6 is set to a short period before the switch means 4 is turned on as shown by the dotted line, thereby reducing the loss generated in the current limiting means 9. It can also be decreased.

As described above, according to this embodiment, the circulating current is passed from the tertiary winding 3C through the diode 8, the current limiting means 9, the auxiliary switch means 6, and the primary winding 3a during the off period of the switch means 4. By configuring the current to flow, it is possible to prevent a spike current when the switch means 4 is turned on, to prevent loss from occurring when the switch means 4 is turned on, and to reduce the amount of the primary current of the transformer 3 that occurs when the switch means 4 is turned off. By actively absorbing the spike voltage generated by the excitation energy of the winding 3a and the leakage inductance 19 into the distributed capacitance and parasitic capacitance, or by connecting a capacitor externally to equivalently increase the distributed capacitance and parasitic capacitance. By absorbing even more, it is possible to prevent the steep rise of the switching voltage waveform, reduce the occurrence of noise in the voltage waveform, and achieve even lower noise. A high-efficiency current resonance type power supply without increase can be achieved.

A fourth embodiment of the present invention will be described below with reference to the drawings.

FIG. 8 shows a circuit configuration diagram of a switching power supply device showing a fourth embodiment of the present invention, in which an inductance element is used as current limiting means. In FIG. 8, the same parts as in FIG. 1 are denoted by the same reference numerals, and explanations thereof will be omitted. In Fig. 8, 1 is a DC power supply, 2-2° is an input terminal, 3 is a transformer, 3a is a primary winding, 3b is a secondary winding, and 3c is a tertiary winding. It is a winding wire,
4 is a switch means, 5 is a diode, 6 is an auxiliary switch means, 7 is a capacitor, 8 is a diode, 9 is a current limiting means, 10 is a resistor, 11 is a control 12 is a rectifying and smoothing circuit, 13 is an output detection circuit, 14-14° is an output terminal, and 15 is a load. 20 is an inductance element, which is used as current limiting means 9. 21
is a capacitor, one end of which is connected to the input terminal 2, the other end of which is connected to the connection point between the auxiliary switch means 6 and the inductance element 20, and absorbs the excitation energy of the inductance element 20 when the auxiliary switch means 6 is turned off. .

Regarding the switching power supply device configured as described above, the current limiting operation when an inductance element is used as the current limiting means will be described in detail with reference to FIG. 9 as well. 9(a) to (e) show operating waveforms of each part of the switch means 4 and the auxiliary switch means 6, (a) is the switching voltage waveform VDS applied to both ends of the switch means 4, and (b) ) is the switching current waveform Io flowing through the switch means 4 and the diode 5, (C) shows the first on/off signal VGI of the control circuit 11 applied to the control terminal of the switch means 4, and (d) shows the switching current waveform Io flowing through the switch means 4 and the diode 5. A switching current waveform Io flowing through the auxiliary switch means 6, let indicates the second on/off signal Vc2 of the control circuit 11 applied to the control terminal of the auxiliary switch means 6, and shows temporal changes in each waveform state. Therefore, symbols t1 to t4 are written on the waveform.
+~t2 period), when the auxiliary switch 6 is turned on, the above 1
Voltage difference v3d-v3 between the flyback voltages induced in the secondary winding 3a and the tertiary winding 3C. is applied across the inductance element 20, and a circulating current IQ flows from the tertiary winding 3c through the diode 8, the inductance element 20, the turned-on auxiliary switch means 6, and the primary winding 3a as shown below. , V3e is the induced flyback voltage value of the tertiary winding 3c, V3m is the induced flyback voltage value of the primary winding 3a, L20 is the inductance value of the inductance element 20, and T is the auxiliary switch. This is the on time of means 6. In this case, the limit value of the circulation and current is determined by adjusting the ON time of the auxiliary switch means 6 as shown in the solid line and dotted line in FIG. When the circulating current is cut off, the excitation energy accumulated in the inductance element 20 is once absorbed by the capacitor 21 and transferred to the stored resistance 1.
However, the loss due to this is about half that of the current limiting means that limits the current at a constant current.

As described above, by using the inductance element 20 as the current limiting means 9, the loss of the current limiting means 9 is halved, and the current waveform flowing through the auxiliary switch means 6 is shaped into a triangular wave, so that the loss during turn-on and the loss during conduction are reduced by half. This makes it possible to reduce the amount of electricity and further improve efficiency.

A fifth embodiment of the present invention will be described below with reference to the drawings.

FIG. 10 shows a circuit configuration diagram of a switching power supply device showing a fifth embodiment of the present invention, in which leakage inductance of the primary winding 3a and the tertiary winding 3c of the transformer 3 is used as current limiting means. It was used. In FIG. 10, the same parts as in FIG. 8 are denoted by the same reference numerals, and the explanation will be omitted. In Figure 10, 1 is a DC power supply, 2-2° is an input terminal, 3 is a transformer, 3a is a primary winding, 3b is a secondary winding, and 3C is a tertiary winding. 4 is a switch means, 5 is a diode, and 6 is a winding.
is an auxiliary switch means, 8 is a diode, and 1
1 is a control circuit, 12 is a rectification and smoothing circuit, and 13
is an output detection circuit, 14-14° is an output terminal, 15 is a load, and 21 is a capacitor. 25 equivalently represents the leakage inductance of the primary winding 3a and the tertiary winding 3C, and this inductance value is used as a current limiting means for the circulating current. 22 is a diode, 23 is a capacitor, and 24 is a
A resistor connected between the primary winding 3a via a parallel circuit of a diode 22, a capacitor 23, and a resistor 24,
It absorbs and consumes the excitation energy of the transformer 3 to reset the magnetic flux.

The operation of the switching power supply device configured as described above is almost the same as that already explained in detail in FIG. 8, and the explanation will be omitted.
The inductance value of No. 5 can be set freely to some extent by changing the coupling between the primary winding 3a and the tertiary winding 3C, and has almost no effect on the coupling between the primary winding 3a and the secondary winding 3b. There are no operational problems. Further, the circulating current flows through the tertiary winding 3c, the diode 8, the auxiliary switch means 6, and the primary winding 3a, but the excitation energy of the leakage inductance 25 when the auxiliary switch means 6 is turned off is , is absorbed and stored in the capacitor 21, is absorbed in the reset circuit during the next ON period of the auxiliary switch means 6, and is consumed by the resistor 24.

As described above, by using the leakage inductance 25 as the current limiting means, the number of parts can be reduced, the circuit configuration can be simplified, and costs can be reduced.

Note that although the fourth and fifth embodiments are applied to the circuit configuration of the first embodiment, the same result can be obtained if similarly applied to the second and third embodiments. Needless to say, the effect can be obtained.

Also, the first embodiment. In the third, fourth, and fifth embodiments, the circuit for resetting the excitation of the transformer 3 during the OFF period of the switch means 4 is constructed using a diode, a resistor,
Although a power consumption type using a capacitor is used, it goes without saying that similar effects can be obtained by using other reset circuits. Further, it goes without saying that the present invention can be similarly applied to a switching power supply circuit configured with a multi-hole type.

Effects of the Invention As described above, the present invention provides three windings in parallel with the primary winding of the transformer.
The next winding is connected through a series connection circuit of the rectifying means, the current limiting means, and the second switching means, and the second switching means is turned on and off complementary to the first switching means, and the second switching means is turned on and off in a complementary manner to the first switching means. By causing current to circulate between the tertiary winding and the primary winding when the means is turned on, zero-cross switching in which the voltage across the first switch means is turned on at zero is constantly and stably realized; It is possible to realize an excellent switching power supply device that has low noise, low loss, high efficiency, high frequency, high reliability, and a wide control range.

[Brief explanation of drawings]

FIG. 1 is a circuit configuration diagram of a switching power supply device according to a first embodiment of the present invention, FIG. 2 is an operation waveform diagram of the circuit configuration diagram of FIG. 1 of the present invention, and FIG. FIG. 4 is a circuit diagram showing a switching power supply according to the second embodiment of the present invention. FIG. 5 is an operation waveform diagram of the circuit diagram of FIG. 6 is a circuit configuration diagram showing a switching power supply device according to the third embodiment of the present invention, FIG. 7 is an operation waveform diagram of the circuit configuration diagram of FIG. 6 according to the present invention, and FIG. A circuit configuration diagram showing a switching power supply device in an embodiment, FIG. 9 is an eighth embodiment of the present invention.
The operating waveform diagram of the circuit configuration diagram shown in FIG.
FIG. 11 is a circuit diagram of a conventional switching power supply, FIG. 12 is an operating waveform diagram of the conventional circuit diagram of FIG. 11, and FIG. 13 is a diagram of a conventional switching power supply. An explanatory diagram of the circuit configuration diagram in FIG. 11,
FIG. 14 is a circuit configuration diagram of another conventional switching power supply device, FIG. 15 is an operating waveform diagram of the conventional circuit configuration diagram of FIG. 14, and FIG. 16 is a circuit configuration diagram of another conventional switching power supply device. FIG. 17 is an operational waveform diagram of the conventional circuit configuration diagram of FIG. 16. 1...DC power supply, 2-2"...Input terminal, 3...Transformer, 4...Switch means, 5.8.22... ...Diode, 6...
... Auxiliary switch means, 716, 18, 21.23.
... Capacitor, 9 ... Current # limiting means,
10.24... Resistor, 11... Control circuit, 12... Rectifying and smoothing means, 13...
Output detection circuit, 14-14'...Output terminal, 1
5... Load, 17, 19.25... Leakage inductance, 20... Inductance element.

Claims (5)

[Claims] (1) A transformer having at least a primary winding, a secondary winding, and a tertiary winding; a first switch means connected to one end of the primary winding for repeating on/off; a DC input voltage for supplying a voltage via a first switch means; an output terminal for supplying a DC output voltage via a rectification and smoothing means connected to the secondary winding; and a rectification means for the tertiary winding. are connected in parallel to both ends of the primary winding through a series connection of a current limiting means and a second switch means, and the second switch means is complementary to the first switch means for turning on and off. Repeatedly, the tertiary winding has a larger number of turns than the primary winding, and when the second switch means is turned on, the current limiting means receives an induced voltage in the primary winding and the tertiary winding. A switching power supply configured to apply a difference of . (2) a transformer having at least a primary winding, a secondary winding, and a tertiary winding; a first switch means connected to one end of the primary winding for repeatedly turning on and off; A resonant circuit is formed with the DC input voltage supplied through the first switch means and the primary winding or the leakage inductance or inductance element of the primary winding during the period when the first switch means is off. a capacitive element connected in parallel to the primary winding or the first switch means; and an output terminal for supplying a DC output voltage via a rectifying and smoothing means connected to the secondary winding. , the tertiary winding is connected in parallel to both ends of the primary winding through a series connection of a rectifying means, a current limiting means, and a second switch means, and the second switch means is connected to the first switch means. The switching means repeats on and off in a complementary manner, and the tertiary winding has a larger number of windings than the primary winding, and when the second switch means is turned on, the current limiting means is connected to the primary winding. A switching power supply device configured to apply a difference between an induced voltage between a tertiary winding and a tertiary winding. (3) a transformer having at least a primary winding, a secondary winding, and a tertiary winding; a first switch means connected to one end of the primary winding to repeatedly turn on and off; a DC input voltage for supplying a voltage via a first switch means; an output terminal for supplying a DC output voltage via a rectifying and smoothing means connected to the secondary winding; and the first switch means is turned on. a capacitive element connected to the secondary winding so as to constitute a resonant circuit with the leakage inductance or inductance element of the secondary winding during the period;
The secondary winding is connected in parallel to both ends of the primary winding through a series connection of rectifier means, current limiting means and second switch means, and the second switch means is connected to the first switch means. repeats on and off in a complementary manner, and the tertiary winding has a larger number of turns than the primary winding, and when the second switch means is turned on, the current limiting means has the primary winding and the tertiary winding. A switching power supply device configured to apply a difference in induced voltage between windings. (4) The switching power supply device according to claim 1, 2 or 3, wherein an inductance element is used as the current limiting means. (5) The switching power supply device according to claim 1, wherein a leakage inductance between the primary winding and the tertiary winding is used as the current limiting means.