JPH0198009A - Automatic power factor controller - Google Patents

Abstract

PURPOSE:To obtain a power saving type automatic power factor controller by using an auto-transformer type voltage regulator, a DC exciting power supply, a semiconductor switch, a power factor detecting circuit, and a switch control circuit. CONSTITUTION:The output wiring of an auto-transformer type voltage regulator 24 is connected between an AC power supply 12 and an inductive load 18 and the output voltage is controlled by a control wiring 26. An exciting current is supplied to the wiring 26 from a DC exciting power supply 28 via a semiconductor switch circuit 30. A power factor detecting circuit 32 outputs an output signal proportional to the power factor of the load 18. A control circuit 34 controls the conduction rate of a switch element 88 of the circuit 30 in response to said output signal and then controls the excitation of the wiring 26. In such a constitution, the load voltage is controlled at an optimum level in response to the output signal proportional to the power factor. As a result, the load 18 is always driven at the maximum power factor and the high energy saving effect is ensured.

Description

DETAILED DESCRIPTION OF THE INVENTION [Object of the Invention] The present invention relates to a power factor control device, and more particularly to an automatic power factor control device for an AC induction motor.

[Prior art]

Conventionally, for the purpose of saving energy in AC induction motors and other inductive loads, U.S. Patent Section 4,052,6
No. 48 and No. 4,337,640 propose changing the input voltage of an induction motor by phase control to improve the power factor.

In these power factor control devices, the thyristor directly controls the phase of the AC voltage supplied to the load, so the load current contains many harmonic components, and this harmonic current is connected to the power capacitor of the power factor control device and the linear flows into the reactor,
These devices caused problems such as abnormal noise, vibration, overheating, and damage. Moreover, the waveform of the received power supply voltage is distorted by the harmonic current, causing great trouble to information devices such as computers and other control devices. The thyristor is fired in synchronization with the voltage in every cycle, but the synchronization signal for firing the thyristor is taken from the power supply voltage.

The synchronization signal sometimes fluctuates due to this waveform distortion. Therefore, depending on the state of the load, control becomes unstable, or in some cases becomes uncontrollable, resulting in safety and reliability problems. Aiming to resolve this, U.S. Patent Section 4,602,200
The issue proposed installing a harmonic filter, but this device required a large number of large-capacity capacitors, reactors, and resistors, making the entire device large and extremely expensive to manufacture. Next, when starting an induction motor or induction coil, a large starting current that is more than 6 times the rated current of the motor flows, so the capacity of the power semiconductor element is selected to be equivalent to 2 to 4 times the rated capacity of the inductive load. Therefore, the semiconductor element becomes expensive, and the control circuit therefor inevitably becomes large and complicated, and the response is poor.

Furthermore, a control device that also uses a thyristor as a semiconductor element has the disadvantage that a large internally generated loss occurs in the main circuit portion, resulting in increased power consumption in the main circuit portion. In particular, the main circuit requires a forced commutation circuit consisting of a commutation reactor and a commutation capacitor, and losses occur due to the energy transferred each time the commutation occurs within the commutation circuit.

In addition, power consumption was large due to losses in the main circuit snubber circuit (resistance, diode, etc.), losses in the smoothing reactor, AC reactor, etc. (iron loss, copper loss, etc.), and internal loss of the capacitor.

[Purpose of the invention]

Therefore, one object of the present invention is to provide an automatic power factor control device that solves the above problems.

Another object of the present invention is to provide an automatic power factor control device that is small, lightweight, and inexpensive.

Another object of the present invention is to provide an automatic power factor control device that can respond quickly to changes in the load of an AC induction motor.

Another object of the present invention is to provide an automatic power factor control device that prevents distortion to a sinusoidal AC waveform.

Another object of the present invention is to provide an automatic power factor control device that can automatically drive an inductive load at the highest power factor in response to the load condition of an AC induction motor.

Another object of the present invention is to provide an automatic power factor control device that is small, lightweight, and low cost.

Another object of the present invention is to provide an automatic power factor control device that has a large overload capacity, high stability and reliability, and requires no maintenance or inspection.

[Structure of the Invention] The automatic power factor control device of the present invention includes a single output winding connected between an AC power source and an inductive load, and a control winding for adjusting the output voltage of the output winding. A winding transformer type voltage regulator, a DC excitation power supply that supplies a DC excitation current to the control W winding, and a DC excitation power supply connected between the control winding and the DC excitation power supply and supplied to the control winding. a semiconductor switch for controlling the DC excitation current; a power factor detection circuit for generating an output signal proportional to the power factor of the inductive load; and a power factor detection circuit for generating an output signal proportional to the power factor of the inductive load; The present invention is characterized by comprising a control circuit that controls the conduction rate of the semiconductor switch so that the conduction rate of the semiconductor switch is controlled.

(Example〕 Hereinafter, the present invention will be described in detail with reference to the drawings.

In FIG. 1, an automatic power factor control device 10 according to a preferred embodiment of the present invention has an input terminal 14 connected to an AC power source 12.
．． 16 and an output terminal 20.22 connected to the inductive load 18.
, an autotransformer type voltage regulator 24 equipped with a control winding 26 that adjusts the output voltage supplied to the inductive load 18 variably according to the power factor load state of the inductive load; A DC excitation power supply 28 that supplies excitation current and a control winding 2
6 and the DC excitation m source 28, and the control winding 26
a semiconductor switch circuit 30 that varies the DC excitation current supplied to the inductive load; a power factor detection circuit 32 that generates an output signal proportional to the power factor of the inductive load; and a conduction factor of the semiconductor switch circuit 30 in response to the output signal. and a control circuit 34 that controls the output voltage and adjusts the output voltage in response to the power factor.

In FIGS. 1 to 5, the autotransformer type voltage regulator 24 includes a first saturable iron core forming the main magnetic flux loop path and a magnetic shunt iron core 44 for bypassing a portion of the main magnetic flux loop path. The first saturable core 42 is made of a wound core. The first saturable iron core 42 includes an output winding consisting of a first series winding wA46, a first shunt winding 48, and a second series winding 50. The first series winding 46 is connected between the high voltage terminal connected to the output end 20 and the medium voltage terminal connected to the input end 14, and the first shunt winding 48 is connected to the first series winding 46 with the same polarity. 0 connected in series
The lower end of the shunt winding 48 has a neutral connected to the input terminal 16;
connected directly. The second series winding 81g50 is connected between the lower end of the shunt winding 48 and the output end 22 with a polarity opposite to that of the first series winding 46. In order to control the magnetic flux density of the magnetic shunt core 44 by changing the magnetic saturation state of at least a portion of the main magnetic flux loop path, a second saturable core 52 consisting of a wound core is controlled by the control winding 26 as described below. be done.

In FIGS. 2-3, the first saturable core 42 includes a center leg 54 and outer legs 56, 58 forming a main magnetic flux loop path. The center leg 54 includes a first core portion 54a and a second core portion 54b separated by the magnetic shunt core 44. Furthermore, the center leg 54 includes an extension portion extending outwardly from the outer legs 56, 58, ie, a third core portion 54c. The center leg 54 is disposed above the first saturable core 42 and the fixture 6
0.62 and are fixed to each other and integrated. Section 2.3.
As is clear from FIG. 5, the magnetic shunt core 44 is made up of a C-shaped cross-sectional core made by laminating a large number of silicon steel plates. Groove 44a of magnetic shunt core 44 is arranged to magnetically couple with center leg 54. End portion 44b of magnetic shunt core 44.

44c is an interlayer 64 having a required thickness corresponding to a constant air gap between the first coil block of the first series winding 46 and the shunt winding 48 and the second coil block of the second series winding 5o. , 66 on the outer legs 56, 58 of the first saturable core 42, and is fixed to the outer legs 56, 58 by fixtures 68, 70, so that each core is integrated. Magnetic shunt core 64 shunts a portion of the magnetic flux in the main flux loop path through high reluctance gaps (formed by intersperses 64 and 66) to outer legs 56 and 58 to provide an output voltage. It also functions to attenuate harmonics and reduce waveform distortion of the output voltage. The second saturable core 52 is the magnetic shunt core 5
4, i.e. the first coil block of the first series winding 46 and the shunt winding 48 and the second series winding 50.
The cores are arranged between the second coil block of the center leg 54 on the upper part of the second core part 54b and the lower end of the third core part 54c, and the respective cores are integrated by fixing devices 72 and 74. magnetically coupled. In this way, the second saturable core 52 is arranged so as to overlap the lower half of the first saturable core 42, and magnetically saturates a portion of the first saturable core to increase the magnetic flux of the second series winding 50. is the first series winding 46
The first series winding 46 and the shunt winding 48 have no effect on the magnetic flux of the shunt winding 48, and the magnetic flux of the first series winding 46 and the shunt winding 48 is shifted to the magnetic shunt core 44.

The first series winding [46 and the shunt winding 48. Second series winding 5
0 and the control winding 26 are each connected to the center leg 5.
The first to third core portions 54a, 54b, and 54c of No. 4 are wound around each other and are arranged substantially in the same plane. Furthermore, the two surfaces and the bottom surface of each winding are arranged so as to be aligned with the top surface of the second saturable iron core 52 and the bottom surface of the first saturable iron core 42, respectively. That is, a second coil block consisting of a coil block of the first series winding @ 46 and the shunt winding 48 and a second series winding 50, and a third coil block of the control winding 26 are arranged in the center leg 54, It is arranged substantially within the thickness of the first and second saturable cores 42.52. 3rd of center leg 54
The core portion 54c extends outside the first saturable core 42, and the control winding @26 is located at the lower end 54c of the center leg 54.
rolled up on top. The second saturable core 52 surrounds the second coil block of the second series winding 5o and the third coil block of the control winding 26. In FIGS. 2 and 3, the upper and lower parts of the second saturable core 52 are fixed by fixtures 72 and 74, respectively.
This constitutes an auxiliary magnetic flux loop path together with the center leg 54, and functions as a path for the magnetic flux of the control winding 26 when a DC excitation current is supplied to the control winding 26. That is, the magnetic flux of control winding 26 is
The core portion 54b is partially magnetically saturated, so that the magnetic flux of the first series winding @46 and the shunt winding 48 is transferred from the main magnetic flux loop through the magnetic shunt core 44 to the outer legs 56, 58.
shift to.

The first series winding 46 and the shunt winding 48 are connected to the center leg 5.
The second series winding 5o is wound on the second core 54b with a polarity opposite to that of the 1iIII series winding 46, so that the so-called Differentially coupled.

In the above configuration, when the input end 14.16 is connected to the AC power supply 12 and the output end 20.22 is connected to the inductive load 18, a large current flows through the first and second series windings 46°50, A difference current between the input current and the output current flows through the shunt winding 48 .

In FIGS. 1 and 2, when no DC excitation current is supplied to the control winding 26, the first series winding 46, the shunt winding 48, and the second series winding differentially coupled to the shunt winding 48 line 5
The magnetic flux generated by 0 passes from the center leg 54 to the outer legs 56 and 58, and
Cycle to 4.

At this time, since the magnetic flux generated by the first series winding 46 and the shunt winding 48 and the magnetic flux generated by the second series winding 50 are in opposite directions, the mutual magnetic flux as a whole.

It acts as a difference. Therefore, the output voltage at this time is the minimum.

Next, when the DC excitation current is supplied to the control winding 26,
The second saturable core 52 is the second saturable core 52 of the center leg 54. Because it is magnetically saturated together with the third core parts 54b and 54c,
The magnetic flux produced by first series winding 46 and shunt winding 48 is shifted to magnetic shunt core 44 . At this time, the magnetic flux circulates through the first core portion 54a, the outer legs 56, 58, and the magnetic shunt core 44, and the output voltage at the output ends 20, 22 becomes maximum. When the DC excitation current supplied to the control winding 26 is reduced, the output voltage at the output end of the output winding is reduced accordingly. By variably controlling the magnetic saturation states of the second and third core portions 54b and 54C of the center leg 54, the second By changing the differential coupling state of the series winding 5o to control the magnetic fluxes of the first series winding 46 and the shunt winding 48 shifted to the magnetic shunt core 44, the output voltage at the output end can be variably controlled.

Returning to FIG. 1, the DC excitation power supply 28 includes a current transformer 80 connected to the output side of the autotransformer type voltage regulator 24 and an AC reactor 82 connected via a transformer 81. The current transformer 80 functions as a current component circuit for extracting a component dependent on the current of the inductive load 18. The AC reactor 82 functions as a voltage component circuit for extracting a component dependent on the output voltage of the voltage regulator 24 from the voltage transformed from high voltage to low voltage via the transformer 81. Vector composition is performed on the side. The DC output current of the rectifier 84 corresponds to a rectified composite current of frequency components, is smoothed by a capacitor 86, and is used as a DC output current of the control winding 26. Due to current dependent components and voltage dependent components included in the DC output current of the rectifier 84.

When a load is turned on or off, or when the load suddenly fluctuates, it is possible to compensate for the load fluctuation with a high-speed response by changing the DC output current.

The semiconductor switch circuit 30 includes a semiconductor switch 88,
This semiconductor switch 88 is connected between the DC output terminals of the rectifier 84 in order to control the DC excitation current (2). As the semiconductor switch 88, a transistor or a thyristor can be used.

In FIG. 1, semiconductor switch 88 includes first and second control transistors 88a and 88b forming an inverted Darlington circuit.

Here, the inverted Darlington circuit is a PNP
In other words, in order to control the base current of the first control transistor 88a, the second control transistor 88b is connected in an inverted Darlington manner. It forms the Doderlington circuit. A current absorption circuit 90 is connected in parallel to the control winding 26 to which the DC excitation current I is supplied. A capacitor is used as the current absorption circuit 9o. This current absorption circuit 90 functions to absorb the left-handed flow of the DC output current of the rectifier 84 and the DC excitation current when the semiconductor switch 88 is off. !
! ! A voltage limiting element 92 is connected in parallel with the current absorption Ii7 circuit 88. This voltage limiting *'JJ element 92 is provided to conduct when the excitation voltage reaches the voltage limited by the voltage limiting element 92 and to prevent overvoltage from being applied to the semiconductor switch 88 and the current absorption circuit 9o. 1! An embodiment in which a constant voltage diode is used as the pressure limiting rope 92 is shown in FIG. In FIG. 1, the current absorption circuit 9
A backflow prevention diode 94 is inserted between the capacitor 0 and the semiconductor switch 88. Diode 94 prevents the discharge current from capacitor 9o from flowing through semiconductor switch 88 when semiconductor switch 88 is turned on. This prevents the risk of destruction of elements such as the illustrated transistor used as the semiconductor switch 88.

In FIG. 6.7, in the power factor detection circuit 32, a sine wave voltage signal (a) from a transformer (not shown) is supplied to an amplifier 100 consisting of an operational amplifier, and similarly a current transformer (not shown) is supplied to an amplifier 100 consisting of an operational amplifier. The sinusoidal current signal (b) from 1) is supplied to an amplifier 102 which is also an operational amplifier. amplifier 1
00° 102 has a large amplification factor and converts signals (a) and (b) into rectangular waves, respectively, and outputs signals (Q) and (d). The signals (Q) and (d) are then supplied to the N0RIEfl approximately 104, which outputs a pulse (e) equal to the phase difference (θ) of the signal ((1) and (d). This pulse (e) The signal is converted into a DC signal (f) through a low-pass filter 106 consisting of a capacitor and a capacitor.This DC signal (f) is supplied to a control circuit 34.

In FIG. 1, the control circuit 34 includes a transistor 108,
The output signal of the triangular wave oscillator 110 and the power factor detection circuit 32 (
f) and the triangular waveform output g of the triangular wave oscillator 110, and outputs drive pulses with different pulse widths to the base of the transistor 108. The collector of the transistor 108 is connected to the collector side of the transistor 88a via resistors R1 and R2, and the conduction rate of the semiconductor switch 80 is controlled by turning on/off the transistor 88b. This adjusts the excitation current of the control winding 26. In this case, the conduction rate control is based on the load voltage and the load f.
It is controlled by the control circuit 34 so as to eliminate the phase difference with the flow a, that is, to achieve the maximum power factor.

Next, the operation will be explained with reference to examples of voltage and current waveforms of each part shown in FIG.

The H path constant is selected so that even if the DC output current I of the rectifier 84 is high, it will be larger than the required value of the excitation current r' of the control winding @26. When the semiconductor switch 88 is on, the DC output current of the rectifier 84 is shunted by the semiconductor switch 88, and the exciting current ' decreases.
Next, when the semiconductor switch 88 is turned off, the rectifier output current flows into the control winding 26 in an increasing manner.

Due to the inductance of the control winding llI26, the exciting current I
Since ' can only increase gradually, the left-hand flow 1-I' flows into the current absorbing capacitor 9o. In this way,
The excitation current T' is instantaneously controlled by the base signal of the semiconductor switch 88 so as to be maintained at a target value.

The output signal f proportional to the power factor expressed by the phase difference between the load voltage and load current applied to the positive input terminal of the amplifier 112 and the triangular waveform signal g applied to the negative input terminal are compared, At the time of 1 hour t when the output pulse h occurs.

The amplifier 112 outputs a "1" signal, and at time t,
When the amplifier 112 outputs a "1" signal, the transistor 108 is turned on, and the transistors 88a and 88b are turned on.

The conduction rate α of the semiconductor switch 88 at a certain moment is defined by the on-time as Ton and the period as T.

It can be expressed as Ton α= □, and the average value of excitation a current I' av
is the average value Tav of the rectified leakage crab, and has the relationship I' av=a1 Iav. That is, when viewed as an average value, it can be seen that out of the rectifier output current I, only αIav flows for excitation, and the remaining (1-α)Iav flows to the semiconductor switch 88. In this way, the semiconductor switch 88 is turned on and off in response to the load condition detected by the load detection circuit 32, and is controlled so that the phase difference between the load voltage and the load current is always at zero level, that is, the power factor approaches 1. circuit 34
controlled by That is, when the phase difference θ between the load voltage and the load current is large, the power factor of the inductive load is extremely low, and the output f of the power factor detection circuit 32 becomes high. At this time, as is clear from FIG. 8, the output j of the transistor 108
Since the pulse width of the semiconductor switch 88 becomes larger, the conduction rate of the semiconductor switch 88 becomes larger, and the branch basin of the excitation current becomes larger. Therefore, the control current supplied to the control winding 26 is reduced, and the center voltage of the autotransformer type voltage regulator 24 is reduced.
The degree of magnetic saturation of the second core portion 54b of the leg 54 is reduced. At this time, the magnetic fluxes of the first series winding 46 and the shunt winding 48 in FIG. 2 are canceled by the opposite polarity magnetic flux of the second series winding 50, and the output voltage of the voltage regulator 24 decreases. Next, when the inductive load increases and the phase difference between the load voltage and the load current becomes smaller, the output f of the power factor detection circuit 32 becomes lower. At this time, the pulse width of the output of the amplifier IIa becomes smaller.

The conduction rate of the semiconductor switch 88 becomes small and the exciting current 1
' increases, and the output voltage of voltage regulator 24 increases. In this manner, the control circuit 34 controls the excitation current generator' by controlling the conduction rate of the semiconductor switch 88 in response to the output signal f of the power factor detection circuit 32, thereby controlling the induction current from the voltage regulator 24. The output voltage supplied to the load 18 has a power factor of 1.
Adjust so that

Although a single-phase embodiment of the present invention has been described above, the above-mentioned 11 autotransformer type voltage regulators can be connected to a three-phase AC power source by three-phase wiring. Power factor detection circuits are described in U.S. Pat. No. 3,588,710 and U.S. Pat. No. 4,480,219.
The phase detection circuit may also be constructed using the phase detection circuit disclosed in the above publication.

〔Effect of the invention〕

As is clear from the above, the automatic power factor control device according to the present invention provides significant effects.

(1) Since the load voltage is adjusted to the optimum level by quickly adjusting the output signal proportional to the power factor, the inductive load is always driven at the maximum power factor, resulting in a significant energy saving effect.

(2) The load voltage is controlled by controlling the excitation current flowing through the control winding of the autotransformer voltage regulator, and there is no direct phase control of the AC voltage in the power supply line, so the load current is caused by harmonics. It does not contain any components, does not distort the AC voltage waveform, and therefore does not cause any damage to information equipment such as computers or other control devices.

(3) Since the load current does not include harmonic components, a large and expensive high-capacity harmonic filter can be omitted, improving reliability and safety, as well as significantly reducing size and weight.

(4) The semiconductor switch does not directly control the AC voltage of the power line, but rather controls the low voltage, low current excitation current of the control winding of the autotransformer type voltage regulator.

It is possible to significantly reduce the capacity and cost of semiconductor switches and control circuits.

Also, circuit design becomes easier.

(5) Since an automatic power factor control device with a large load capacity can be controlled by an autotransformer type voltage regulator with a self-capacity of 1/100 or less, the entire device is smaller and lighter and does not generate large electromagnetic noise. First, because it is highly reliable, it can be used in spacecraft such as shuttles.

(6) High voltage, large capacity power factor control is possible by combining a low voltage, small capacity semiconductor switch and the control winding of an autotransformer type voltage Xa transformer, making it safe and reliable.
This has great practical effects, as it becomes possible to control the power factor of large-capacity inductive loads using extremely inexpensive electronic components, which was previously impossible.

(7) Enables the adoption of an autotransformer voltage regulator with a small self-capacity and a low-power control circuit for a large load capacity, minimizing energy loss, resulting in significantly higher efficiency.

Fig. 1 is a wiring diagram of a preferred embodiment of the automatic power factor control device according to the present invention, Fig. 2 is a plan view of the autotransformer type voltage regulator shown in Fig. 1, and Fig. 3 is a voltage regulator shown in Fig. 2. Side view of the vessel, No. 4
The figure is a bottom view of the voltage regulator in Figure 2, and Figure 5 is the v of Figure 2.
-V line cross-sectional view, FIG. 6 is a circuit diagram showing an example of the load detection circuit of FIG. 1.

7 shows a waveform diagram of the circuit shown in FIG. 6, and FIG. 8 shows a current voltage waveform diagram of the circuit shown in FIG. 1.

24... Autotransformer type voltage regulator 28...
......DC excitation power supply 30...Semiconductor switch circuit 32...
・・・・・・Power factor detection circuit 34・・・・・・・・・Control circuit Patent applicant: Alex Electronics Industry Co., Ltd.

Book 7 illustration

Claims (1)

[Claims] 1. (a) An autotransformer comprising an output winding connected between an AC power source and an inductive load, and a control winding for adjusting the output voltage of the output winding. (b) a DC excitation power source that supplies direct current and magnetic current to the control winding; (c
) connected between the control winding and the DC excitation power supply,
a semiconductor switch that controls the DC excitation current supplied to the control winding; (d) a power factor detection circuit that generates an output signal proportional to the power factor of the inductive load; and (e) responsive to the output signal. and a control circuit that controls the conduction factor of the semiconductor switch so that the power factor of the inductive load is 1. 2. The DC excitation power source is characterized by comprising a current transformer connected to the input side of the inductive load to take out a component depending on the current of the inductive load, and a rectifier connected to the current transformer. An automatic power factor control device according to claim 1. 3. An AC reactor to which the DC excitation power source is connected to the input side of the inductive load to take out a component dependent on the output voltage, and a current transformer to take out a component dependent on the current of the inductive load; The automatic power factor control device according to claim 1, further comprising a rectifier that rectifies the combined current to form the DC excitation current. 4. Claim 1 or 2, wherein the semiconductor switch is connected to a DC output terminal of the DC excitation power source, and a part of the DC excitation current is shunted to the semiconductor switch. Automatic power factor control device as described. 5. The automatic power factor control device according to claim 4, wherein a current absorption circuit is connected in parallel to the semiconductor switch. 6. The automatic power factor control device according to claim 5, characterized in that a voltage limiting element is connected in parallel to the semiconductor switch. 7. The automatic power factor control device according to claim 1 or 2, wherein the power factor detection circuit includes a phase difference detection circuit that detects a phase difference between a load voltage and a load current. 8. The control circuit includes an amplifier that generates an output pulse with a pulse width responsive to the output signal, and a transistor that controls the conduction rate of the semiconductor switch in response to the output of the amplifier. An automatic power factor control device according to claim 1 or 2. 9. The autotransformer type voltage regulator includes a first series winding, a shunt winding connected in series to the first series winding, and a shunt winding having a polarity different from that of the shunt winding. a main flux loop path comprising a first saturable iron core having a second series winding connected in series with the line; and a main flux loop path disposed between the shunt winding and the second series winding to at least one magnetic shunt core with an air gap for bypassing a portion of the loop path; and a portion of the first saturable core between the magnetic shunt core and the second series winding. and an auxiliary magnetic flux loop path consisting of a second saturable core magnetically coupled to the control winding, the control winding magnetically saturating the part of the first saturable core via the second saturable core. 3. The automatic power factor control device according to claim 1, wherein the magnetic flux of the first series winding and the shunt winding is shifted to the magnetic shunt core. 10. The first saturable core includes a first-turn core having a center leg and an outer leg, and the auxiliary flux loop path surrounds the second series winding and the control winding on the center leg. a second-volume core arranged as shown in FIG. An automatic power factor control device according to claim 9. 11. A first fixture for fixing the first volume core and the center leg, and a first fixture for fixing the first volume core and the center leg;
11. The automatic power factor control device according to claim 10, further comprising a second fixture for fixing the leg. 12. The first volume iron core is arranged on one side of the center leg, and the second volume iron core is arranged on the other side of the center leg. 12. Automatic power factor control device according to item 11. 13. Claim 12, characterized in that the first series winding, the shunt winding, the second series winding, and the control winding are arranged in substantially the same plane. Automatic power factor control device as described.