JP7486704B2 - Permanent magnet brushless motor - Google Patents

Description

The present invention relates to a permanent magnet field type brushless rotating electric machine.

Permanent magnet field brushless rotating electric machines have been widely used in recent years because they do not have brushes that cause friction like DC motors. Most of these types of rotating electric machines are generally coreless, but while coreless types do not have cogging torque, they have the problem that it is difficult to obtain high torque and large output. On the other hand, core-equipped types can easily obtain high torque and large output, but have the disadvantage that the cogging torque is large because they have slots or teeth.

1) Non-patent literature, New Brushless Motor, page 117, Hisashi Kenjo, Shigenobu Nagamori, Sogo Electronics Publishers.
2) Non-patent literature, Reduction of cogging torque of PMSM by slot combination, Hideo Doumeki, Yudai Shoji, Institute of Electrical Engineers Rotating Machine Research Society, RM-04-15, 1916.
3) Non-patent document, Improvement of Characteristics of PM Type Stepping Motors, Masafumi Sakamoto, Japan Management Association, Motor Technology Symposium, 1983.
On the other hand, the related prior art includes the above-mentioned documents 1) and 2). Also, the related technical information includes document 3). The above documents will be abbreviated as reference 1), reference 2), and reference 3) below, respectively.

The present invention is directed to a permanent magnet field type brushless rotating electric machine having a stator core and multiple concentrated winding poles, and aims to realize a highly efficient, low vibration brushless rotating electric machine that keeps cogging torque small. A known method for reducing cogging torque is to maintain the relationship between the total number of winding poles Q of the stator and the least common multiple LCM of the rotor poles p and Q.
Another objective of the present invention is to create a winding method that keeps the fundamental wave component of the interlinkage magnetic flux that generates the induced voltage large while minimizing the third, fifth, and seventh harmonic components in order to achieve high output, high speed rotation, and low vibration.

The present invention is realized by the following means.
"Method 1"
A rotating electric machine having a three-phase stator with concentrated windings consisting of Q iron cores arranged in the radial or axial direction at equal pitch, and a surface magnet rotor with a pole number p in which an equal number of N and S permanent magnets are arranged alternately through an air gap, or a buried magnet rotor with a pole number p in which an equal number of N and S permanent magnets are arranged alternately and embedded in a magnetic rotor yoke, wherein n is an integer of 2 or more , the number of winding poles q for one phase is defined as q = 2n + 1, the total number of winding poles Q for the three phases is set to Q = 3q, and the number of rotor poles p is p = Q ± 1, and the q winding poles of each phase are arranged non-adjacently with a gap of one winding pole between each other.
"Method 2"
A rotating electric machine having a three-phase stator with concentrated windings made of Q iron cores arranged at equal pitch in the radial or axial direction, and a surface magnet rotor with a pole number p in which an equal number of N-pole and S-pole permanent magnets are arranged alternately through an air gap, or a buried magnet rotor with a pole number p in which an equal number of N-pole and S-pole permanent magnets are arranged alternately and embedded in a magnetic rotor yoke, where n is an integer of 2 or more, the number of winding poles q for one phase is defined as q = 2n + 1 and is an odd number of 5 or more, the total number of three-phase winding poles Q is set to Q = 3q, and the number of rotor poles p is p = Q ± 1, and where the q winding poles of each phase of the three-phase winding are designated T 1 , T 2 , T 3 , . . . T q-1 , T q in order, _there is _a one winding _pole interval _between T ₁ and T 2, and between _T q _- 1 _{and T} q _{, and} A rotating electric machine in which the winding poles of _q-1 are arranged adjacent to each other, where n is 2 or 3 (q is 5 or 7).
"Method 3"
a three-phase stator with concentrated windings made of Q iron cores arranged at equal pitch in the radial or axial direction; and a surface magnet rotor with p poles in which the same number of N-pole and S-pole permanent magnets are arranged alternately via an air gap, or a recessed magnet rotor with p poles in which the same number of N-pole and S-pole permanent magnets are embedded in a magnetic rotor yoke, each of which is configured as a set of opposing stator and rotor combinations, and these combinations are repeated for m sets,
A rotating electric machine in which n is an integer of 2 or more, the number of winding poles q for one phase in one set is defined as q = 2n-1, the number of winding poles Q for three phases in one set is defined as Q = 3q, the number of poles p of the rotor in one set is defined as p = Q ± 1, the total number of windings K for m sets is defined as K = 3mq, the number of poles M for m sets is defined as M = m (Q ± 1), and the q winding poles of each phase in one set are arranged non-adjacently with an interval of one winding pole between each other, where m is 2, 4, or 6, and M is 20 or more.
"Method 4"
A rotating electric machine according to the first aspect, wherein the rotor is made of a polar anisotropic magnet.
"Measure 5"
A rotating electric machine according to the first aspect, wherein each phase winding is driven as a delta connection.

1) Since the number of rotor poles p and the number of stator winding poles Q have no common divisors, their least common multiple (hereinafter abbreviated as LCM) is the product of p and Q, which is a large value, and low cogging torque is obtained.
2) In Method 1, the difference between p and Q is only 1, so the fundamental wave rate of the simplified winding rate, which will be described later, becomes larger, which is advantageous for achieving higher torque.
3) In Means 1, _when n=2, q=5 and Q=15, and when p=14 or 16 , the fundamental rate P1 of the interlinkage magnetic flux, which will be described later, is 0.833 and the third harmonic rate _P3 is 0. Similar values are also obtained for Q and p when n>2 . The fact that the third harmonic is zero means that the motor is suitable for delta connection drive and high speed operation. In contrast, in the conventional technology using the adjacent winding method , _P3 is not zero and copper loss increases when driving with delta connection, so this is where the great inventive step of the present invention lies.
4) In the case of Means 2, when n=2, q=5, Q=15, and p=14 or 16, the fundamental wave rate _P1 is 0.914 and the third harmonic rate _P3 is 0.400. Compared with the prior art, the fundamental wave rate is slightly lower, but the third harmonic rate is significantly smaller, and the present invention has a great advancement in the high-speed rotation characteristics.
5) In the case of m=2, 4, or 6 in the third means, the LCM can be made large and the unbalanced electromagnetic force disappears, so that a rotating electric machine with high speed rotation and low vibration can be realized.
6) If a polar anisotropic magnet rotor is used in the first aspect of the present invention, a highly efficient rotating electric machine that does not require a back yoke, is lightweight, inexpensive, and has excellent high-speed capabilities, can be realized.
7) If a rotor configured with pseudo-pole magnets is used in means 1 to 3 of the present invention, not only is the number of magnets reduced by half, making it lighter and less expensive, but it also provides a highly efficient rotating electric machine with excellent high-speed performance, with even lower cogging torque and a large field-weakening effect.
8) In the sixth aspect of the present invention, by making some of the q winding poles different from the other winding numbers, the induced voltage can be made even more sinusoidal. In particular, when q =5 or more, (q-2) poles are adjacent, so by appropriately making the central pole or two poles at both ends of the q winding poles different from the other winding numbers, the back electromotive force waveform can be adjusted to be even more sinusoidal.

FIG. 2 is a cross-sectional view perpendicular to the axial direction showing the general configuration of a stator and a rotor for explaining the present invention. Linear expansion of the winding pole position for one phase outside the present invention and its excitation polarity diagram FIG. 1 is a diagram of a winding pole excitation sequence for 120-degree conduction of Q=9 outside the present invention. Stepping principle diagram for Q=9, p= 10 outside the present invention Polarity diagram of the stator winding poles linearly expanded for star-connected three-phase excitation with Q=9, not according to the present invention A diagram of the stator and rotor of the present invention, linearly expanded with one-phase excitation , Q=15, p=14. Polarity diagram of the stator winding poles linearly expanded for the Q=15 star-connected three-phase excitation of the present invention Polarity diagram of the stator winding poles linearly expanded for a star-connected three-phase excitation with Q=15 according to another embodiment of the present invention. Polarity diagram of the stator winding poles linearly expanded for a star-connected three-phase excitation with Q=21 according to another embodiment of the present invention. Polarity diagram of the stator winding poles linearly expanded for a two-fold star-connected three-phase excitation with Q=9, p= 10 of the prior art A linear expansion of the stator winding poles for one-phase excitation with Q=9 and p=8 in the prior art reference 1) A linear expansion of the stator winding poles for three-phase excitation with Q=9 and p=8 in the prior art reference 2) A linear expansion of the stator winding poles for three-phase excitation with Q=15 and p=14 in the prior art [Table 1] Excitation sequence table for 3-phase 120-degree current conduction [Table 2] Simple winding ratio comparison table [Table 3] Simple winding ratio comparison table

Hereinafter, a rotating electric machine according to this embodiment will be described with reference to the drawings.

FIG. 1 is a diagram showing the schematic configuration of a stator and rotor of an outer rotor type in accordance with the first embodiment of the present invention, where n=2, q=5, Q=15, and p=14, and is a cross-sectional view perpendicular to the axial direction.
Reference numeral 1 denotes a stator core, on which 15 radially protruding pole teeth are provided at equal pitches, and although reference numerals for all of the pole teeth are omitted, a concentrated winding 4 is wound, with winding poles 11 to 25 distributed in a clockwise direction. In FIG. 1, reference numerals are given to the pole teeth 11, 13, 15, 22, and 24 for one phase , and the other reference numerals are omitted. Similarly, only the winding of the winding pole 11 is referenced as the winding 4 , and the reference numerals for the windings of the other winding poles are omitted. The rotor has a total of 14 poles, that is, p = 14 , with alternating N and S poles of the rotor magnetic poles made of permanent magnets, and reference numerals are given to only two of them, the N pole 31 and the S pole 32 , but the following magnetic poles with reference numerals omitted are provided in a clockwise direction, alternating N and S poles , to form the rotor 2, which faces the 15 winding poles via an air gap.
Reference numeral 2-1 denotes a structure of the rotor 2 which also serves as a back yoke for each of the 14 rotor magnetic poles , and the rotor shaft is not shown.
The winding method of the present invention is composed of three phases, U, V, and W, but the winding of one phase, U, is configured with 11, 13, 15, 22, and 24 wound with the same winding polarity at intervals of one winding pole, that is, skipping 12, 14, and 23 and 25. If this stator configuration is linearly expanded in the circumferential direction of the air gap, it will become Figures 6 and 7, which will be described later . In Figure 6 , the upper row shows the winding pole numbers corresponding to Figure 1, and the lower row shows the excitation phase and its excitation polarity.
1 shows that when the N poles of the winding pole 11 and the rotor pole 31 face each other at their centers, the N pole of the rotor pole 31 facing the winding pole 13 is offset by an angle δ. Therefore, although the reference numerals are omitted in the figure, the N pole of the rotor pole 31 facing the winding pole 24 is also offset by δ. Furthermore, the rotor pole 31 facing the winding pole 15 and the rotor pole 31 facing the winding pole 22 are each offset by 2δ.
In this case, δ is 3.428° in mechanical angle, but since the rotor has 14 poles, it is 24° in electrical angle.
Therefore, the offset angles between the five U-phase winding poles and the opposing rotor poles correspond to FIG. 1 and are as follows: winding pole 11 is 0°, winding poles 24 and 13 are δ or 24° , and winding poles 15 and 22 are 48° .
Means 1 is a rotating electric machine in which p = Q ± 1, Q = 3q, q = 2n + 1, n being an integer of 2 or more , and the winding method is such that the number of winding poles per phase, q, is an odd number of 5 or more , and the winding poles are arranged non-adjacently with a gap of one winding pole between each other , and therefore Figures 2, 3, 4, and 5 below are outside the scope of the present invention.
In this case, the induced voltage for one phase is the sum of the time differentials of the interlinkage magnetic flux of each of the windings 22, 24, 11, 13, and 15, so the fundamental rate _P1 of the permeance of the interlinkage magnetic flux, the third harmonic rate _P3 , and the fifth and seventh harmonic rates are called the simplified winding rate, defined as follows (1) and (2), and obtained by substituting a numerical value for δ and performing a numerical calculation. The purpose of this is to use this as an evaluation factor for comparing the present invention with the prior art. Even if the winding rate is a concentrated winding method, it is common to calculate the short-pitch winding factor and the distributed winding factor, and use their product as the winding factor. However, the fundamental rate _P1 of the permeance of the interlinkage magnetic flux adopted here is close to the distributed winding factor, and it is more convenient to use this simplified winding rate for evaluation including the harmonic rate.
δ= 24° (electrical angle), and as described above with reference to FIG. 1, the offset angle of the winding pole 11 from the opposing rotor pole is δ=0, and the offset angle of the winding poles 15 and 22 is 2δ, so that:
_P1 = {cos0 + 2 · cosδ + 2 · cos2δ} / q = {1 + 2 · cos24 + 2 · cos48} / 5 = 0.833 (1)
_P3 = {cos0 + 2 · cos3δ + 2 · cos6δ} / q = {1 + 2 · cos72 + 2 · cos144} / 5 = 0 (2)
A comparison of the conventional products 1) and 2) using numerical calculations will be given later.
That is, the larger _P1 is, the higher the induced voltage is, and therefore the torque and efficiency are improved.
In addition, when _P3 is large, the induced voltage for one phase becomes closer to a trapezoidal wave, which is plump, rather than a sine wave. The third harmonic component of the induced voltage is cancelled between the lines in a star connection and has no effect, but in a delta connection, a circulating current is generated inside the ring connection, resulting in increased copper loss.
In the winding method of the present invention, since _P3 is zero, a delta connection can also be used without any problems.

FIG. 3 is a diagram of the pole excitation sequence of a three-phase winding with 120-degree conduction of Q=9 according to the present invention. In the diagram, U, V, and W mean that U , V , and W are connected so that the polarity is reversed. The same applies to the following diagrams. The top row of FIG. 3 represents the nine winding poles in order, numbered 1 to 9. From the next row to the bottom, the excitation is numbered 1 to 6, showing the winding pole excitation sequence of the present invention when the rotor poles of the present invention are 8 or 10 poles and there are 9 winding poles when switching with 120-degree conduction, which is a typical excitation method for brushless motors. Each row excitation switching is performed in six steps of 60 degrees electrical angle, for a total of 360 electrical degrees. FIG. 3 corresponds to the stepping operation diagram of FIG. 4, which will be described next.
The polarities of the voltages applied to the three star-connected terminals U, V, and W of a three-phase brushless motor of a bipolar inverter with three-phase 120-degree conduction are as shown in Table 1.
Fig. 4 is a confirmation diagram for reliable operation of the brushless motor of the present invention. The relative positions of the stator and the opposing rotor are linearly expanded in the circumferential direction of the air gap, and this is a confirmation diagram for so-called six-step stepping, like the stepping operation of a three-phase permanent magnet stepping motor. In this case, the motor steps 60 degrees in electrical angle with excitation of one phase at a time, so that the motor steps 360 degrees in electrical angle in six steps. If stepping can be confirmed with one-phase excitation, the motor will also operate reliably with two-phase excitation drive with 120 degrees of current flow as a brushless motor.
The stepping diagram is shown for p = Q ± 1, Q = 9, p = 8, but it also works in the same way when p = 10. It is assumed that the winding poles of each of the three phases are star-connected, with the ends of the windings shorted together as a common terminal, but it also works in the same way with a delta connection, where the end of each phase is connected in sequence to the start of the next phase to form a ring connection.
As can be seen from Figure 4, in excitation 1), the U phase is all N polarity and faces the S pole of the rotor. Next, in excitation 2), the V phase is all S polarity and faces the N pole of the rotor. In excitation 3), the W phase is all N polarity and faces the S pole of the rotor. Since the polarity of the winding poles for one phase is all the same and the next phase excitation is reversed, the rotor magnetic flux becomes a magnetic path that forms a closed magnetic path between the U phase and V phase, or the V phase and W phase, or the W phase and U phase. In excitation 4), the stator winding poles are excited to the opposite polarity of 1), 5) to 2), and 6) to 3), and the rotor advances by 60 electrical degrees at a time as shown in Figure 4.

Fig. 5 shows the excitation polarity of two rows, excitation 1 and 2, shown in Fig. 3, at once. This corresponds to a table that summarizes the three steps of sequential excitation 1), 2), and 3) for the three phases, U, V, and W, as shown in Fig. 4, and is called a three-phase excitation representation for convenience, and allows the winding method configuration of the present invention to be expressed in a single table. Since Fig. 3 can be obtained by applying Table 1 from Fig. 5, the representation of the winding method of the present invention will be shown as a three-phase excitation as shown in Fig. 5 as necessary, even when Q and p are changed.

FIG. 6 is a diagram showing the opposing relationship between the stator and rotor in the case of one-phase excitation with n=2, q=5, and therefore Q=15 and p=14, in the first embodiment of the present invention. Row A of the table shows the winding pole numbers of the stator, which are arranged over 360 degrees at equal pitches from 1 to 15. Row B shows the excited phase and its polarity. Similarly, rows A and B of the figures after FIG. 6 show the winding pole numbers, and row B shows the excitation phase and its polarity. Row C of FIG. 6 shows the rotor magnet. This is a relationship diagram in which the winding pole 5 and the rotor S pole are opposed in excitation B _1. The rotor magnet in row C moves 60 degrees in electrical angle to the left in excitation B ₂ , so that the winding pole 10 and the rotor N pole are opposed, and further moves 60 degrees in electrical angle to the left in excitation B ₃ , so that the winding pole 15 and the rotor S pole are opposed, and similarly rotates leftward by 60 degrees electrical angle each in excitations B ₃ to B ₆ .
The five U-phase winding poles are spaced apart by one winding pole. Since p = Q ± 1, this winding method is also valid when Q = 15 and p = 16.

Fig. 7 shows three phases in the excitation order of U, V, W, with n=2, q=5, and therefore Q=15, according to the present invention in Means 1. In other words, the one-phase display in Fig. 6 is shown in terms of the stator excitation polarity for three phases, and corresponds to Fig. 5 in the case of Q=9 described above.
Although the stepping diagram in this case is omitted, if a stepping diagram similar to that in FIG. 4 is created, the stepping of 60 degrees can be confirmed.

図９も手段２を適用した本発明のＱ＝２１，ｐ＝２０または２２の場合の、３相励磁状態での固定子巻き線極の極性図である。ｎ＝３でｑ＝７の場合である。ｑ個の巻き線極は、手段２を適用すれば、３相巻き線の各相の巻き線極のｑ個は隣接的に配置された位置から、その両端の巻き線極が１巻き線極分離れた非隣接状に配置されるため、１相分の巻き線極の配置は、５個隣接してその両端は１巻き線極分、分離した非隣接状に配置される。即ち、分離して１極分飛ばした箇所を０で表現し、Ｕ相の励磁で示せば、Ｕ０ＵＵＵＵＵ０Ｕ、とｑ＝７なので、０を除いて７個の文字で表現できる。これを３相励磁状態で表現すれば、図９となる。
この場合、Ｑ＝２１、ｐ＝２０で、図９の巻き線極５と対向する回転子磁極とが対向して、ずれ角が零であるとき、巻き線極４および６と、対向する回転子磁極のずれ角をδとすれば、巻き線極３および７と、対向する回転子磁極のずれ角は２δ、巻き線極１および９と、対向する回転子磁極のずれ角は４δとなる。
δ＝８．５７１４°（電気角）で、図９のＵ相の７個、即ち、図９の巻き線極１，３，４，５，６，７，９の巻き線極の合計のＰ_１，Ｐ_３は以下となる。

Fig. 8 is a polarity diagram of the stator winding poles in a three-phase excitation state when Q = 15, p = 14 or 16 according to the present invention to which Means 2 is applied, and shows the case of n = 2 and q = 5. When Means 2 is applied, the q winding poles of each phase of the three-phase winding are arranged in a non-adjacent state with the winding poles at both ends separated by one winding pole from the adjacently arranged positions, so that the arrangement of the winding poles for one phase is 5 - 2 = 3, that is, three adjacent poles, with both ends separated by one winding pole, and arranged in a non-adjacent state. In other words, if the separated one pole-width location is represented by 0, and the excitation of the U phase is represented as U0 U U U 0U, and q = 5, it can also be expressed in this way. If this is expressed in a three-phase excitation state, it becomes Fig. 8.
In this case, when Q = 15, p = 14, and the offset angle between the rotor pole facing winding pole 4 in Figure 8 and the opposing rotor pole is zero, if the offset angle between the rotor poles facing winding poles 3 and 5 is δ, the offset angle between the rotor poles facing winding poles 1 and 7 is 3δ, and δ = 12° (electrical angle), the total P1 and _P3 of the five winding poles of the U phase in Figure 8, i.e., winding poles ₁ , 3, 4, 5, and 7 in Figure 8, are as follows.

Fig. 9 is also a polarity diagram of the stator winding poles in a three-phase excitation state when Q=21, p=20 or 22 according to the present invention to which Means 2 is applied. This is the case of n=3 and q=7. When Means 2 is applied, the q winding poles of each phase of the three-phase winding are arranged in a non-adjacent state with the winding poles at both ends separated by one winding pole from the adjacently arranged positions, so that the winding poles for one phase are arranged in a non-adjacent state with five adjacent poles and both ends separated by one winding pole. In other words, if the separated one pole-width location is represented by 0 and the excitation of the U phase is represented as U 0U U U U U U0 U , and q=7, it can be represented by seven characters excluding 0. If this is represented in a three-phase excitation state, it becomes Fig. 9.
In this case, when Q = 21, p = 20, and winding pole 5 in Figure 9 faces the opposing rotor pole and the offset angle is zero, if the offset angle between winding poles 4 and 6 and the opposing rotor pole is δ, then the offset angle between winding poles 3 and 7 and the opposing rotor pole is 2δ, and the offset angle between winding poles 1 and 9 and the opposing rotor pole is 4δ.
When δ=8.5714° (electrical angle), the sums of P ₁ and P ₃ of the seven U-phase winding poles in FIG. 9, that is, winding poles 1, 3, 4, 5, 6, 7, and 9 in FIG.

By increasing n using means 1 and 2 of the present invention, the winding method of the present invention can be applied to various combinations that satisfy p = Q ± 1, such as Q = 33 and p = 32, 34, etc. Furthermore, by increasing the number of poles p, the maximum torque that the motor can produce is proportional to the number of poles p, so a rotating electric machine suitable for high torque applications such as electric vehicles and drones can be obtained. Furthermore, as p and Q increase, the LCM also increases and the cogging torque decreases. If the cogging torque is large, the no-load current also increases, hindering efficiency.

If Q is an odd number, the force acting in the radial direction on the rotor by the stator when one phase is excited cannot be completely canceled out; in other words, an electromagnetic force called unbalanced electromagnetic force or side pull will come into play, which is unfavorable in terms of vibration and noise.
However, in reality, Q=9, p=8 or 10 is widely used because there is no common factor between Q and p, so the LCM is large and the cogging torque is small.
Furthermore, by using the third aspect of the present invention, p = m (3q ± 1), and selecting m as 2 or more, it becomes an m-fold rotating electric machine of the first aspect, and the unbalanced electromagnetic force can be offset and eliminated. Also, by increasing the number of poles, the LCM is large, and p is large with low cogging torque, so a high-torque rotating electric machine with high limit torque can be expected.
Fig. 10 shows three-phase excitation of 18 winding poles when m=2, q=3, Q=3mq=18, and p=16 or 20, using Means 3. In other words, this is twice the three-phase rotating electric machine with winding poles Q=9 and pole number p=8 or 10 in Fig. 5, and the unbalanced electromagnetic force disappears, resulting in a rotating electric machine with low vibration. Note that the fundamental wave rate _P1 and the third harmonic rate _P3 of the permeance of the interlinkage magnetic flux in this case are the same as the values of equations (1) and (2).

また図１２は引例２）の３相励磁の表示の場合であり、ｑ＝３が隣接配置されている。同様にして、１相分に関しては、以下となる。

従来技術、及び本発明の手段１，２のＰ_１，Ｐ_３を小数点４桁以下は切り捨てて３桁までの表示で、一覧表として比較すれば、表２となる。
即ち、表２のＰ_１が大きいほど、誘起電圧も大きくなり、従ってトルク、効率も向上する。
しかし、従来技術の引例１のＰ_１は本発明の手段１が０．８４４であるのに対して０．６５７と劣り、Ｐ_３は手段１が零で小さいのに対して、０．５７７と大きいので、デルタ結線駆動には適さない回転電機となる。
また、引例２のＰ_１値は本発明の手段１より、やや勝るが、Ｐ_３は０．６６６とかなり大きな値で、本発明のように零でないので、引例１と同様にデルタ結線駆動に適さない回転電機である。ここに、本発明の進歩性がある。
図１３は引例２と同類の従来技術で、ｑ＝５を隣接配置した、Ｑ＝１５，ｐ＝１４の１相励磁時の固定子と回転子の対向図である。この場合、１相分に関して、Ｐ_１，Ｐ_３は、（５）、（６）式に準じて同様に計算すれば、以下となる。

この値を表２に入れて、本発明のＱ＝１５，ｑ＝１４の手段２と比較すると、以下のようになる。
Ｐ１値は本発明の手段２の０．９１４と比較すれば、やや勝るが、Ｐ_３は本発明品が０．４００に対して、０．６４７とかなり大きな値で、引例１と同様にデルタ結線駆動に適さない回転電機である。
ここに、本発明の進歩性がある。
更に巻き線極数のＱ＝９の場合、引例２と本発明の比較では、引例２の３個のｑが隣接しているのに対して、本発明の３個のｑは非隣接配置で、より分散しているため、本発明品の方が、１相励磁時の不平衡電磁力がより少なくなり、低振動化に有利な構成といえる。Ｑ＝１５，ｐ＝１４の不平衡電磁力は、従来技術は図１３に示すように、ｑである５個の巻き線極が隣接しているのに対して、本発明の手段２の方は、より分散配置されているため、小さくなり、低振動化に有利な構成といえる。
また表２より、本発明の回転電機は表２のＬＣＭは、ｐ＝Ｑ±１の効果で、大きな値が得られるので、コギングトルクの低減に有利な構造を採用していることも分かる。
更に表２から次のことが分かる。
１）本発明の手段１と手段２によるＱ＝９，ｐ＝８は巻き線極配置が同一でＰ_１，Ｐ_３の値も同一である。
２）Ｑを１５以上と大きくした場合、手段１と手段２とでは、巻き線極配置も、Ｐ_１，Ｐ_３の値も相違する。
３）本発明の手段１内でＱを変化させても、Ｐ_１，Ｐ_３の値はほぼ同一である。
４）本発明の手段３内でＱを変化させても、Ｐ_１，Ｐ_３の値はほぼ同一である。
５）本発明の手段３によるＱ＝２７の場合はｐ＝２４で、手段１でのＱ＝２７の場合、ｐは２６となるので異なる回転電機となる。FIG. 11 is an example of the prior art 1) and shows the opposing relationship between the stator and rotor during one-phase excitation with Q=9 and p=8.
On page 117 of cited document 1, the configuration shown in FIG. 11 is disclosed.
In this case, the following applies to one phase in FIG. 11, for example, the U phase.

12 shows the case of three-phase excitation in the reference 2), where q=3 is arranged adjacently. Similarly, the following applies to one phase.

Table 2 is obtained by comparing P ₁ and P ₃ of the prior art and the means 1 and 2 of the present invention in a list format, with the values rounded down to three decimal places.
That is, the larger _P1 in Table 2 is, the larger the induced voltage is, and therefore the torque and efficiency are improved.
However, _P1 in Reference 1 of the prior art is 0.657, which is inferior to 0.844 in Means 1 of the present invention, and _P3 is 0 in Means 1, which is small, but is large at 0.577, making this a rotating electric machine unsuitable for delta connection drive.
Also, although the _P1 value of Reference 2 is slightly better than Means 1 of the present invention, _P3 is a fairly large value of 0.666, which is not zero like the present invention, and therefore the rotating machine is not suitable for delta connection drive, just like Reference 1. Herein lies the inventive step of the present invention.
13 shows a conventional technique similar to that of Reference 2, in which q=5 is arranged adjacently to the stator and rotor during one-phase excitation with Q=15 and p=14. In this case, _P1 and _P3 for one phase can be calculated in the same manner according to equations (5) and (6) as follows:

When these values are entered in Table 2 and compared with Means 2 of the present invention in which Q=15 and q=14, the results are as follows.
The P1 value is slightly better than that of the present invention's means 2 (0.914), but _P3 is a fairly large value of 0.647 compared to 0.400 for the present invention, and this rotating electric machine is not suitable for delta connection drive, just like Reference 1.
Herein lies the inventive step of the present invention.
Furthermore, in the case of the number of winding poles Q=9, in a comparison between Reference 2 and the present invention, the three q's in Reference 2 are adjacent, whereas the three q's in the present invention are non-adjacent and more dispersed, so that the product of the present invention has less unbalanced electromagnetic force during one-phase excitation and is advantageous in terms of vibration reduction. The unbalanced electromagnetic force of Q=15, p=14 is smaller in the prior art, as shown in Figure 13, because the five winding poles, or q, are adjacent, whereas the means 2 of the present invention is more dispersed, so that the configuration is advantageous in terms of vibration reduction.
It can also be seen from Table 2 that the rotating electric machine of the present invention employs a structure that is advantageous for reducing cogging torque, since the LCM in Table 2 achieves a large value due to the effect of p=Q±1.
Furthermore, from Table 2, the following can be seen:
1) The winding pole arrangements of Q=9, p=8 according to the means 1 and 2 of the present invention are the same, and the values of _P1 and _P3 are also the same.
2) When Q is increased to 15 or more, the winding pole arrangement and the values of P ₁ and P ₃ are different between means 1 and means 2.
3) Even if Q is changed in the first aspect of the present invention, the values of P ₁ and P ₃ are almost the same.
4) Even if Q is changed in the means 3 of the present invention, the values of P ₁ and P ₃ are almost the same.
5) When Q=27 according to the means 3 of the present invention, p=24, whereas when Q=27 according to the means 1, p is 26, resulting in a different rotating electric machine.

Furthermore, for the means 2 of the present invention and the prior art reference 2, the fifth harmonic ratio _P5 and the seventh harmonic ratio P7 of the permeance of the interlinkage magnetic flux when Q is 15, 21, and ₂₇ were calculated in the same manner and are shown in Table 3. From Tables 2 and 3, the following can be seen.
1) Q=9 has P ₃ =0, so it is suitable for delta connection.
2) In comparison with Reference 2, at Q=15 the fifth harmonic is zero, at Q=21 the seventh harmonic is zero, and at Q=27 the seventh harmonic is also close to zero. The small values of _P5 and _P7 mean that the rotating electric machine generates a sine wave back electromotive force, and torque unevenness due to current in the case of sine wave drive is small.
3) Q=15, 21, 27, etc. have a third harmonic, but the fifth and seventh harmonics are small, so they are suitable for star connection.
4) To obtain smooth rotational operation, the LCM of Q and p should be set large to reduce the cogging torque, and also reduce torque irregularities caused by back electromotive force.
5) For example, considering the results of Table 3, if the magnet pole width, end shape, convex lens-shaped magnet thickness, or stator tooth shape, etc. are devised so as to remove the seventh harmonic when Q=15, and the fifth harmonic when Q=21, this will be an even more effective countermeasure against torque unevenness.
6) As described in Means 6, by making some of the q winding poles different from the others in number of turns, the induced voltage can be made even more sinusoidal. In particular, when Q=5 or more, (q-2) poles are adjacent, so by appropriately making the central pole or two poles at both ends of the q winding poles different from the others in number of turns, the back electromotive force waveform can be made even more sinusoidal. The magnitude of the back electromotive force is proportional to the number of turns, and the effect of correcting the back electromotive force waveform to a sinusoidal wave is significantly obtained by a configuration in which the winding poles for one phase are adjacent.

The rotating electric machine of the present invention becomes an excellent rotating electric machine when a polar anisotropic magnet rotor is used. If the winding poles are adjacent as in means 2, or if the winding poles are separated by one winding pole as in means 1 or 2, a magnetic path of the permanent magnet magnetic flux is formed between the adjacent winding poles, so if a polar anisotropic magnet is used, the back yoke can be omitted, which makes the machine cheaper and lighter. That is, as explained with reference to FIG. 1, the N pole 21 and the S pole 22 of the permanent magnet 2 are magnetized and oriented inside the magnet, and the rotor core 3 may be made of a non-magnetic material such as resin, or the permanent magnet 2 and the rotor shaft 4 may be in direct contact without using 3. FIG. 1 shows an inner rotor type, but in the outer rotor type, the rotor back yoke becomes large, so it can be replaced with resin or the like, or it can be made of a non-magnetic material such as aluminum, or it can be made unnecessary, which is effective for making the machine cheaper and lighter. Polar anisotropic magnets are magnets that have been made anisotropic by aligning the magnetic path during magnet molding to correspond to the number of poles determined inside the permanent magnet. One of the authors presented this at the Small Motor Technology Symposium hosted by the Japan Management Association in 1983, and the material was mentioned above as reference 3) as related technology. This significantly improves motor torque and also makes the induced voltage sinusoidal.
The present invention has the above-mentioned characteristics, and a synergistic effect can be expected.
Furthermore, if the winding poles are adjacent to each other or are separated by one winding pole, a magnetic path for the permanent magnet flux is formed between the adjacent winding poles in a similar manner, so the present invention is also applicable to a pseudo-pole type rotor.
Pseudo poles, also called consequent poles, are a type of rotor magnet arrangement in which, instead of alternating N and S poles, only magnets of the same polarity, for example only N pole magnets, are arranged, with the S poles being iron cores or space, and the number of permanent magnets is halved, reducing cogging torque.Compared to polar anisotropic magnets, a back yoke is required, but in the case of a rotor with alternating N and S poles, the pseudo pole arrangement configuration has a more pronounced field weakening effect in the case of field weakening drive of a brushless motor, which is advantageous for higher speeds.

The present invention uses a newly created winding configuration in which the stator winding poles are concentrated windings, and the number of these winding poles and the number of rotor magnetic poles can differ by ±1.As a result, despite the small number of winding poles, the cogging torque is small and the unbalanced electromagnetic force is also smaller than that of conventional products of the same configuration; if the number is doubled or more, the unbalanced electromagnetic force will disappear and a low-vibration, high-torque brushless motor can be obtained, which has extremely great industrial applicability.

1: stator core 31, 32 : rotor poles 11, 13, 15, 22, 24 : winding poles 4: winding
2: Rotor 2-1: Rotor back yoke and structure

Claims (5)

A rotating electric machine having a three-phase stator with concentrated winding of winding poles consisting of Q iron cores arranged at equal pitch in the radial or axial direction, and a surface magnet type rotor with pole number p in which an equal number of N and S permanent magnets are arranged alternately through an air gap, or a buried magnet type rotor with pole number p in which an equal number of N and S permanent magnets are arranged alternately and embedded in a magnetic rotor yoke, wherein n is an integer of 2 or more , the number of winding poles q for one phase is defined as q = 2n + 1, the total number of winding poles Q for three phases is set to Q = 3q, and the number of rotor poles p is p = Q ± 1, characterized in that the q winding poles of each phase are arranged non-adjacently with a gap of one winding pole between each other. A rotating electric machine having a three-phase stator with concentrated windings made of Q iron cores arranged at equal pitch in the radial or axial direction, and a surface magnet type rotor with a pole number p in which an equal number of N-pole and S-pole permanent magnets are arranged alternately through an air gap, or a recessed magnet type rotor with a pole number p in which an equal number of N-pole and S-pole permanent magnets are arranged alternately and embedded in a magnetic rotor yoke, where n is an integer of 2 or more, the number of winding poles per phase q is defined as q = 2n + 1 and is an odd number of 5 or more, the total number of three-phase winding poles Q is set to Q = 3q, and the number of rotor poles p is p = Q ± 1, and the q winding poles of each phase of the three-phase windings are sequentially arranged as T ₁ , T ₂ , T ₃ , , , T _q-1 , T _q In this case, T ₁ and T ₂ Between and T _q-1 and T _q The interval between the winding poles is one. ₂ ~T _q-1 The winding poles of the first and second windings are arranged adjacent to each other, where n is 2 or 3 (q is 5 or 7). a three-phase stator with concentrated windings made of Q iron cores arranged at equal pitch in the radial or axial direction; and a surface magnet rotor with q poles in which the same number of N-pole and S-pole permanent magnets are arranged alternately via an air gap, or a recessed magnet rotor with q poles in which the same number of N-pole and S-pole permanent magnets are embedded in a magnetic rotor yoke, each of which is repeated for m sets of opposing combinations of a stator and rotor,
A rotating electric machine characterized in that n is an integer of 2 or more, the number q of winding poles per phase in one set is defined as q = 2n-1, the number Q of winding poles for three phases in one set is defined as Q = 3q, the number p of poles of the rotor in one set is defined as p = Q ± 1, the total number of windings K for m sets is defined as K = 3mq, the number of poles M for m sets is defined as M = m (Q ± 1), and the q winding poles of each phase in one set are arranged non-adjacently with an interval of one winding pole between each other, where m is 2, 4, or 6, and M is 20 or more. 2. The rotating electric machine according to claim 1 , wherein the rotor employs a polar anisotropic magnet. 2. The rotating electric machine according to claim 1, wherein each phase winding is driven as a delta connection.

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