JP7452200B2 - motor control device - Google Patents

Description

The present disclosure relates to a motor control device.

A motor control device that controls driving of a motor includes an inverter that generates three-phase alternating current voltage (hereinafter sometimes referred to as "three-phase voltage") that is applied to the motor. The inverter is composed of a plurality of switching elements.

Furthermore, when vector control is used for the motor control device, the motor control device adjusts the d-axis current command value and the q-axis current command value so that the motor rotation speed matches the speed command value (target speed). A d-axis voltage command value and a q-axis voltage command value are generated from the d-axis current command value and the q-axis current command value. Further, the motor control device converts the d-axis voltage command value and the q-axis voltage command value into three-phase voltage command values.

PWM (Pulse Width Modulation) is known as a technique for controlling an inverter based on three-phase voltage command values. PWM is a technology that changes the output voltage (that is, three-phase voltage) of an inverter by adjusting the length of on/off time of a plurality of switching elements that constitute the inverter. A signal for controlling on/off of a switching element (hereinafter sometimes referred to as a "PWM signal") is generated based on a comparison result between a carrier signal, which is a PWM carrier wave, and a comparison value. By controlling on/off of the switching element according to the PWM signal, a three-phase voltage is applied to the motor, and the driving of the motor is controlled. Hereinafter, the predetermined period of the carrier signal, which is a PWM carrier wave, may be referred to as a "carrier period."

Additionally, motor drive can be controlled without using a sensor (hereinafter sometimes referred to as a ``position sensor'') to detect the rotational position of the motor's rotor (hereinafter sometimes referred to as ``rotor position''). A technique to do this (hereinafter sometimes referred to as a "position sensorless method") is known. In the position sensorless method, the rotor position is estimated without using a position sensor by detecting three-phase currents flowing through the motor (hereinafter sometimes referred to as "motor currents").

Furthermore, a "single shunt detection method" is known as a motor current detection method. In the 1-shunt detection method, two phases of the motor current are detected based on the bus current flowing between the inverter that generates the three-phase voltage and the DC power supply, and the remaining one-phase current is detected as the two-phase current. Calculated from phase current using Kirchhoff's law. When calculating the remaining 1-phase current from 2-phase current using the 1-shunt detection method, the current of the 2 phases with different detection timings is used, so Kirchhoff's law is used to calculate the remaining 1-phase current. Errors may occur when doing so.

On the other hand, the phase currents of the two phases estimated at symmetrical timings with respect to the apex in one period of the PWM triangular carrier signal (hereinafter sometimes referred to as the "carrier apex") are averaged, and the carrier A technology is proposed that suppresses phase current calculation errors due to different detection timings by calculating the phase currents of two phases at the apex and then calculating the phase current of the remaining one phase from the calculated phase currents of the two phases. has been done.

JP2010-088260A Japanese Patent Application Publication No. 2001-327173 Japanese Patent Application Publication No. 2013-055772

However, in conventional technology, phase currents estimated in the first half and second half of one period of a triangular wave carrier signal are used to calculate the phase current, so the phase current calculated within the current carrier period is used to calculate the phase current in the next carrier period. Even if an attempt is made to use it for control, there is only less than half of one cycle of the triangular wave carrier signal remaining, so it cannot be used in time due to constraints such as the calculation speed of the microcomputer and the response characteristics of the inverter. In other words, in the conventional technology, the process of calculating the phase current (motor current) in the current carrier cycle is performed in the next carrier cycle, and the calculated motor current can only be used in the next carrier cycle. It becomes a cycle. Therefore, the time lag between the period corresponding to the calculated motor current (current carrier period) and the period of control using that motor current (the next next carrier period) becomes large. If the time lag between the cycle corresponding to the calculated three-phase motor current and the cycle in which control is performed using the motor current becomes large, the accuracy of motor control may decrease.

Therefore, the present disclosure proposes a technique that can improve the accuracy of motor control.

A motor control device of the present disclosure includes an inverter, a current detection section, a calculation section, and a generation section. The inverter converts a DC voltage supplied from a DC power source into a three-phase AC voltage, and applies the AC voltage to the motor. The current detection unit detects a bus current of the inverter using a resistor connected between the DC power supply and the inverter. The calculation unit calculates three-phase currents flowing through the motor based on the bus current. The generation unit generates a PWM signal that controls the inverter by PWM using a carrier signal having a predetermined period. Further, the generation unit temporally shifts the PWM signal forward or backward within the period of the carrier signal. Further, the calculation unit calculates, based on the shifted PWM signal, a bus current detected in the second half of the first carrier period and a bus current detected in the first half of the second carrier period following the first carrier period. The three-phase currents at a boundary timing between the first carrier period and the second carrier period are calculated using the bus current obtained by the calculation. Further, the generation unit generates three-phase voltage command values for generating the PWM signal to be used in a third carrier period following the second carrier period based on the three-phase currents at the boundary timing. do.

According to the disclosed technology, it is possible to improve the control accuracy of the motor.

FIG. 1 is a diagram illustrating a configuration example of a motor control device according to Example 1 of the present disclosure. FIG. 2 is a diagram illustrating a configuration example of a 3φ current calculator according to Example 1 of the present disclosure. FIG. 3 is a diagram illustrating an example of the operation of the motor control device according to the first embodiment of the present disclosure. FIG. 4 is a diagram illustrating an example of the operation of the interpolation calculation unit according to the first embodiment of the present disclosure. FIG. 5 is a diagram illustrating an example of the operation of the motor control device according to the first embodiment of the present disclosure. FIG. 6 is a diagram illustrating an example of the operation of the motor control device according to the first embodiment of the present disclosure. FIG. 7 is a diagram illustrating an example of the operation of the motor control device according to the first embodiment of the present disclosure. FIG. 8 is a diagram illustrating an example of the operation of the motor control device according to the first embodiment of the present disclosure. FIG. 9 is a diagram showing a comparative example. FIG. 10 is a diagram illustrating a configuration example of a motor control device according to Example 2 of the present disclosure. FIG. 11 is a diagram illustrating a configuration example of a 3φ current calculator according to Example 2 of the present disclosure. FIG. 12 is a diagram illustrating an example of a carrier signal according to Example 2 of the present disclosure. FIG. 13 is a diagram illustrating an example of comparison processing according to Example 2 of the present disclosure. FIG. 14 is a diagram illustrating an example of PWM signal shift processing according to Example 2 of the present disclosure. FIG. 15 is a diagram for explaining shift example 1 according to Example 2 of the present disclosure. FIG. 16 is a diagram for explaining shift example 1 according to Example 2 of the present disclosure. FIG. 17 is a diagram for explaining shift example 2 according to Example 2 of the present disclosure. FIG. 18 is a diagram for explaining shift example 2 according to Example 2 of the present disclosure. FIG. 19 is a diagram for explaining shift example 3 according to Example 2 of the present disclosure. FIG. 20 is a diagram for explaining shift example 3 according to Example 2 of the present disclosure. FIG. 21 is a diagram for explaining shift example 4 according to Example 2 of the present disclosure.

Hereinafter, embodiments of the present disclosure will be described based on the drawings. In the following embodiments, the same components are given the same reference numerals.

[Example 1]
FIG. 1 is a diagram illustrating a configuration example of a motor control device according to Example 1 of the present disclosure.

The motor M whose drive is controlled by the motor control device 100 shown in FIG. 1 is, for example, a PMSM (permanent magnet synchronous motor).

The motor control device 100 shown in FIG. 1 employs a 180-degree energization method (sine wave energization method), a position sensorless method, and a one-shunt detection method. For example, in controlling the motor M, the motor control device 100 detects a bus current Is using a shunt resistor Rs, and controls the drive of the motor M by generating a PWM signal based on the detected bus current Is. . The motor control device 100 includes an inverter 10, a current detection section 20, a calculation section 60, a voltage detection section 30, and a generation section 40.

The inverter 10 receives a DC voltage from a DC power supply _EDC , and receives three-phase (U-phase, V-phase, W-phase) PWM signals Up, Un, Vp, Vn, Wp, and Wn from the generation unit 40. The inverter 10 converts the DC voltage into a three-phase voltage according to the PWM signals Up to Wn, and drives the motor M by applying the converted three-phase voltage to the motor M.

The inverter 10 includes upper arm switching elements SWup, SWvp, SWwp and lower arm switching elements SWun, SWvn, SWwn. The inverter 10 converts the DC voltage into a three-phase voltage by turning on/off each of the switching elements SWup to SWwn according to the PWM signals Up to Wn. Free wheel diodes Dupd to Dwnd are connected to both ends of each of the switching switches SWup to SWwn.

The current detection unit 20 includes a shunt resistance Rs and a current detection circuit 21, and detects the bus current Is using the shunt resistance Rs.

A shunt resistor Rs is connected between the DC power source _EDC and the inverter 10. The shunt resistor Rs is inserted, for example, on an N line _LN that is a DC line between the N-side terminal of the DC power supply _EDC and the inverter 10. Note that the shunt resistor Rs may be inserted on the P line _LP , which is the DC line between the P-side terminal of the DC power supply _EDC and the inverter 10.

A bus current Is corresponding to the U-phase current, V-phase current, and W-phase current, which are motor currents, and the PWM signal flows through the shunt resistor Rs. At this time, a voltage drop occurs across the shunt resistor Rs. The current detection circuit 21 detects the bus current Is flowing through the shunt resistor Rs based on the magnitude of this voltage drop and the resistance value of the shunt resistor Rs, and outputs the detection result to the calculation unit 60.

The calculation unit 60 includes a 3φ current calculator 61. The 3φ current calculator 61 calculates motor currents iu, iv, and iw based on the bus current Is detected by the current detection unit 20 and the PWM signals Up to Wn generated by the generation unit 40, and calculates the motor currents iu, iv, and iw. iu, iv, and iw are output to the generation unit 40. Details of the 3φ current calculator 61 will be described later.

The voltage detection unit 30 includes a DC voltage detection circuit 31 and a DC voltage calculator 32, and detects the bus voltage. The DC voltage detection circuit 31 detects the bus voltage between the P line L _P and the N line L _N and outputs the detection result to the DC voltage calculator 32 . The DC voltage calculator 32 converts the detection result into a DC voltage Vdc, and outputs the converted DC voltage Vdc to the generation unit 40.

The generation unit 40 receives the speed command value ωm ^* from outside the motor control device 100 (for example, a higher-order controller of the motor control device 100), receives the motor currents iu, iv, and iw from the calculation unit 60, and generates a DC voltage. Vdc is input from the voltage detection section 30. The generation unit 40 generates PWM signals Up to Wn based on the speed command value ωm ^* , motor currents iu, iv, iw, and DC voltage Vdc. The generation unit 40 generates PWM signals Up to Wn so that the rotor of the motor M rotates at a target speed indicated by the speed command value ωm ^* .

The generation unit 40 includes a 3φ/dq coordinate converter 42, a position/velocity estimator 44, a 1/Pn converter 51, a subtracter 52, a d-axis current setter 48, a subtracter 46, and a speed control 49, a subtractor 47, a d-axis/q-axis voltage setter 45, a dq/3φ coordinate converter 43, and a PWM unit 41.

The 3φ/dq coordinate converter 42 receives the motor currents iu, iv, and iw from the 3φ current calculator 61, and converts the phase angle θdq of the rotating coordinate system (dq coordinate system) with respect to the fixed coordinate system (UVW coordinate system) into the position It is input from the speed estimator 44. The 3φ/dq coordinate converter 42 converts the current vector (iu, iv, iw) in the fixed coordinate system into the current vector (id, iq) in the rotating coordinate system using the phase angle θdq, and stores the coordinates in time series. do. The 3φ/dq coordinate converter 42 outputs the converted d-axis current id to the position/velocity estimator 44, the d-axis/q-axis voltage setter 45, and the subtracter 46, and converts the converted q-axis current iq into the position/speed estimator 44, the d-axis/q-axis voltage setter 45, and the subtracter 46. It is output to the speed estimator 44, the d-axis and q-axis voltage setter 45, and the subtracter 47.

The position/speed estimator 44 receives the d-axis current id and the q-axis current iq from the 3φ/dq coordinate converter 42, and converts the d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^* into d-axis and q-axis voltages. It is input from the setting device 45. The position/speed estimator 44 calculates the angular velocity ωe of the motor M and the phase of the rotational coordinate system based on the d-axis current id, the q-axis current iq, the d-axis voltage command value Vd ^* , and the q-axis voltage command value Vq ^*. The angle θdq is estimated, the angular velocity ωe is output to the 1/Pn converter 51, and the phase angle θdq is output to the 3φ/dq coordinate converter 42 and the dq/3φ coordinate converter 43.

The 1/Pn converter 51 receives the estimated electrical angular velocity ωe from the position/velocity estimator 44 . The 1/Pn converter 51 converts the electrical estimated angular velocity ωe into a mechanical angular velocity ωm at the rotor of the motor M by multiplying the estimated angular velocity ωe by 1/Pn (Pn is the number of pole pairs of the motor M). . The 1/Pn converter 51 outputs the converted angular velocity ωm to the subtracter 52.

The subtracter 52 receives the angular velocity ωm from the 1/Pn converter 51, and receives the speed command value ωm ^* from the outside of the motor control device 100 (for example, a higher-order controller of the motor control device 100). The subtractor 52 calculates the speed deviation Δω by subtracting the estimated angular velocity ωm from the speed command value ωm ^* , and outputs the calculated speed deviation Δω to the speed controller 49.

The d-axis current setter 48 generates a d-axis current command value id ^* having a predetermined value, and outputs the generated d-axis current command value id ^* to the subtractor 46.

The subtracter 46 receives the d-axis current command value id ^* from the d-axis current setter 48 and receives the d-axis current id from the 3φ/dq coordinate converter 42 . The subtracter 46 calculates the d-axis current deviation Δid by subtracting the d-axis current id from the d-axis current command value id ^* , and outputs the calculated d-axis current deviation Δid to the d-axis q-axis voltage setter 45. .

The speed controller 49 receives the speed deviation Δω from the subtractor 52 . The speed controller 49 controls the q-axis current command value iq ^* so that the speed deviation Δω approaches zero. The speed controller 49 outputs the controlled q-axis current command value iq ^* to the subtractor 47.

The subtracter 47 receives the q-axis current command value iq ^* from the speed controller 49 and the q-axis current iq from the 3φ/dq coordinate converter 42 . The subtracter 47 calculates the q-axis current deviation Δiq by subtracting the q-axis current iq from the q-axis current command value iq ^* , and outputs the calculated q-axis current deviation Δiq to the d-axis and q-axis voltage setter 45. .

The d-axis and q-axis voltage setter 45 receives the d-axis current deviation Δid from the subtractor 46 and receives the q-axis current deviation Δiq from the subtractor 47 . The d-axis and q-axis voltage setter 45 calculates a d-axis voltage command value Vd ^* and a q-axis voltage command value Vq ^* such that each of the d-axis current deviation Δid and the q-axis current deviation Δiq approaches zero. The d-axis and q-axis voltage setter 45 outputs the calculated d-axis voltage command value Vd ^* and q-axis voltage command value Vq ^* to the position/speed estimator 44 and the dq/3φ coordinate converter 43.

The dq/3φ coordinate converter 43 receives the d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^* from the d-axis/q-axis voltage setter 45, and receives the phase angle θdq from the position/velocity estimator 44. Ru. The dq/3φ coordinate converter 43 converts the voltage vectors (Vd ^* , Vq ^* ) in the rotating coordinate system to the voltage vectors (Vu ^* , Vv ^* , Vw ^* ) in the fixed coordinate system using the phase angle θdq. .

That is, the dq/3φ coordinate converter 43 coordinately transforms the d-axis voltage command value Vd ^* and the q-axis voltage command value Vq ^* according to the motor currents iu, iv, and iw calculated by the 3φ current calculator 61. Three-phase voltage command values Vu ^* , Vv ^* , and Vw ^* are generated. The dq/3φ coordinate converter 43 outputs the generated three-phase voltage command values Vu ^* , Vv ^* , Vw ^* to the PWM device 41.

The PWM device 41 receives three-phase voltage command values Vu ^* , Vv ^* , Vw ^* from the dq/3φ coordinate converter 43, and receives the DC voltage Vdc from the DC voltage calculator 32. Further, the PWM device 41 receives a carrier signal, which is a carrier wave of PWM, and a carrier cycle from outside the motor control device 100 (for example, a higher-order controller of the motor control device 100). The PWM device 41 generates PWM signals Up to Wn based on the carrier signal, carrier period, voltage command values Vu ^* , Vv ^* , Vw ^* , and DC voltage Vdc.

For example, a triangular wave signal is input as a carrier signal to the PWM device 41, and the period of the triangular wave signal is input as the carrier period.

The PWM device 41 generates PWM signals Up to Wn by comparing each of the voltage command values Vu ^* , Vv ^* , and Vw ^* with the carrier signal. For example, the PWM device 41 compares the voltage command value Vu ^* as a comparison value with the carrier signal, and when the level of the carrier signal is higher than the voltage command value Vu ^* , turns on the U-phase PWM signal Up, and turns on the U-phase PWM signal Up. When the level of the carrier signal is smaller than the voltage command value Vu ^* as a comparison value, the U-phase PWM signal Up is turned off and the U-phase PWM signal Un is turned on. The generated PWM signals Up and Up are supplied to switching elements SWup and SWun, respectively. Similarly, the PWM device 41 generates V-phase PWM signals Vp, Vn according to the comparison result between the voltage command value Vv ^* as a comparison value and the carrier signal, and transmits the generated PWM signals Vp, Vn to the switching elements SWvp, SWvn. supply to each of them. Similarly, the PWM device 41 generates W-phase PWM signals Wp, Wn according to the comparison result between the voltage command value Vw ^* as a comparison value and the carrier signal, and transmits the generated PWM signals Wp, Wn to the switching element SWwp, SWwn.

Hereinafter, in a carrier signal that is a triangular wave signal, the apex on the mountain side may be referred to as a "mountain", and the apex on the valley side may be referred to as a "valley". Further, hereinafter, in one cycle of the carrier signal (that is, one carrier cycle), the range from the valley to the peak may be referred to as the "first half of the carrier," and the range from the peak to the valley may be referred to as the "second half of the carrier." Therefore, the lengths of the first half of the carrier and the second half of the carrier correspond to one-half of the carrier period.

For example, when comparing the voltage command value, which is the comparison value, and the triangular wave signal, which is the carrier signal, using the peak of the carrier signal as the reference (if the carrier signal is based on the peak), the peak of the carrier signal is the same as the carrier period. Set each carrier period so that it is centered. Then, the PWM device 41 is given voltage command values Vu ^* , Vv ^* , Vw ^* whose levels are constant within one carrier cycle, and the level of the carrier signal exceeds the voltage command values Vu ^* , Vv ^* , Vw ^* . At this point, the P-side PWM signals Up, Vp, and Wp are set to on-level, while the N-side PWM signals Un, Vn, and Wn are set to off-level, and the level of the carrier signal is set to voltage command values Vu ^* , Vv ^* , Vw. ^* When the value becomes below, the P-side PWM signals Up, Vp, and Wp are turned off, while the N-side PWM signals Un, Vn, and Wn are turned on. At this time, in each carrier cycle, the PWM signals Up, Vp, and Wp have a pattern that is symmetrical with respect to the peak of the carrier signal (see FIG. 3). Also, when the motor M is driving, the levels of the voltage command values Vu ^* , Vv ^* , Vw ^* change every carrier cycle, so when looking at adjacent carrier cycles, the PWM signals Up, Vp, Wp are as follows: The pattern becomes asymmetrical with respect to the valley of the carrier signal (see FIG. 3).

For example, if the comparison process between the voltage command value, which is the comparison value, and the triangular wave signal, which is the carrier signal, is performed using the valley of the carrier signal as a reference (if the carrier signal is based on the valley), the valley of the carrier signal is the carrier period. Set each carrier period so that it is in the center of . Then, the PWM device 41 is given voltage command values Vu ^* , Vv ^* , Vw ^* whose levels are constant within one carrier cycle, and the level of the carrier signal exceeds the voltage command values Vu ^* , Vv ^* , Vw ^* . At this point, the P-side PWM signals Up, Vp, and Wp are turned on, while the N-side PWM signals Un, Vn, and Wn are turned off, and the level of the carrier signal is set to the voltage command values Vu ^* , Vv ^* , Vw ^*. At the time when the voltage becomes below, the P-side PWM signals Up, Vp, and Wp are turned off, while the N-side PWM signals Un, Vn, and Wn are turned on. At this time, in each carrier period, the PWM signals Up, Vp, and Wp have a pattern that is symmetrical with respect to the valley of the carrier signal (not shown). Also, when the motor M is driving, the levels of the voltage command values Vu ^* , Vv ^* , Vw ^* change every carrier cycle, so when looking at adjacent carrier cycles, the PWM signals Up, Vp, Wp are as follows: The pattern is asymmetrical with respect to the peak of the carrier signal (not shown).

FIG. 2 is a diagram illustrating a configuration example of a 3φ current calculator according to Example 1 of the present disclosure. FIG. 3 is a diagram illustrating an example of the operation of the motor control device according to the first embodiment of the present disclosure. FIG. 4 is a diagram illustrating an example of the operation of the interpolation calculation unit according to the first embodiment of the present disclosure. In the following, a case where the carrier signal is based on a peak will be described as an example, but the technique described below can be similarly applied to a case where the carrier signal is based on a valley.

The 3φ current calculator 61 calculates motor currents iu, iv, and iw based on the bus current Is detected by the current detector 20 and the PWM signals Up to Wn generated by the generator 40.

For example, the 3φ current calculator 61 calculates the PWM signals Up to Wn in the first carrier cycle, the bus current Is detected in the second half of the carrier cycle in the first carrier cycle, and the bus current Is detected in the carrier cycle next to the first carrier cycle. Based on the PWM signals Up to Wn of a certain second carrier period and the bus current Is detected in the first half of the carrier of the second carrier period, at the boundary timing between the first carrier period and the second carrier period. Calculate motor current. The 3φ current calculator 61 outputs the calculated motor current to the generation unit 40, for example, within the second carrier period. The generation unit 40 converts the motor current input from the 3φ current calculator 61 into two-phase currents id and iq, and generates a two-phase current command based on the converted two-phase current and the speed command value ωm ^* . Two-phase voltage command values Vd ^* and Vq ^* are generated so that the current deviation between them and the values id ^* and iq ^* becomes zero. In addition, the generation unit 40 converts the two-phase voltage command values Vd ^* , Vq ^* into three-phase voltage command values Vu ^* , Vv ^* , Vw ^* in the carrier cycle next to the second carrier cycle. Three-phase voltage command values Vu ^* , Vv ^* , and Vw ^* to be used in a certain third carrier cycle are generated. Then, the generation unit 40 generates PWM signals Up to Wn based on the generated three-phase voltage command values Vu ^* , Vv ^* , Vw ^* and the DC voltage Vdc.

For example, the 3φ current calculator 61 calculates the states of the PWM signals Up to Wn of the carrier period Tc1, the bus current Is detected during the latter half period Tc1b of the carrier period Tc1, and the PWM signal of the carrier period Tc2 following the carrier period Tc1. Based on the states of the signals Up to Wn and the bus current Is detected during the first half period Tc2a of the carrier period Tc2, the motor currents iu, iv, and iw at the boundary timing tc12 between the carrier period Tc1 and the carrier period Tc2 are determined. calculate.

The 3φ current calculator 61 outputs the calculated motor currents iu, iv, and iw to the generation unit 40, for example, within the carrier period Tc2. The 3φ current calculator 61 outputs the calculated motor currents iu, iv, and iw to the generation unit 40, for example, within the second half period Tc2b of the carrier period Tc2. Further, the generation unit 40 generates voltage command values Vu ^* , Vv ^* , Vw ^* for the carrier cycle Tc3 following the carrier cycle Tc2 in the Tc2b period based on the motor currents iu, iv, iw at the boundary timing tc12. .

That is, the generation unit 40 converts the three-phase currents iu, iv, iw at the boundary timing tc12 into two-phase currents id, iq by the start timing of the carrier cycle Tc3, and calculates the converted two-phase currents and the speed. Two-phase voltage command values Vd ^* , Vq ^* are generated so that the current deviation from the two-phase current command values id ^* , iq ^* based on the command value ωm ^* becomes zero, and the two-phase voltage command value Vd By converting ^* , Vq ^* into three-phase voltage command values Vu ^* , Vv ^* , Vw ^* , three-phase voltage command values Vu ^* , Vv ^* , Vw ^* used for carrier cycle Tc3 are generated.

As shown in FIG. 2, the 3φ current calculator 61 includes an AD conversion section 61a, a first calculation section 61b, an interpolation calculation section 61c, and a second calculation section 61d.

The bus current Is detected by the current detection circuit 21 is input to the AD conversion section 61a. The AD conversion unit 61a converts the analog bus current Is into a digital value bus current Is by sampling the analog bus current Is, and converts the converted digital value bus current Is into the first calculation unit. 61b. The AD converter 61a obtains the bus current Is based on the PWM signals Up to Wn.

The first calculating section 61b receives the digital value of the bus current Is from the AD converting section 61a. The first calculation unit 61b calculates two-phase currents in the second half of the first carrier period from the PWM signal of the first carrier period and the bus current detected in the second half of the first carrier period. . Further, the first calculation unit 61b calculates the carrier of the second carrier period from the PWM signal of the second carrier period following the first carrier period and the bus current detected in the first half of the carrier of the second carrier period. Calculate the two-phase currents in the first half.

For example, as shown in FIG. 3, the first calculating unit 61b calculates the carrier current of the carrier period Tc1 from the PWM signals Up to Wn of the carrier period Tc1 and the bus current Is detected in the second half period Tc1b of the carrier period Tc1. Two-phase currents iu and iw in the second half period Tc1b are calculated.

For example, in the period Tc1b in the latter half of the carrier cycle Tc1, the PWM signals Up, Vp, and Wp are all on level in the period TP1, so the switching elements SWup, SWvp, and SWwp are all on, while the other switching elements The element is turned off. At this time, since no phase current flows, the bus current Is becomes zero.

In period TP2 following period TP1 in period Tc1b, PWM signal Up is at off level (PWM signal Un is on level) and PWM signals Vp, Wp are on level, so switching elements SWun, SWvp, SWwp are turned on. Then, the other switching elements are turned off. At this time, assuming that the direction from the inverter 10 toward the motor M is the "positive" direction of the current, a U-phase current "-Iu=Iv+Iw" flows as the bus current Is. The first calculation unit 61b calculates, for example, the bus current Is detected during the period TP2 as the U-phase current -Iu at the timing tu1 in the center of the period TP2.

In period TP3 following period TP2 in period Tc1b, PWM signals Up and Vp are at off level (PWM signals Un and Vn are on level), and PWM signal Wp is at on level, so switching elements SWun, SWvn, and SWwp are It is turned on while other switching elements are turned off. At this time, assuming that the direction from the inverter 10 toward the motor M is the "positive" direction of the current, the W-phase current "Iw=-Iu-Iv" flows as the bus current Is. The first calculation unit 61b calculates, for example, the bus current Is detected during the period TP3 as the W-phase current Iw at the central timing tw1 of the period TP3.

For example, as shown in FIG. 3, the first calculation unit 61b calculates the carrier period Tc2 from the PWM signals Up to Wn with the carrier period Tc2 and the bus current Is detected during the first half period Tc2a of the carrier period Tc2. The two-phase currents iu and iw in the first half period Tc2a of the carrier are calculated.

For example, in the period Tc2a of the first half of the carrier period Tc2, in the period TP4, the PWM signals Up and Vp are at the off level (the PWM signals Un and Vn are at the on level), and the PWM signal Wp is at the on level, so that the switching elements SWun, While SWvn and SWwp are turned on, other switching elements are turned off. At this time, assuming that the direction from the inverter 10 toward the motor M is the "positive" direction of the current, the W-phase current "Iw=-Iu-Iv" flows as the bus current Is. The first calculation unit 61b calculates, for example, the bus current Is detected during the period TP4 as the W-phase current Iw at timing tw2 at the center of the period TP4.

Further, in period TP5 following period TP4 in period Tc2a, PWM signal Up is at off level (PWM signal Un is on level), and PWM signals Vp and Wp are on level, so switching elements SWun, SWvp, and SWwp are turned on. while the other switching elements are turned off. At this time, when the direction from the inverter 10 toward the motor M is defined as the "positive" direction of the current, a U-phase current "-Iu=Iv+Iw" flows as the bus current Is. The first calculation unit 61b calculates, for example, the bus current Is detected during the period TP5 as the U-phase current -Iu at the timing tu2 at the center of the period TP5.

During period TP6 following period TP5 in period Tc2a, since the PWM signals Up, Vp, and Wp are on level, switching elements SWup, SWvp, and SWwp are turned on, while other switching elements are turned off. At this time, since no phase current flows, the bus current Is becomes zero.

The first calculation unit 61b outputs two-phase currents in the second half of the carrier period of the first carrier cycle and two-phase currents in the first half of the carrier period of the second carrier period to the interpolation calculation unit 61c. For example, the first calculation unit 61b outputs the U-phase current -Iu at timing tu1, the W-phase current Iw at timing tw1, the W-phase current Iw at timing tw2, and the U-phase current -Iu at timing tu2 to the interpolation calculation unit 61c. do.

The interpolation calculation unit 61c calculates the calculation results of the first calculation unit 61b, that is, the two-phase currents in the second half of the carrier period of the first carrier period and the two-phase currents in the first half of the carrier period of the second carrier period. It is input from the calculation unit 61b. For example, the interpolation calculation unit 61c inputs the U-phase current -Iu at timing tu1, the W-phase current Iw at timing tw1, the W-phase current Iw at timing tw2, and the U-phase current -Iu at timing tu2 from the first calculation unit 61b. be done.

The interpolation calculation unit 61c calculates the difference between the first carrier period and the second carrier period from the two-phase current in the second half of the carrier period of the first carrier period and the two-phase current in the first half of the carrier period of the second carrier period. Calculate the two-phase currents at the boundary timing.

For example, as shown in FIG. 3, the interpolation calculation unit 61c calculates the two-phase currents iu, iw in the second half period Tc1b of the carrier period Tc1 and the two-phase currents iu, iw in the first half period Tc2a of the carrier period Tc2. From this, the two-phase currents iu and iw at the boundary timing tc12 are calculated. For example, the interpolation calculation unit 61c calculates the U-phase current -Iu at the boundary timing tc12 from the U-phase current -Iu at the timing tu1 and the U-phase current -Iu at the timing tu2. Furthermore, the interpolation calculation unit 61c calculates the W-phase current -Iw at the boundary timing tc12 from the W-phase current Iw at the timing tw1 and the W-phase current Iw at the timing tw2.

At this time, as mentioned above, the levels of the voltage command values Vu ^* , Vv ^* , Vw ^* may change for each carrier cycle, so when looking at adjacent carrier cycles, the PWM signals Up, Vp, and Wp are the same as the carrier signal The pattern is asymmetrical with respect to the boundary timing, which is the trough. For example, as shown in FIG. 3, the PWM signals Up, Vp, Wp in the second half period Tc1b of the carrier period Tc1 and the PWM signals Up, Vp, Wp in the first half period Tc2a of the carrier period Tc2 are at boundary timings. The pattern is asymmetrical with respect to tc12.

If the interpolation calculation unit 61c calculates the U-phase current by simply averaging the U-phase current -Iu at timing tu1 and the U-phase current -Iu at timing tu2, the obtained U-phase current will be at the boundary timing. This results in a U-phase current at a timing shifted from tc12. Alternatively, if the interpolation calculation unit 61c calculates the W-phase current by simply averaging the W-phase current Iw at timing tw1 and the W-phase current Iw at timing tw2, the calculated W-phase current is calculated from the boundary timing tc12. This results in a W-phase current at a shifted timing. In other words, when the two-phase current is calculated by a simple average of the two-phase current in the second half of the carrier of the first carrier period and the two-phase current in the first half of the carrier of the second carrier period, the calculated two-phase current Since the currents are generated at different timings (see FIGS. 3 and 4), there is a possibility that errors in calculating the currents will increase.

Therefore, in this embodiment, the interpolation calculation unit 61c calculates the phase current by linear interpolation instead of calculating it by a simple average. That is, the interpolation calculation unit 61c calculates the two-phase current at the boundary timing between the two-phase current in the second half of the carrier of the first carrier period and the two-phase current in the first half of the carrier of the second carrier period. We use linear interpolation to do this.

For example, as shown in FIG. 3, the interpolation calculation unit 61c calculates the two-phase currents iu, iw in the second half period Tc1b of the carrier period Tc1 and the two-phase currents iu, iw in the first half period Tc2a of the carrier period Tc2. From this, the two-phase currents iu and iw at the boundary timing tc12 are calculated by linear interpolation. At this time, the interpolation calculation unit 61c uses the PWM signals Up to Wn to recognize the positions of the timings tu1, tw1, tw2, tu2 and the boundary timing tc12 on the time axis. For example, the interpolation calculation unit 61c calculates the U-phase current -Iu at the boundary timing tc12 from the U-phase current -Iu at the timing tu1 and the U-phase current -Iu at the timing tu2 by linear interpolation. For example, the interpolation calculation unit 61c calculates the W-phase current Iw at the boundary timing tc12 from the W-phase current Iw at the timing tw1 and the W-phase current Iw at the timing tw2 by linear interpolation. Note that the interpolation calculation unit 61c can perform the process of calculating the two-phase currents iu and iw at the boundary timing tc12 by linear interpolation within the second half period Tc2b of the carrier cycle Tc2.

For example, as shown in FIG. 4, if the U-phase current at timing tu1 is -Iu1 and the U-phase current at timing tu2 is -Iu2, the length of the period from timing tu1 to tc12 is Ti, and the length of the period from timing tc12 to tu2 is When the length of the period is Tiv, the U-phase current Iu12 at the boundary timing tc12 can be expressed by equation (1).
Iu12
= {(-Iu2)-(-Iu1)}×Ti/(Tiv+Ti)+(-Iu1)...(1)

Similarly, if the W-phase current at timing tw1 is Iw1 and the W-phase current at timing tw2 is Iw2, the length of the period from timing tw1 to tc12 is Tii, and the length of the period from timing tc12 to tw2 is Tiii. Then, the W-phase current Iw12 at the boundary timing tc12 can be expressed by equation (2).
Iw12
= {(+Iw2)-(+Iw1)}×Tii/(Tiii+Tii)+(+Iw1)…(2)

The interpolation calculation unit 61c outputs the two-phase currents at the boundary timing to the second calculation unit 61d. For example, the interpolation calculation unit 61c outputs the U-phase current Iu12 and the W-phase current Iw12 at the boundary timing tc12 to the second calculation unit 61d.

The second calculation unit 61d receives the two-phase currents at the boundary timing from the interpolation calculation unit 61c. For example, the second calculation unit 61d receives the U-phase current Iu12 and W-phase current Iw12 at the boundary timing tc12 from the interpolation calculation unit 61c. The second calculation unit 61d calculates the remaining one phase current from the interpolated two phase currents among the motor currents.

For example, in the case shown in FIG. 3, the second calculation unit 61d calculates the V-phase current Iv12 at the boundary timing tc12 from the U-phase current Iu12 and the W-phase current Iw12 at the boundary timing tc12 using the three-phase condition "Iu+Iv+Iw=0". calculate. That is, the second calculation unit 61d calculates the V-phase current Iv12 at the boundary timing tc12 according to equation (3). Note that the second calculation unit 61d can calculate the V-phase current Iv12 at the boundary timing tc12 within the second half period Tc2b of the carrier period Tc2.
Iv12=-Iu12-Iw12...(3)

The second calculation unit 61d outputs the calculated motor currents of Iv12, Iu12, and Iw12 to the 3φ/dq coordinate converter 42.

5 to 8 are diagrams illustrating operational examples of the motor control device according to the first embodiment of the present disclosure.

In the motor control device 100, the bus current Is is sampled at two points in the second half of the carrier period of the previous carrier cycle and at two points in the first half of the carrier period of the next carrier cycle with respect to the valley of the carrier signal. . The motor control device 100 stores the time, detected value, detected phase, and current sign in a memory (not shown) in association with each other for each of these sampling points. Then, the motor control device 100 calculates current values for two phases at the valley timing (boundary timing) of the carrier signal from the sampling data by linear interpolation. Then, the motor control device 100 calculates the current value for the remaining one phase from the current value for the two phases at the timing of the valley of the carrier signal.

For example, as shown in Figure 5, the bus current that occurs in the latter half of the carrier cycle before the boundary timing is
0 [A]
↓
Negative minimum voltage phase (U phase in the example shown)
↓
Maximum positive voltage phase (W phase in the example shown)
↓
0 [A]
becomes.

Among these, sampling is performed in each of the sections where a negative minimum voltage phase current (in the case of FIG. 5, the U-phase current) and a positive maximum voltage phase current (in the case of FIG. 5, the W-phase current) occur.

As shown in Figure 6, the bus current that occurs in the first half of the carrier period after the boundary timing is
0 [A]
↓
Maximum positive voltage phase (W phase in the example shown)
↓
Negative minimum voltage phase (U phase in the example shown)
↓
0 [A]
becomes.

Of these, sampling is performed in each of the sections where the maximum positive voltage phase current (in the case of FIG. 6, the W-phase current) and the minimum negative voltage phase current (in the case of FIG. 6, the U-phase current) occur.

At this time, the motor control device 100 sequentially assigns symbols i, ii, iii, and iv to the sampling points shown in FIG. The time of the point, the detected value, the detected phase, and the current code are associated and stored in a memory (not shown).

Then, the motor control device 100 calculates current values for two phases at the trough timing (boundary timing) of the carrier signal from the sampling data by linear interpolation. The motor control device 100 searches for data of the same detection phase from the data stored in association.

In the examples of FIGS. 7 and 8, i and iv are picked up as U-phases, and ii and iii are picked up as W-phases. The motor control device 100 calculates the value at the valley using the above equations (1) and (2) stored in a memory (not shown).

For example, the U phase is calculated using equation (1). In equation (1), Ti represents the time from timing i (tu1) to trough (tc12). Tiv represents the time from the trough (tc12) to timing iv (tu2).

Similarly, the W phase is calculated using equation (2). In equation (2), Tii represents the time from timing ii (tw1) to trough (tc12). Tiii represents the time from the trough (tc12) to timing iii (tw2).

The motor control device 100 calculates the current value for the remaining one phase from the current value for the two phases at the valley timing (boundary timing). For example, the motor control device 100 calculates the remaining V-phase current value from the two-phase current values calculated using equations (1) and (2). Since the two-phase current data calculated using equations (1) and (2) are temporally consistent as data at the valley timing, "Iu+Iv+Iw=0" holds true. Therefore, the remaining V-phase current value is calculated using equation (3) stored in a memory (not shown).

Here, as shown in the comparative example of FIG. 9, suppose that after calculating the currents of two phases within the current carrier cycle Tc902, the current value of another phase is calculated based on the current values of the two phases. think. In this case, even if it is attempted to use the calculated three-phase phase currents for control in the current next carrier cycle Tc903, it will not be in time. That is, in the comparative example shown in FIG. 9, the process of generating the voltage command value from the three-phase current values is performed in the current next carrier cycle Tc903, and the generated three-phase current command values are used in the current The next carrier cycle becomes Tc904. Therefore, the time lag between the cycle corresponding to the calculated three-phase phase currents (current carrier cycle Tc902) and the control cycle (next carrier cycle Tc904) becomes large. When the time lag between the period corresponding to the calculated three-phase phase currents and the control period becomes large, the accuracy of motor control decreases.

In contrast, in the present embodiment, in the motor control device 100, the calculation unit 60 calculates the PWM signal of the first carrier period, the bus current detected in the second half of the carrier of the first carrier period, and the first carrier current. The three-phase signal at the boundary timing between the first carrier period and the second carrier period is based on the PWM signal of the second carrier period following the period and the bus current detected in the first half of the carrier period of the second carrier period. Calculate the current. The generation unit 40 generates three-phase voltage command values in the third carrier period following the second carrier period, based on the three-phase currents at the boundary timing. Thereby, the next carrier cycle can be controlled using the current carrier cycle, which is a period corresponding to the calculated three-phase phase currents, and the voltage command value generated based on the phase currents. As a result, the time lag between the carrier period corresponding to the calculated three-phase phase currents and the carrier period in which control is performed using the phase currents can be reduced. As a result, the accuracy of motor control can be improved.

The first embodiment has been described above.

[Example 2]
In the following, differences from Example 1 described above regarding Example 2 of the present disclosure will be described.

FIG. 10 is a diagram illustrating a configuration example of a motor control device according to Example 2 of the present disclosure. In FIG. 10, the PWM device 41 is configured so that the time required for sampling in the AD converter 61a (hereinafter sometimes referred to as "required sampling time") is external to the motor control device 100 (for example, a higher-level host of the motor control device 100). controller). Further, the PWM device 41 generates a sampling start instruction (hereinafter sometimes referred to as an "AD conversion trigger") to the AD conversion unit 61a, and outputs the generated AD conversion trigger to the AD conversion unit 61a. In addition, the PWM device 41 sets a detection impossible flag indicating whether or not the motor current can be detected to “TRUE” (undetectable) or “FALSE” (detectable) based on the input to the PWM device 41, and The undetectable flag is output to the 3φ/dq coordinate converter 42.

Further, the PWM device 41 generates U-phase duties Duty_u, V, which are the duties of the voltage command value voltages Vu ^* , Vv ^* , Vw ^* of each phase with respect to the DC voltage Vdc, according to equations (4), (5), and (6). Phase duty Duty_v and W-phase duty Duty_w are calculated.
Duty_u=Vu ^* /Vdc…(4)
Duty_v=Vv ^* /Vdc…(5)
Duty_w=Vw ^* /Vdc…(6)

Further, the PWM device 41 controls the switching element SWup according to equations (7), (8), and (9) based on the duties Duty_u, Duty_v, and Duty_w and the carrier cycle Tc input from the outside of the motor control device 100. , SWvp, and SWwp, which are the on-times of U-phase on-time On_u, V-phase on-time On_v, and W-phase on-time On_w are calculated.
On_u=Tc・Duty_u…(7)
On_v=Tc・Duty_v…(8)
On_w=Tc・Duty_w…(9)

The PWM device 41 also uses on-times On_u, On_v, On_w, U-phase duty Duty_u, V-phase duty Duty_v, W-phase duty Duty_w, a carrier signal input from the outside of the motor control device 100, a carrier cycle, and sampling requirements. Based on the time, an AD conversion trigger and PWM signals Up, Un, Vp, Vn, Wp, and Wn are generated, and an undetectable flag is set. The PWM device 41 outputs a PWM signal Up whose on-level time is the U-phase on time On_u, a PWM signal Vp whose on-level time is the V-phase on time On_v, and a PWM signal Vp whose on-level time is the W-phase on time On_v. A PWM signal Wp whose time is On_w is generated.

FIG. 11 is a diagram illustrating a configuration example of a 3φ current calculator according to Example 2 of the present disclosure. In FIG. 11, the AD converter 61a converts the analog bus current Is into a digital value bus current Is by sampling the analog bus current Is according to the AD conversion trigger input from the PWM device 41. .

FIG. 12 is a diagram illustrating an example of a carrier signal according to Example 2 of the present disclosure. A triangular wave signal as shown in FIG. 12 is input to the PWM device 41 as a carrier signal. Further, the duty is associated with the amplitude of the carrier signal in FIG. 12, and when the amplitude of the carrier signal is set from 0 to 1, the duty is 0% when the comparison value is 1, and the duty is 100% when the comparison value is 0. .

FIG. 13 is a diagram illustrating an example of comparison processing according to Example 2 of the present disclosure. As shown in FIG. 13, the PWM device 41 generates PWM signals Up to Wn by comparing the comparison value and the carrier signal. Here, the PWM device 41 sets comparison values for each phase according to the U-phase duty Duty_u, the V-phase duty Duty_v, and the W-phase duty Duty_w. The comparison process between the comparison value and the carrier signal is performed for each phase: U phase, V phase, and W phase.

For example, if the comparison process is performed between the comparison value and the carrier signal using the peak as a reference (in other words, when the carrier signal is based on the peak), if a triangular carrier signal is generated at a predetermined carrier cycle, then the center of the carrier cycle becomes a mountain. Then, when the level of the carrier signal exceeds the comparison value, the PWM device 41 turns the PWM signals Up, Vp, and Wp on level, and turns the PWM signals Un, Vn, and Wn off level, so that the level of the carrier signal increases. When the value becomes equal to or less than the comparison value, the PWM signals Up, Vp, and Wp are turned off, while the PWM signals Un, Vn, and Wn are turned on. Therefore, as shown in FIG. 13, in the carrier period, the PWM signals Up, Vp, and Wp have a pattern that is symmetrical with respect to the peak (that is, the first half of the carrier and the second half of the carrier).

In addition, the PWM device 41 turns on the PWM signals Up, Vp, and Wp by increasing or decreasing the comparison value according to the on time of each phase expressed by equations (7), (8), and (9). Control the time you are in the level. That is, the PWM device 41 decreases the comparison value as the on time becomes longer, and increases the comparison value as the on time becomes shorter, thereby adjusting the on level expressed by equations (7), (8), and (9). A certain time generates the PWM signals Up, Vp, and Wp. Then, in accordance with the PWM signals Up, Vp, and Wp generated in this way, the inverter 10 generates a three-phase AC voltage, that is, a voltage applied to the U-phase terminal of the motor M (hereinafter referred to as "U-phase terminal voltage"). ), the voltage applied to the V-phase terminal of motor M (hereinafter sometimes referred to as "V-phase terminal voltage"), and the voltage applied to the W-phase terminal of motor M (hereinafter sometimes referred to as "W-phase terminal voltage") is generated. That is, in the inverter 10, the U-phase terminal voltage has a pulse width corresponding to the on-time On_u, the V-phase terminal voltage has a pulse width corresponding to the on-time On_v, and the W-phase terminal voltage has a pulse width corresponding to the on-time On_w. A terminal voltage is generated.

Below, the U-phase terminal voltage, the V-phase terminal voltage, and the W-phase terminal voltage may be collectively referred to as "terminal voltage." In addition, below, the pulse of the U-phase terminal voltage will be referred to as the "U-phase pulse," the pulse of the V-phase terminal voltage will be referred to as the "V-phase pulse," and the pulse of the W-phase terminal voltage will be referred to as the "W-phase pulse." There is. Further, the U-phase pulse, V-phase pulse, and W-phase pulse may be collectively referred to as "terminal voltage pulse."

FIG. 14 is a diagram illustrating an example of PWM signal shift processing according to Example 2 of the present disclosure.

As shown in FIG. 14, when the PWM signal before shifting is generated using the original comparison value, which is the comparison value before increase/decrease, the first half of the carrier of the original comparison value is decreased by α, and the carrier of the original comparison value is By increasing the latter half by α, the PWM signal is shifted to the left in the figure (that is, backward in time in the carrier period) by β corresponding to the magnitude of α, without changing the pulse width. . For example, when shifting the PWM signal to the left in the figure by 30% with respect to the time width of the first half of the carrier within the cycle of one carrier, the first half of the carrier of the original comparison value is decreased by a value corresponding to a duty of 30%. At the same time, the latter half of the carrier of the original comparison value may be increased by a value corresponding to a duty of 30%. Conversely, by increasing the first half of the carrier of the original comparison value by α and decreasing the second half of the carrier of the original comparison value by α, the PWM signal changes according to the magnitude of α without changing the pulse width. It shifts to the right in the figure (that is, forward in time in the carrier period) by β. As described above, since the terminal voltage is generated according to the PWM signal, when the PWM signal shifts, the pulse of the terminal voltage also shifts in the same way as the PWM signal shifts. In addition, the original comparison value within one carrier period can be decreased by α in the first half of the carrier and increased by the same amount α in the second half of the carrier, or increased by α in the first half of the carrier and the same magnitude in the second half of the carrier. By decreasing the value α by α, the duty within one carrier (Equations (4), (5), and (6)) is kept at the same value before and after shifting the PWM signal.

Here, for example, in the first embodiment described above, the AD converter 61a needs to perform sampling on the bus current Is four times within one carrier period (see FIG. 3). In Embodiment 1, for example, when the boundary timing is a valley timing, sampling is performed twice to calculate the current for two phases in the second half of the first carrier period, and the sample is sampled twice to calculate the current for two phases following the first carrier period. It is necessary to perform sampling twice to calculate the current for two phases in the first half of the carrier of the second carrier period. However, depending on the state of the terminal voltage pulse, it may not be possible to secure the time required for four samplings within one carrier period. Therefore, when the required sampling time cannot be secured, the PWM device 41 shifts the terminal voltage pulse by shifting the PWM signal as described below. In the following, examples of shifting the terminal voltage pulse will be described using four examples, shift examples 1 to 4.

<Shift example 1 (Figures 15 and 16)>
15 and 16 are diagrams for explaining shift example 1 according to Example 2 of the present disclosure. 15 and 16 show, as an example, a case where the PWM unit 41 performs two-phase modulation.

In FIG. 15, since the pulse width of the V-phase pulse and the pulse width of the W-phase pulse are equal in each of two consecutive carrier periods Tca1 and Tcb1, each of the V-phase pulse and the W-phase pulse has one carrier period. An example will be shown in which the difference between the pulse width of the V-phase pulse and the pulse width of the W-phase pulse is smaller than the required sampling time in a state in which the pulse width is symmetrical about the mountain.

In the state shown in FIG. 15, only a U-phase current with an inverted sign appears as a bus current in both carrier periods Tca1 and Tcb1. Therefore, in the state shown in FIG. 15, the 3φ current calculator 61 cannot calculate the motor current as in the first embodiment.

Therefore, as shown in FIG. 16, in the carrier period Tca1, the PWM device 41 needs to sample the respective appearance times of the bus current, which is the U-phase current, and the bus current, which is the V-phase current, whose signs are reversed in the latter half of the carrier. Until the time is reached, the V-phase pulse is shifted to the right in the figure, while the W-phase pulse is shifted to the left in the figure. In addition, as shown in FIG. 16, in the carrier cycle Tcb1, the PWM device 41 needs to sample the respective appearance times of the bus current, which is the V-phase current, and the bus current, which is the U-phase current, whose sign has been reversed in the first half of the carrier. Until the time is reached, the V-phase pulse is shifted to the left in the figure, while the W-phase pulse is shifted to the right in the figure.

This makes it possible for the AD converter 61a to secure two sampling times for the U-phase current iu and V-phase current iv calculated by the 3φ current calculator 61 in the second half of the carrier period Tca1. Therefore, the AD converter 61a can perform sampling twice for calculating the U-phase current iu and the V-phase current iv in the latter half of the carrier period Tca1. In addition, in the AD converter 61a, it is possible to secure two additional sampling times for the U-phase current iu and V-phase current iv calculated by the 3φ current calculator 61 in the first half of the carrier period Tcb1. Therefore, the AD converter 61a can perform sampling two more times for calculating the U-phase current iu and the V-phase current iv in the first half of the carrier period Tcb1. That is, by the PWM device 41 performing the shift processing as shown in FIG. 16, the U-phase current iu, the V-phase current iv, It becomes possible to secure a total of four sampling times for calculating the current for two phases of the W-phase current iw.

Therefore, in the second half of the carrier period Tca1 shown in FIG. 16, the PWM device 41 selects a timing T11 located temporally after the falling timing of the W-phase pulse by the necessary sampling time, and a falling timing of the V-phase pulse. An AD conversion trigger is generated at timing T12, which is located temporally backward by the required sampling time, and the generated AD conversion trigger is output to the AD conversion section 61a. Furthermore, in the first half of the carrier period Tcb1 shown in FIG. 16, the PWM device 41 generates an AD conversion trigger at timing T13, which is the rising timing of the V-phase pulse, and timing T14, which is the rising timing of the W-phase pulse. , outputs the generated AD conversion trigger to the AD conversion section 61a. Then, the AD conversion unit 61a starts sampling at each of timings T11, T12, T13, and T14 according to the AD conversion trigger.

Therefore, the 3φ current calculator 61 can calculate the motor current at the carrier boundary between the carrier period Tca1 and the carrier period Tcb1 as in the first embodiment.

<Shift example 2 (Figures 17 and 18)>
17 and 18 are diagrams for explaining shift example 2 according to Example 2 of the present disclosure. 17 and 18 show, as an example, a case where the PWM unit 41 performs two-phase modulation.

FIG. 17 shows that in each of two consecutive carrier periods Tca2 and Tcb2, the pulse width of the V-phase pulse is smaller than the pulse width of the W-phase pulse, and each of the V-phase pulse and the W-phase pulse is symmetrical with respect to the peak. An example will be shown in which the difference between the pulse width of the V-phase pulse and the pulse width of the W-phase pulse is smaller than the required sampling time.

In the state shown in FIG. 17, a W-phase current and a U-phase current whose sign is reversed appear as bus currents in both carrier periods Tca2 and Tcb2. However, in both carrier periods Tca2 and Tcb2, the time during which the W-phase current appears is less than the required sampling time. Therefore, in the state shown in FIG. 17, the 3φ current calculator 61 cannot calculate the motor current as in the first embodiment.

Therefore, as shown in FIG. 18, in the carrier cycle Tca2, the PWM device 41 determines that the time of appearance of the bus current, which is the W-phase current, in the first half of the carrier reaches the required sampling time, and the bus current, which is the V-phase current, in the second half of the carrier. The V-phase pulse is shifted to the right in the figure, while the W-phase pulse is shifted to the left in the figure, until the current appearance time reaches the required sampling time. In addition, as shown in FIG. 18, in the carrier period Tcb2, the PWM device 41 determines that the time of appearance of the bus current, which is the V-phase current, in the first half of the carrier reaches the sampling required time, and the bus current, which is the W-phase current, in the second half of the carrier. The V-phase pulse is shifted to the left in the figure, while the W-phase pulse is shifted to the right in the figure, until the current appearance time reaches the required sampling time.

As a result, in the AD converter 61a, it becomes possible to secure the required sampling time twice for the W-phase current iw and the V-phase current iv calculated by the 3φ current calculator 61 in the carrier cycle Tca2. The AD converter 61a can perform sampling twice for calculating the W-phase current iw and the V-phase current iv in the carrier period Tca2. In addition, in the AD converter 61a, it becomes possible to secure two additional sampling times in the carrier cycle Tcb2 for the W-phase current iw and the V-phase current iv calculated by the 3φ current calculator 61. The AD converter 61a can perform sampling two more times to calculate the W-phase current iw and the V-phase current iv in the carrier period Tcb2. In other words, by performing the shift processing as shown in FIG. 18 by the PWM device 41, the carrier period Tca2 and the carrier period Tcb2 correspond to two phases of the U-phase current iu, V-phase current iv, and W-phase current iw. It becomes possible to secure the time necessary for sampling four times in total for calculating the current.

Therefore, in the carrier period Tca2 shown in FIG. 18, the PWM device 41 has a timing T21 located temporally after the rising timing of the V-phase pulse by the required sampling time, and a timing T21 located temporally after the rising timing of the V-phase pulse by the required sampling time. An AD conversion trigger is generated at timing T22, which is located later in time by 100 kHz, and the generated AD conversion trigger is output to the AD converter 61a. Further, in the carrier cycle Tcb2 shown in FIG. 18, the PWM device 41 generates an AD conversion trigger at timing T23, which is the rising timing of the V-phase pulse, and timing T24, which is the falling timing of the V-phase pulse. The generated AD conversion trigger is output to the AD conversion section 61a. Then, the AD conversion unit 61a starts sampling at each of timings T21, T22, T23, and T24 according to the AD conversion trigger.

Therefore, the 3φ current calculator 61 can calculate the motor current at the carrier boundary between the carrier period Tca2 and the carrier period Tcb2 as in the first embodiment.

<Shift example 3 (Figures 19 and 20)>
19 and 20 are diagrams for explaining shift example 3 according to Example 2 of the present disclosure. 19 and 20 show, as an example, a case where the PWM unit 41 performs three-phase modulation.

In FIG. 19, in each of four consecutive carrier periods Tca3, Tcb3, Tcc3, and Tcd3, the pulse width of the V-phase pulse is smaller than the pulse width of the W-phase pulse, and the pulse width of the U-phase pulse is smaller than that of the V-phase pulse. The difference between the pulse width of the V-phase pulse and the pulse width of the W-phase pulse, and the pulse of the U-phase pulse, with each of the U-phase pulse, V-phase pulse, and W-phase pulse being symmetrical with respect to the peak. An example will be shown in which the difference between the pulse width of the V-phase pulse and the difference between the pulse width of the U-phase pulse and the W-phase pulse are both smaller than the required sampling time.

In the state shown in FIG. 19, a W-phase current and a U-phase current whose sign is reversed appear as bus currents in any of the carrier periods Tca3, Tcb3, Tcc3, and Tcd3. However, in any of the carrier periods Tca3, Tcb3, Tcc3, and Tcd3, both the appearance time of the W-phase current and the appearance time of the U-phase current whose sign has been reversed are less than the required sampling time. Therefore, in the state shown in FIG. 19, the 3φ current calculator 61 cannot calculate the motor current as in the first embodiment.

Therefore, as shown in FIG. 20, in the carrier period Tca3, the PWM device 41 transmits the U-phase pulse and the V-phase pulse in the figure until the appearance time of the bus current, which is the W-phase current, reaches the required sampling time in the latter half of the carrier. While shifting to the left, the W-phase pulse is shifted to the right in the figure. In addition, as shown in FIG. 20, in the carrier cycle Tcb3, the PWM device 41 transmits the U-phase pulse and the W-phase pulse in the figure until the appearance time of the bus current, which is the V-phase current, reaches the required sampling time in the latter half of the carrier. While shifting to the left, the V-phase pulse is shifted to the right in the figure. In addition, as shown in FIG. 20, in the carrier period Tcc3, the PWM device 41 transmits the U-phase pulse and the W-phase pulse in the figure until the appearance time of the bus current, which is the V-phase current, reaches the required sampling time in the first half of the carrier. While shifting to the right, the V-phase pulse is shifted to the left in the figure. In addition, as shown in FIG. 20, in the carrier cycle Tcd3, the PWM device 41 transmits the U-phase pulse and the V-phase pulse in the figure until the appearance time of the bus current, which is the W-phase current, reaches the required sampling time in the first half of the carrier. While shifting to the right, the W-phase pulse is shifted to the left in the figure.

As a result, the AD converter 61a secures two sampling times required for the W-phase current iw and the V-phase current iv calculated by the 3φ current calculator 61 in the carrier period Tca3 and the carrier period Tcb3, respectively. Therefore, the AD converter 61a can perform sampling twice in total for calculating the W-phase current iw and the V-phase current iv in the carrier cycle Tca3 and the carrier cycle Tcb3. In addition, the AD converter 61a secures two additional sampling times for the W-phase current iw and the V-phase current iv calculated by the 3φ current calculator 61 in the carrier period Tcc3 and the carrier period Tcd3, respectively. Therefore, the AD converter 61a can perform sampling twice in total for calculating the W-phase current iw and the V-phase current iv in the carrier cycle Tcc3 and the carrier cycle Tcd3. In other words, by performing the shift processing as shown in FIG. 20 by the PWM device 41, the U-phase current iu, V-phase current iv, and W-phase current iw are It becomes possible to secure the time required for a total of four samplings to calculate the currents for two of the phases.

Therefore, in the carrier period Tca3 shown in FIG. 20, the PWM device 41 generates an AD conversion trigger at a timing T31 located temporally after the falling timing of the W-phase pulse by the required sampling time, and performs the generated AD conversion. The trigger is output to the AD converter 61a. In addition, in the carrier period Tcb3 shown in FIG. 20, the PWM device 41 generates an AD conversion trigger at timing T32, which is located temporally after the falling timing of the V-phase pulse by the required sampling time, and converts the generated AD conversion trigger. The trigger is output to the AD converter 61a. Further, in the carrier period Tcc3 shown in FIG. 20, the PWM device 41 generates an AD conversion trigger at timing T33, which is the rising timing of the V-phase pulse, and outputs the generated AD conversion trigger to the AD conversion section 61a. Further, in the carrier period Tcd3 shown in FIG. 20, the PWM device 41 generates an AD conversion trigger at timing T34, which is the rising timing of the W-phase pulse, and outputs the generated AD conversion trigger to the AD conversion section 61a. Then, the AD conversion unit 61a starts sampling at each of timings T31, T32, T33, and T34 according to the AD conversion trigger.

Therefore, the 3φ current calculator 61 can calculate the motor current at the carrier boundary between the carrier period Tcb3 and the carrier period Tcc3 as in the first embodiment.

<Shift example 4 (Figures 19 and 21)>
FIG. 21 is a diagram for explaining shift example 4 according to Example 2 of the present disclosure. FIG. 21 shows, as an example, a case where the PWM unit 41 performs three-phase modulation. FIG. 21 shows the state of the terminal voltage pulse after the shift. In shift example 4, the state of the terminal voltage pulse before the shift is the same as shift example 3 (FIG. 19), so the description thereof will be omitted.

As shown in FIG. 21, in the carrier period Tca3, the PWM device 41 sends the U-phase pulse and the V-phase pulse to the left in the figure until the appearance time of the bus current, which is the W-phase current, reaches the required sampling time in the latter half of the carrier. At the same time, the W-phase pulse is shifted to the right in the figure. In addition, as shown in FIG. 21, in the carrier cycle Tcb3, the PWM device 41 outputs the U-phase pulse and the W-phase pulse in the figure until the appearance time of the bus current, which is the V-phase current, reaches the required sampling time in the first half of the carrier. While shifting to the right, the V-phase pulse is shifted to the left in the figure. In addition, as shown in FIG. 21, in the carrier period Tcc3, the PWM device 41 transmits the U-phase pulse and the W-phase pulse in the figure until the appearance time of the bus current, which is the V-phase current, reaches the required sampling time in the latter half of the carrier. While shifting to the left, the V-phase pulse is shifted to the right in the figure. In addition, as shown in FIG. 21, in the carrier cycle Tcd3, the PWM device 41 outputs the U-phase pulse and the V-phase pulse in the figure until the appearance time of the bus current, which is the W-phase current, reaches the required sampling time in the first half of the carrier. While shifting to the right, the W-phase pulse is shifted to the left in the figure.

As a result, the AD converter 61a secures two sampling times required for the W-phase current iw and the V-phase current iv calculated by the 3φ current calculator 61 in the carrier period Tca3 and the carrier period Tcb3, respectively. Therefore, the AD converter 61a can perform sampling twice in total for calculating the W-phase current iw and the V-phase current iv in the carrier cycle Tca3 and the carrier cycle Tcb3. In addition, the AD converter 61a secures two additional sampling times for the W-phase current iw and the V-phase current iv calculated by the 3φ current calculator 61 in the carrier period Tcc3 and the carrier period Tcd3, respectively. Therefore, the AD converter 61a can perform sampling twice in total for calculating the W-phase current iw and the V-phase current iv in the carrier cycle Tcc3 and the carrier cycle Tcd3. In other words, by performing the shift processing as shown in FIG. 21 by the PWM device 41, the U-phase current iu , V-phase current iv, and W-phase current iw, it is possible to secure a total of four sampling times for calculating the currents for two phases.

Therefore, in the carrier cycle Tca3 shown in FIG. 21, the PWM device 41 generates an AD conversion trigger at a timing T41 located temporally backward by the sampling required time from the falling timing of the W-phase pulse, and performs the generated AD conversion. The trigger is output to the AD converter 61a. Further, in the carrier period Tcb3 shown in FIG. 21, the PWM device 41 generates an AD conversion trigger at timing T42, which is the rising timing of the V-phase pulse, and outputs the generated AD conversion trigger to the AD conversion section 61a. Further, in the carrier period Tcc3 shown in FIG. 21, the PWM device 41 generates an AD conversion trigger at a timing T43 located temporally backward by the sampling required time from the falling timing of the V-phase pulse, and converts the generated AD The trigger is output to the AD converter 61a. Further, in the carrier period Tcd3 shown in FIG. 21, the PWM device 41 generates an AD conversion trigger at timing T44, which is the rising timing of the W-phase pulse, and outputs the generated AD conversion trigger to the AD conversion section 61a. Then, the AD conversion unit 61a starts sampling at each of timings T41, T42, T43, and T44 according to the AD conversion trigger.

Therefore, the 3φ current calculator 61 can calculate the motor current at the carrier boundary between the carrier period Tcb3 and the carrier period Tcc3 as in the first embodiment, similarly to shift example 3 (FIG. 20).

<PWM device processing>
The processing in the PWM device 41 will be described in more detail below. If the PWM device 41 shifts at least one of the three-phase terminal voltage pulses, the PWM device 41 can shift within two consecutive carrier periods (shift examples 1 and 2) or within four consecutive carrier periods (shift examples 3 and 4). Then, it is determined whether it is possible to secure the required sampling time for each different phase four times in total to calculate the current for two phases twice each. Here, various patterns can be considered for how to shift the terminal voltage pulse, and different terminal voltage pulses can be obtained depending on which of the three phases is shifted and by how much. Therefore, the PWM device 41 checks the patterns for shifting the terminal voltage pulses in a predetermined order, and determines whether the required sampling time can be secured for each pattern.

If the PWM device 41 determines that the required sampling time can be secured with any of the patterns, it shifts the terminal voltage pulses according to the pattern. Next, the PWM device 41 sets the detection impossible flag to "FALSE" and outputs the detection impossible flag set to "FALSE" to the 3φ/dq coordinate converter 42. Note that if the detection impossible flag input from the PWM unit 41 is set to "FALSE", the 3φ/dq coordinate converter 42 calculates the d-axis current id and the motor current calculated by the 3φ current calculator 61. Generate and output a q-axis current iq.

Next, the PWM device 41 generates an AD conversion trigger based on the shifted terminal voltage pulse, and outputs the generated AD conversion trigger to the AD conversion section 61a. The AD conversion unit 61a starts sampling when the AD conversion trigger is input from the PWM device 41.

On the other hand, if the PWM unit 41 determines that the required sampling time cannot be secured for all patterns, it determines that the required sampling time cannot be secured even if the terminal voltage pulses are shifted. Then, the PWM device 41 sets the detection impossible flag to "TRUE" and outputs the detection impossible flag set to "TRUE" to the 3φ/dq coordinate converter 42. In this case, the 3φ/dq coordinate converter 42 outputs the d-axis current id and the q-axis current iq in the most recent carrier cycle in which the motor current could be calculated.

The second embodiment has been described above.

As described above, the motor control device of the present disclosure (motor control device 100 of Example 2) includes an inverter (inverter 10 of Example 1), a current detection unit (current detection unit 20 of Example 1), and a (the calculation unit 60 of the second embodiment) and a generation unit (the generation unit 40 of the second embodiment). The inverter converts the DC voltage supplied from the DC power source (DC power source E _DC in the embodiment) into a three-phase AC voltage, and applies the converted AC voltage to the motor (motor M in the first embodiment). The current detection unit detects the bus current of the inverter using a resistor (shunt resistor Rs in the first embodiment) connected between the DC power supply and the inverter. The calculation unit calculates three-phase currents flowing through the motor based on the bus current of the inverter. The generation unit generates a PWM signal that controls the inverter by PWM using a carrier signal having a predetermined period. Further, the generation unit temporally shifts the PWM signal forward or backward within the period of the carrier signal. The calculation unit also calculates, based on the shifted PWM signal, a bus current detected in the second half of the first carrier period and a bus current detected in the first half of the second carrier period following the first carrier period. Using these, three-phase currents at the boundary timing between the first carrier period and the second carrier period are calculated. Then, the generation unit generates three-phase voltage command values for generating a PWM signal to be used in a third carrier period following the second carrier period, based on the three-phase currents at the boundary timing.

By doing this, even if it is not possible to calculate the three-phase current based on the PWM signal before shifting, the area in which the three-phase current can be calculated can be expanded by shifting the PWM signal, so the calculated The time lag between the carrier period corresponding to the three-phase current and the carrier period in which control is performed using the three-phase current can be reduced, and as a result, the accuracy of motor control can be improved.

Further, the calculation unit includes a first calculation unit (the first calculation unit 61b of the first embodiment), an interpolation calculation unit (the interpolation calculation unit 61c of the first embodiment), and a second calculation unit (the second calculation unit 61c of the first embodiment). 61d). The first calculation unit calculates the two-phase current in the second half of the first carrier period from the bus current detected in the second half of the first carrier period, and calculates the bus current detected in the first half of the second carrier period. The two-phase currents in the first half of the second carrier period are calculated from . The interpolation calculation unit calculates two-phase currents at the boundary timing from two-phase currents in the second half of the first carrier period and two-phase currents in the first half of the second carrier period. The second calculation unit calculates the remaining one phase current of the three phase currents from the two phase currents at the boundary timing calculated by the interpolation calculation unit.

By doing this, based on the bus current detected in the first carrier period and the bus current detected in the second carrier period, at the boundary timing between the first carrier period and the second carrier period, Three-phase currents can be calculated.

Further, the interpolation calculation unit calculates two-phase currents at the boundary timing by linear interpolation using the current value in the second half of the first carrier period and the current value in the first half of the second carrier period.

By doing this, even if the PWM signals before and after the boundary timing have an asymmetric pattern with respect to the boundary timing, the two-phase currents at the boundary timing between the first carrier period and the second carrier period can be calculated accurately. can do.

The calculation unit also calculates, based on the shifted PWM signal, the bus current detected in the first half and the second half of the first carrier period, and the bus current detected in the first half and the second half of the second carrier period following the first carrier period. The three-phase currents at the boundary timing between the first carrier period and the second carrier period may be calculated using the bus current obtained by the calculation. At this time, the generation unit generates three-phase voltage command values for generating a PWM signal to be used in a third carrier period following the second carrier period, based on the three-phase currents at the boundary timing.

By doing this, even if it is not possible to calculate the three-phase current based on the PWM signal before shifting, the area in which the three-phase current can be calculated can be further expanded by shifting the PWM signal, so the calculation The time lag between the carrier cycle corresponding to the three-phase currents and the carrier cycle in which control is performed using the three-phase currents can be reduced, and as a result, the accuracy of motor control can be improved.

The calculation unit also calculates, based on the shifted PWM signal, the bus current detected in the first carrier period, the bus current detected in the second carrier period following the first carrier period, and the bus current detected in the second carrier period following the first carrier period. The bus current detected in the third carrier period following the third carrier period and the bus current detected in the fourth carrier period following the third carrier period are used to calculate the second carrier period and the third carrier period. The three-phase currents at the boundary timing with the carrier period may be calculated. At this time, the generation unit generates three-phase voltage command values for generating the PWM signal used in the fifth carrier period following the fourth carrier period based on the three-phase currents at the boundary timing.

By doing this, even if it is not possible to calculate the three-phase current based on the PWM signal before shifting, the area in which the three-phase current can be calculated can be further expanded by shifting the PWM signal, so the calculation The time lag between the carrier cycle corresponding to the three-phase currents and the carrier cycle in which control is performed using the three-phase currents can be reduced, and as a result, the accuracy of motor control can be improved.

The calculation unit also calculates a bus current detected in the second half of the first carrier period and a bus current detected in the second half of the second carrier period following the first carrier period based on the shifted PWM signal. , the bus current detected during the first half of the third carrier period following the second carrier period, and the bus current detected during the first half of the fourth carrier period following the third carrier period. Three-phase currents at the boundary timing between the second carrier period and the third carrier period may be calculated. At this time, the generation unit generates three-phase voltage command values for generating the PWM signal used in the fifth carrier period following the fourth carrier period based on the three-phase currents at the boundary timing.

By doing this, the bus currents detected in the second half of the second carrier cycle and the first half of the third carrier cycle are temporally close to the boundary timing, so the two-phase currents at the boundary timing can be detected with higher accuracy. It can be calculated.

100 Motor control device 10 Inverter 20 Current detection section 21 Current detection circuit 40 Generation section 41 PWM device 60 Calculation section 61 3φ current calculator 61a AD conversion section 61b First calculation section 61c Interpolation calculation section 61d Second calculation section

Claims (7)

an inverter that converts a DC voltage supplied from a DC power source into a three-phase AC voltage and applies the AC voltage to a motor;
a current detection unit that detects a bus current of the inverter using a resistor connected between the DC power supply and the inverter;
a calculation unit that calculates three-phase currents flowing through the motor based on the bus current;
A generation unit that generates a PWM signal that controls the inverter by PWM using a carrier signal having a predetermined period,
The generation unit temporally shifts the PWM signal forward or backward within a period of the carrier signal,
When the calculation unit cannot secure the time required for a total of four samplings to calculate the current for two of the three phases of current within one cycle of the carrier signal, the calculation unit calculates the Based on the PWM signal, the bus current detected in the second half of the first carrier period and the bus current detected in the first half of the second carrier period following the first carrier period are used to determine the first carrier period. Calculating the currents of the three phases at the boundary timing between the carrier cycle and the second carrier cycle,
The generation unit generates three-phase voltage command values for generating the PWM signal to be used in a third carrier period following the second carrier period, based on the three-phase currents at the boundary timing.
Motor control device. The calculation unit is
The two-phase current in the second half of the first carrier period is calculated from the bus current detected in the second half of the first carrier period, and the second phase current is calculated from the bus current detected in the first half of the second carrier period. a first calculation unit that calculates two-phase currents in the first half of the second carrier cycle;
an interpolation calculation unit that calculates two-phase currents at the boundary timing from two-phase currents in the second half of the first carrier cycle and two-phase currents in the first half of the second carrier cycle;
a second calculation unit that calculates the remaining one-phase current of the three-phase currents from the two-phase currents at the boundary timing calculated by the interpolation calculation unit;
The motor control device according to claim 1. The interpolation calculation unit calculates two-phase currents at the boundary timing by linear interpolation using a current value in the second half of the first carrier period and a current value in the first half of the second carrier period.
The motor control device according to claim 2. an inverter that converts a DC voltage supplied from a DC power source into a three-phase AC voltage and applies the AC voltage to a motor;
a current detection unit that detects a bus current of the inverter using a resistor connected between the DC power supply and the inverter;
a calculation unit that calculates three-phase currents flowing through the motor based on the bus current;
A generation unit that generates a PWM signal that controls the inverter by PWM using a carrier signal having a predetermined period,
The generation unit temporally shifts the PWM signal forward or backward within the period of the carrier signal, and the current for two phases of the three-phase current is in the first carrier period and the second phase. In the second carrier period following the first carrier period, the phases are shifted so that they can be detected one by one, and the current for two of the three phases is transferred to the third carrier period following the second carrier period. Shifting the carrier cycle so that it becomes detectable one phase at a time in the first half of a fourth carrier cycle following the third carrier cycle,
The calculation unit calculates, based on the shifted PWM signal, a bus current detected in the first carrier period, a bus current detected in the second carrier period, and a bus current detected in the third carrier period. Using the detected bus current and the bus current detected in the first half of the fourth carrier period, calculate the three-phase current at the boundary timing between the second carrier period and the third carrier period. Calculate,
The generation unit generates three-phase voltage command values for generating the PWM signal to be used in a fifth carrier period following the fourth carrier period, based on the three-phase currents at the boundary timing.
Motor control device. The calculation unit calculates a bus current detected in the second half of the first carrier period and a bus current detected in the second half of the second carrier period following the first carrier period based on the shifted PWM signal. current, a bus current detected during the first half of a third carrier period following the second carrier period, and a bus current detected during the first half of a fourth carrier period following the third carrier period. calculating the three-phase currents at a boundary timing between the second carrier period and the third carrier period;
The motor control device according to claim 4 . The generation unit is configured to detect a first phase current in the first carrier period, a second phase current in the second carrier period, and detect the second phase current in the third carrier period. The current of either one phase or the second phase can be detected, and the current of the first phase and the second phase can be detected in the third carrier period in the first half of the fourth carrier period. Shifting the PWM signal such that a current of a phase different from the phase of the current that enables the detection of the current is detected.
The motor control device according to claim 4. The calculation unit performs a shift when the time required for a total of four samplings for calculating the current for two of the three phases of current within one cycle of the carrier signal is not secured. calculating the three-phase currents based on the subsequent PWM signal;
The motor control device according to claim 4.

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