JP7412933B2 - radar equipment - Google Patents

Description

Embodiments of the present invention relate to a radar device.

In order to deal with interference waves, radar equipment using the sub-array DBF method is equipped with an auxiliary antenna in addition to the sub-array antenna, and side lobe blanking (SLB) processing is performed using the signal obtained by the auxiliary antenna. There may be cases where When using an auxiliary antenna, the configuration of the radar device may become complicated and the manufacturing cost may increase. Furthermore, in radar devices using the subarray DBF method, weights are applied to reduce sidelobes in digital signal processing that combines signals from each subarray in order to suppress unnecessary waves such as clutter. Such weights include weights corresponding to directivity such as Taylor distribution. However, even when such weights are applied, it is not possible to completely remove signals arriving from the sidelobe direction, and the signal may be affected by clutter.

Yoshida, “Revised Radar Technology”, DBF (Digital Beam Forming), Institute of Electronics, Information and Communication Engineers, pp. 289-291, 1996 Yoshida, "Revised Radar Technology", Pulse Compression Radar, Institute of Electronics, Information and Communication Engineers, pp. 275-280, 1996 Yoshida, “Revised Radar Technology”, CFAR (Constant False Alarm Reception) Processing, Institute of Electronics, Information and Communication Engineers, pp. 87-89, 1996 Yoshida, “Revised Radar Technology”, Taylor Distribution, Institute of Electronics, Information and Communication Engineers, pp. 134-135, 1996 Yoshida, “Revised Radar Technology”, Monopulse method, Institute of Electronics, Information and Communication Engineers, pp. 260-264, 1996

The problem to be solved by the present invention is to provide a radar device that can suppress the influence of unnecessary waves such as interference waves arriving from side lobe directions and clutter reflections without using an auxiliary antenna.

The radar device of the embodiment includes an array antenna, a signal forming section, and a detecting section. The signal forming unit calculates a sum signal by adding received signals received by subarrays corresponding to each of four divided aperture planes obtained by dividing the aperture plane of the array antenna. The signal forming unit calculates a first addition signal by weighting and adding the received signals received by the sub-arrays of two of the four divided aperture planes using a weighting coefficient, A second summed signal is calculated by performing weighted addition of received signals received by subarrays of two divided aperture planes excluding two of the divided aperture planes. The signal forming unit weights and adds the received signals used to calculate the first addition signal by using a weighting coefficient different from the weighting coefficient used to calculate the first addition signal, and performs the third addition. A fourth addition is performed by weighting and adding the received signals used for calculating the second addition signal using a weighting coefficient different from the weighting coefficient used for calculating the second addition signal. Calculate the signal. The signal forming unit calculates a first difference signal that is the difference between the first addition signal and the second addition signal, and calculates a second difference signal that is the difference between the third addition signal and the fourth addition signal. Calculate the signal. When the sum signal is larger than the first difference signal and the second difference signal, the detection unit detects an object in the pointing direction of the array antenna based on the sum signal , and detects the object in the pointing direction of the array antenna. If the sum signal is smaller than either or both, the object is not detected .

FIG. 1 is a diagram showing an example of the configuration of a radar device according to the present embodiment. The figure which shows the example of a structure of the subarray antenna in this embodiment. The figure which shows the example of a structure of the A/D conversion unit in this embodiment. FIG. 3 is a diagram illustrating a configuration example of a signal processing section in this embodiment. FIG. 3 is a diagram showing an outline of each beam signal formed by the beam signal forming section in this embodiment. FIG. 3 is a diagram showing an example of a Σ beam signal (sum signal) and a ΔEL beam signal (difference signal) calculated by the beam signal forming unit in the present embodiment. FIG. 3 is a diagram showing an example of a Σ beam signal (sum signal) and an SLB ΔEL beam signal calculated by a beam signal forming unit in the present embodiment. FIG. 7 is a diagram showing an example of a Σ beam signal (sum signal) and two SLB ΔEL beam signals calculated by the beam signal forming unit in the present embodiment.

Hereinafter, a radar device according to an embodiment will be described with reference to the drawings. FIG. 1 is a diagram showing a configuration example of a radar device 100 according to this embodiment. The radar device 100 receives a reflected wave of a signal transmitted from a transmitting antenna (not shown) toward an observation space and reflected by an object (target), and detects the position of the object. A signal transmitted from a transmitting antenna is subjected to code modulation using a predetermined code sequence, linear frequency modulation in which the frequency is constantly changed within a predetermined frequency range, and the like.

The radar device 100 includes a phased array antenna 1 , a signal converter 2 , a clock generator 3 , a scan controller 4 , and a signal processor 5 . Phased array antenna 1 includes a plurality of subarray antennas 11. In the radar device 100, the signal processing unit 5 calculates a plurality of signals for SLB determination based on the digital IQ signal obtained from the signal received by the phased array antenna 1, and performs SLB determination using the calculated signals for SLB determination. Perform processing. By obtaining a plurality of SLB determination signals from the signals received by the phased array antenna 1, the radar device 100 can perform SLB processing without using an auxiliary antenna.

Each sub-array antenna 11 supplies a received RF signal obtained by reception directed toward the observation space to the signal converter 2 . Clock generator 3 supplies a reference clock signal to signal converter 2 and scan controller 4 . The signal converter 2 and the scan controller 4 operate in synchronization with the reference clock signal. The scan controller 4 supplies the phased array antenna 1 with a phase control signal for directing the pointing direction of the phased array antenna 1 toward the observation direction.

The signal converter 2 converts the received RF signals supplied from each sub-array antenna 11 into digital IQ signals in synchronization with the reference clock signal supplied from the clock generator 3. The signal conversion unit 2 includes an A/D (Analog/Digital) conversion unit 21 and a P/S (Parallel/Serial) conversion circuit 22 corresponding to each of the subarray antennas 11 included in the phased array antenna 1. The A/D conversion unit 21 directly samples and down-converts the received RF signal, and obtains a digital IQ signal by performing orthogonal detection on the down-converted signal. The A/D conversion unit 21 supplies the digital IQ signal to the P/S conversion circuit 22. The P/S conversion circuit 22 converts the digital IQ signal supplied from each A/D conversion unit 21 into a serial signal, and transmits the serial signal to the signal processing section 5.

The signal processing unit 5 uses each digital IQ signal transmitted as a serial signal to suppress the influence of unnecessary waves such as interference waves and clutter reflected waves arriving from the side lobe direction of the phased array antenna 1 toward the observation space. Perform processing. Further, the signal processing unit 5 detects the position of the object in the observation space based on the component of the reflected wave included in each digital IQ signal while suppressing the influence of unnecessary waves. The signal processing unit 5 may be configured using, for example, a general-purpose computer.

FIG. 2 is a diagram showing a configuration example of the subarray antenna 11 in this embodiment. The plurality of subarray antennas 11 included in the phased array antenna 1 have the same configuration. The sub-array antenna 11 includes a plurality of antenna elements 101, an amplifier 102 and a phase shifter 103 provided for each antenna element 101, a synthesis circuit 104, an amplifier 105, and an analog filter 106. A received signal received by antenna element 101 is supplied to amplifier 102 . Amplifier 102 amplifies the received signal and supplies it to phase shifter 103 . The phase shifter 103 changes the phase of the received signal by the amount of phase shift indicated by the phase control signal supplied from the scan controller 4. The amount of phase shift indicated by the phase control signal is determined according to the pointing direction of the phased array antenna 1. Phase shifter 103 supplies the phase-changed received signal to synthesis circuit 104 .

The synthesis circuit 104 is an analog circuit that adds and synthesizes the received signals supplied from each phase shifter 103, and supplies the synthesized signal obtained by the synthesis to the amplifier 105. In the additive synthesis in the synthesis circuit 104, addition is performed without applying different weighting to each received signal. Alternatively, weighted addition using a uniform weighting coefficient may be performed for each received signal. Amplifier 105 amplifies the composite signal and supplies it to analog filter 106 . The analog filter 106 transmits a signal in a predetermined frequency band near the frequency of the signal transmitted toward the observation space, and supplies the signal to the signal converter 2 as a received RF signal. Note that a high-pass filter or a band-pass filter may be used as the analog filter 106.

FIG. 3 is a diagram showing a configuration example of the A/D conversion unit 21 in this embodiment. The plurality of A/D conversion units 21 included in the signal conversion section 2 have the same configuration. The A/D conversion unit includes a PLL circuit 211, an A/D converter 212, and a quadrature detector 213. The PLL circuit 211 uses the reference clock signal supplied from the clock generator 3 to generate an A/D conversion clock signal used for sampling the received RF signal, and supplies it to the A/D converter 212. Further, the PLL circuit 211 uses the reference clock signal to generate an operation clock signal used for quadrature detection in the quadrature detector 213, and supplies the generated clock signal to the quadrature detector 213. The A/D conversion clock signal and the operation clock signal are synchronized, and the A/D converter 212 and quadrature detector 213 operate in synchronization.

The A/D converter 212 directly samples the received RF signal and performs digital down-conversion to convert it into a digital signal. The A/D converter 212 supplies the digital signal obtained by digital down-conversion to the quadrature detector 213. The quadrature detector 213 performs quadrature detection on the supplied digital signal to obtain a digital IQ signal including an in-phase component and a quadrature component. Quadrature detector 213 supplies the digital IQ signal to P/S conversion circuit 22 . Note that the quadrature detector 213 may be configured using a general-purpose device such as an FPGA (Field Programmable Gate Array).

FIG. 4 is a diagram showing a configuration example of the signal processing section 5 in this embodiment. The signal processing section 5 includes a communication I/F (interface) 51, a beam signal forming section 52, a pulse compression section 53, a noise threshold processing section 54, a CFAR (Constant False Alarm Ratio) processing section 55, and an SLB processing section. 56, a distance measuring section 57, and an angle measuring section 58. When a general-purpose computer is used as the signal processing unit 5, a CPU (Central Processing Unit) executes a program stored in a storage device such as a hard disk or an SSD (Solid State Drive). It operates as a pulse compression section 53, a noise threshold processing section 54, a CFAR processing section 55, an SLB processing section 56, a distance measurement section 57, and an angle measurement section 58.

The communication I/F 51 receives the digital IQ signal transmitted as a serial signal from the signal converter 2, and buffers the digital IQ signal. The beam signal forming unit 52 forms a plurality of beam signals using the digital IQ signals buffered in the communication I/F 51 at intervals at which signals are transmitted from the transmitting antenna toward the observation space. The beam signals formed by the beam signal forming section 52 include a Σ beam signal, a ΔEL beam signal, a ΔAZ beam signal, a ΔEL1 beam signal for SLB, and a ΔEL2 beam signal for SLB. The beam signal forming section 52 supplies each formed beam signal to the pulse compressing section 53.

FIG. 5 is a diagram showing an outline of each beam signal formed by the beam signal forming section 52 in this embodiment. In the example shown in FIG. 5, the aperture surface of the phased array antenna 1 is divided into four, and subarray antennas 11 included in each divided aperture surface are designated as subarrays A to D. Beam signal forming section 52 adds and combines the digital IQ signals of subarray antennas 11 included in each subarray A to D, and calculates a composite signal for each subarray A to D. Beam signal forming section 52 adds the combined signals of each subarray A to D to calculate a Σ beam signal (sum signal).

In addition, the beam signal forming unit 52 multiplies the digital IQ signals of the sub-array antennas 11 included in each of the sub-arrays A to D by weights (weighting coefficients), adds and synthesizes the signals, and calculates a weighted composite signal for each of the sub-arrays A to D. At least three weights are predetermined, one of which is a uniform weight for the digital IQ signal of each sub-array antenna 11. The uniform weight is, for example, "1". As other weights, for example, weights corresponding to the directivity of the Taylor distribution, weights corresponding to the directivity of the Baylis distribution, or the Chebyshev distribution may be used. Further, in the main lobe of the phased array antenna 1, other weighting coefficients may be used such that the amplitude (gain) of the signal obtained by weighted addition is smaller than the amplitude (gain) of the Σ beam signal.

When uniform weights are used, the beam signal forming unit 52 subtracts the sum of the weighted combined signals of subarray C and subarray D from the sum of the weighted combined signals of subarray A and subarray B, and generates a ΔAZ beam signal (difference signal ) is calculated. That is, the beam signal forming unit 52 calculates the difference in weighted composite signals between the right and left divided aperture planes obtained by dividing the aperture plane of the phased array antenna 1 in the vertical direction. In addition, the beam signal forming unit 52 subtracts the sum of the weighted composite signals of subarray B and subarray D from the sum of the weighted composite signals of subarray A and subarray C to calculate a ΔEL beam signal (difference signal). That is, the beam signal forming unit 52 calculates the difference in weighted composite signals between upper and lower aperture division planes obtained by dividing the aperture plane of the phased array antenna 1 in the horizontal direction.

Note that without performing calculations using weights, the beam signal forming unit 52 calculates the ΔAZ beam signal by subtracting the sum of the combined signals of subarray C and subarray D from the sum of the combined signals of subarray A and subarray B. You can. Furthermore, the beam signal forming unit 52 may calculate the ΔEL beam signal by subtracting the sum of the combined signals of sub-array B and sub-array D from the sum of the combined signals of sub-array A and sub-array C.

When weights other than uniform weights are used, the beam signal forming unit 52 subtracts the sum of the weighted composite signals of subarray C and subarray D from the sum of the weighted composite signals of subarray A and subarray B, and calculates ΔAZ for SLB. Calculate the beam signal. Furthermore, the beam signal forming unit 52 subtracts the sum of the weighted composite signals of subarray B and subarray D from the sum of the weighted composite signals of subarray A and subarray C to calculate the SLB ΔEL beam signal. That is, the beam signal forming unit 52 calculates the difference in weighted composite signals between upper and lower aperture division planes obtained by dividing the aperture plane of the phased array antenna 1 in the horizontal direction. The SLB ΔAZ beam signal and the SLB ΔEL beam signal are used as signals for SLB determination.

As described above, the beam signal forming unit 52 calculates the ΔAZ beam signal and the ΔEL beam signal, and simultaneously calculates the SLB ΔAZ beam signal and the SLB ΔEL beam signal for each weight. The beam signal forming unit 52 simultaneously forms a plurality of beam signals at each transmission interval from the transmitting antenna, that is, at each reception by the phased array antenna 1.

Here, the weighted addition combination performed by the beam signal forming section 52 will be explained. To simplify the explanation, the number of sub-array antennas 11 included in each divided aperture plane is assumed to be "5", and the digital IQ signals of sub-array antennas 11 included in each of sub-arrays A to D are expressed by equations (1-1) to (1). -4).
A=[a1,a2,a3,a4,a5]...(1-1)
B=[b1, b2, b3, b4, b5]...(1-2)
C=[c1,c2,c3,c4,c5]...(1-3)
D=[d1, d2, d3, d4, d5]...(1-4)

Furthermore, when expressed as weight W=[w1, w2, w3, w4, w5], the weighted composite signal is expressed as shown in equations (2-1) to (2-4).
A×W=Σai×wi, (i=1,2,...,5)...(2-1)
B × W = Σbi × wi, (i = 1, 2, ..., 5) ... (2-2)
C×W=Σci×wi, (i=1,2,...,5)...(2-3)
D×W=Σdi×wi, (i=1,2,...,5)...(2-4)

The ΔEL beam signal and the SLB ΔEL beam signal are expressed by equation (3).
ΔEL=((A×W)+(C×W))+((B×W)+(D×W))…(3)

Here, examples of each beam signal calculated by the beam signal forming section 52 are shown in FIGS. 6 to 8. 6 to 8, the horizontal axis indicates the angular difference with respect to the directivity direction of the phased array antenna 1, and the vertical axis indicates the amplitude or gain when the amplitude of the signal is normalized. Hereinafter, the angular range of the sidelobe of the Σ beam signal will be referred to as the "sidelobe range". FIG. 6 is a diagram showing an example of a Σ beam signal (sum signal) and a ΔEL beam signal (difference signal) calculated by the beam signal forming section 52 in this embodiment. Note that in the example shown in FIG. 6, "1" is used as the uniform weight used when calculating the ΔEL beam. The radar device 100 of this embodiment determines whether the signal received by the phased array antenna 1 is a signal included in the sidelobe range, and performs SLB processing.

FIG. 7 is a diagram showing an example of a Σ beam signal (sum signal) and an SLB ΔEL beam signal calculated by the beam signal forming unit 52 in this embodiment. In the example shown in FIG. 7, weights corresponding to the directivity of the Taylor distribution are used as weights used when calculating the SLB ΔEL beam signal. As shown in FIG. 7, in most of the sidelobe range, the amplitude of the SLB ΔEL beam signal is larger than the amplitude of the Σ beam signal. That is, by comparing the amplitude of the SLB ΔEL beam signal and the amplitude of the Σ beam signal, it can be determined whether the received signal is in the sidelobe range. However, near the null point where the amplitude of the ΔEL beam signal for SLB is almost zero, the magnitude relationship between the amplitude of the ΔEL beam signal for SLB and the amplitude of the Σ beam signal is reversed, so that the sidelobe range cannot be determined correctly.

FIG. 8 is a diagram showing an example of a Σ beam signal (sum signal) and two SLB ΔEL beam signals calculated by the beam signal forming unit 52 in this embodiment. In the example shown in FIG. 8, two weights are used when calculating the SLB ΔEL beam signal. The first weight is a weight corresponding to the directivity of the Taylor distribution. The second weight is a weight that selects two predetermined subarray antennas 11 from subarray antennas 11 included in subarrays A to D, and adds and synthesizes the digital IQ signals of the selected subarray antennas 11. For example, any two values among the elements of weight W=[w1, w2, w3, w4, w5] are set to "1", and the other values are set to "0". The null points of the two SLB ΔEL beam signals calculated in this way appear at different angles in the sidelobe range. By comparing the amplitudes of the two SLB ΔEL beam signals and the amplitude of the Σ beam signal, the sidelobe range can be correctly determined.

Note that the combination of weights shown in FIG. 8 is an example, and other combinations may be used. The conditions to be satisfied when selecting other combinations are as follows. The first condition is that in the main lobe of the phased array antenna 1, the amplitude (or gain) of each SLB ΔEL beam signal has a smaller value than the amplitude (or gain) of the Σ beam signal. The second condition is that the amplitude of any SLB ΔEL beam signal has a value larger than the amplitude of the Σ beam signal in the sidelobe range. The third condition is that the null points of the SLB ΔEL beam signals do not match in the sidelobe range.

As weights that satisfy the above conditions, weights corresponding to the directivity of Baylis distribution or Chebyshev distribution and digital IQ signals of n sub-array antennas 11 among sub-array antennas 11 included in sub-arrays A to D are added together (n-combined). ) may be used in combination with weights. "n" is a value smaller than the number of subarray antennas 11 included in subarrays A to D, and is a value such that the amplitude of the weighted composite signal is larger than the amplitude of the Σ beam signal in the side lobe of the Σ beam signal. Such “n” may be determined based on an actual value measured using the radar device 100 or a simulation result. Furthermore, if the above conditions are met, a weight corresponding to the directivity of the Taylor distribution and a weight corresponding to the directivity of the Baylis distribution or the Chebyshev distribution may be combined.

In this embodiment, the beam signal forming unit 52 supplies the difference between weighted composite signals using weights corresponding to the directivity of the Taylor distribution shown in FIG. 8 to the SLB processing unit 56 as the SLB ΔEL1 beam signal, The difference between the weighted composite signals using n composite weights is supplied to the SLB processing unit 56 as an SLB ΔEL2 beam signal. That is, a case will be described in which the beam signal forming unit 52 supplies the Σ beam signal, the ΔEL beam signal, the ΔAZ beam signal, the SLB ΔEL1 beam signal, and the SLB ΔEL2 beam signal to the pulse compression unit 53.

Returning to FIG. 4, the explanation of the configuration of the signal processing section 5 will be continued. The pulse compression unit 53 receives the Σ beam signal, the ΔEL beam signal, the ΔAZ beam signal, the SLB ΔEL1 beam signal, and the SLB ΔEL2 beam signal formed by the beam signal forming unit 52. Hereinafter, the SLB ΔEL1 beam signal and the SLB ΔEL2 beam signal will be referred to as "two SLB ΔEL beam signals." The pulse compression unit 53 calculates, for each received beam signal, a correlation value (pulse compression process) corresponding to the code modulation or linear frequency modulation used for the transmitted signal. The pulse compression section 53 supplies the Σ beam signal subjected to pulse compression processing to the noise threshold processing section 54 . The pulse compression unit 53 supplies the two SLB ΔEL beam signals subjected to pulse compression processing to the SLB processing unit 56. The pulse compression section 53 supplies the ΔEL beam signal and the ΔAZ beam signal, which have been subjected to pulse compression processing, to the angle measurement section 58.

The noise threshold processing unit 54 removes noise in the phased array antenna 1 included in the Σ beam signal supplied from the pulse compression unit 53. Noise is removed by setting a threshold based on the amplitude of the noise signal measured each time it is received, and removing signals below the threshold. The noise threshold processing unit 54 supplies the Σ beam signal from which noise has been removed to the CFAR processing unit 55. The CFAR processing unit 55 performs CFAR processing on the supplied Σ beam signal to determine whether the Σ beam signal contains a component of a reflected wave reflected by an object, that is, whether there is an object. When the CFAR processing section 55 determines that there is an object, it supplies the Σ beam signal to the SLB processing section 56 . If the CFAR processing unit 55 determines that there is no object, it rejects the Σ beam signal without outputting it.

The SLB processing unit 56 receives the Σ beam signal supplied from the CFAR processing unit 55 and the two SLB ΔEL beam signals supplied from the pulse compression unit 53. The SLB processing unit 56 compares the amplitude of the Σ beam signal with the amplitudes of the two SLB ΔEL beam signals. If the amplitude of the Σ beam signal is larger than the amplitudes of the two SLB ΔEL beam signals, the SLB processing unit 56 determines the Σ beam signal as a signal arriving from the main lobe direction of the Σ beam signal, and measures the Σ beam signal. It is supplied to the distance section 57 and the angle measurement section 58. On the other hand, if the amplitude of the Σ beam signal is smaller than either or both of the amplitudes of the two SLB ΔEL beam signals, the SLB processing unit 56 determines that the Σ beam signal is a signal arriving from the side lobe direction of the Σ beam signal. , reject the Σ beam signal. In this way, the SLB processing unit 56 determines whether to reject the Σ beam signal and rejects the Σ beam signal (SLB processing) by using the two SLB ΔEL beam signals according to the threshold value at the time of determination. used as

The distance measuring unit 57 calculates the distance to the object based on the Σ beam signal supplied from the SLB processing unit 56 and the signal transmitted from the transmitting antenna. The distance measuring section 57 outputs the calculated distance to an external device, or stores the distance in a storage section such as a memory provided in the signal processing section 5. The angle measurement section 58 uses a monopulse method as the angle measurement method, and calculates the reflection based on the Σ beam signal supplied from the SLB processing section 56 and the ΔEL beam signal and ΔAZ beam signal supplied from the pulse compression section 53. Determine the direction of arrival of waves. The angle measurement unit 58 specifies the azimuth angle (AZ angle measurement value) and the elevation angle (EL angle measurement value) as the direction of arrival. The angle measurement section 58 outputs the specified azimuth and elevation angle to an external device, or stores the azimuth and elevation angle in a storage section such as a memory provided in the signal processing section 5.

In the radar device 100, the beam signal forming unit 52 calculates a signal for SLB determination used in SLB processing from the signal received by the phased array antenna 1, so that the beam signal that arrives from the side lobe direction without using an auxiliary antenna is By rejecting unnecessary waves, the influence of unnecessary waves can be suppressed. Furthermore, since the radar device 100 does not require an auxiliary antenna, the configuration can be kept from becoming complicated and an increase in manufacturing costs can be suppressed.

In addition, in the radar device 100, the beam signal forming unit 52 calculates a plurality of ΔEL beam signals for SLB, and the SLB processing unit 56 uses the plurality of ΔEL beam signals for SLB for SLB processing, so that the direction of arrival is within the sidelobe range. It is possible to accurately determine whether or not it is included in the . Furthermore, by performing SLB processing using a plurality of SLB ΔEL beam signals, the radar device 100 can detect objects while suppressing the influence of unnecessary waves such as sea clutter and ground clutter.

Further, the sub-array antenna 11 of the radar device 100 generates an analog received RF signal by uniform additive synthesis without using weighting. By simplifying the configuration of the large number of sub-array antennas 11 included in the phased array antenna 1, the manufacturing cost of the radar device 100 can be suppressed and maintainability can be improved.

Furthermore, in the radar device 100, by using weights when calculating the ΔEL beam signals used for angle measurement, it is possible to calculate a plurality of ΔEL beam signals for SLB without complicating the calculation processing in the beam signal forming unit 52. I can do it.

Furthermore, in the case where a general-purpose computer is used as the signal processing section 5 in the radar device 100, the functions of the signal processing section 5 may be implemented in software, and the weights used for calculating the plurality of SLB ΔEL beam signals can be set in the radar device 100. 100 can be easily changed depending on the operating situation. Furthermore, by implementing it in software, the number of SLB ΔEL beam signals to be calculated simultaneously can be easily changed. Further, since there is no need to use a dedicated circuit or device as the signal processing section 5, the manufacturing cost of the radar device 100 can be reduced.

In addition, in the radar device 100, by directly sampling the received RF signal and converting it into a digital signal in the A/D conversion unit 21, a frequency converter for converting the received RF signal into a baseband signal is not required, and the digital IQ It is possible to suppress the mixing of unnecessary signals such as noise into the signal. Furthermore, since a frequency converter is not required, the manufacturing cost of the radar device 100 can be reduced.

In this embodiment, the beam signal forming unit 52 calculates two SLB ΔEL beam signals, and the SLB processing unit 56 uses the two SLB ΔEL beam signals to determine the sidelobe range. It is not limited to this. For example, the beam signal forming unit 52 may calculate three or more SLB ΔEL beam signals, and the SLB processing unit 56 may use the three or more SLB ΔEL beam signals to determine the sidelobe range. In this case, if the amplitude of the Σ beam signal is larger than the amplitude of all SLB ΔEL beam signals, the SLB processing unit 56 determines that the signal of the reflected wave arriving from the main lobe direction is included. On the other hand, if the amplitude of the Σ beam signal is smaller than the amplitude of at least one ΔEL beam signal for SLB, the SLB processing unit 56 determines that a reflected wave signal arriving from the side lobe direction is included, and rejects the Σ beam signal. do.

Furthermore, in this embodiment, the operation of determining the sidelobe range in the elevation direction using two SLB ΔEL beam signals has been described, but the SLB processing unit 56 uses two SLB ΔAZ beam signals to determine the sidelobe range in the elevation direction. SLB processing may be performed by determining the sidelobe range in the azimuth direction.

Furthermore, in this embodiment, a configuration has been described in which the SLB processing section 56 is provided between the CFAR processing section 55 and the ranging section 57, but the SLB processing section 56 is provided between the pulse compression section 53 and the noise threshold processing section 54 It may also be provided between the noise threshold processing section 54 and the CFAR processing section 55.

Further, the pointing direction of the phased array antenna 1 may be changed not only by controlling the phase shifter 103 by the scanning controller 4 but also by using a mechanical device that rotates or moves the phased array antenna 1.

According to at least one embodiment described above, the digital IQ signals (received signals) received by the subarrays A to D corresponding to each of the divided aperture planes obtained by dividing the aperture plane of the phased array antenna 1 are added together to form a Σ beam. A weighted composite signal is calculated for each different weight (weighting coefficient) by weighting and adding the digital IQ signals of subarrays A to D for each divided aperture surface, and a digital IQ signal for each divided aperture surface is calculated. A beam signal forming unit 52 (signal forming unit) calculates a ΔEL beam signal for SLB for each different weight as a difference between weighted composite signals obtained by weighted addition, and a Σ beam from each ΔEL beam signal for SLB calculated for each different weight. When the signal is large, the auxiliary antenna can be used by having a CFAR processing unit 55, a distance measurement unit 57, and an angle measurement unit 58 (detection unit) that detect objects in the pointing direction of the phased array antenna 1 based on the Σ beam signal. It can be determined whether the direction of arrival of the reflected wave is included in the sidelobe range without the need for this, and the influence of unnecessary waves such as interference waves and clutter reflected waves arriving from the sidelobe direction can be suppressed.

Although several embodiments of the invention have been described, these embodiments are presented by way of example and are not intended to limit the scope of the invention. These embodiments can be implemented in various other forms, and various omissions, substitutions, and changes can be made without departing from the gist of the invention. These embodiments and their modifications are included within the scope and gist of the invention as well as within the scope of the invention described in the claims and its equivalents.

DESCRIPTION OF SYMBOLS 1... Phased array antenna (array antenna), 2... Signal converter, 3... Clock generator, 4... Scanning controller, 5... Signal processing part, 11... Sub-array antenna, 21... A/D conversion unit, 22... P /S conversion circuit, 52... Beam signal forming section (signal forming section), 53... Pulse compression section, 54... Noise threshold processing section, 55... CFAR processing section (detection section), 56... SLB processing section, 57... Distance measurement section (detection section), 58... angle measurement section (detection section), 100... radar device, 101... antenna element, 102... amplifier, 103... phase shifter, 104... synthesis circuit, 105... amplifier, 106... analog filter, 211... PLL circuit, 212... A/D converter, 213... Quadrature detector

Claims (4)

array antenna,
A sum signal is calculated by adding the received signals received by the sub-arrays corresponding to each of the four divided aperture planes obtained by dividing the aperture plane of the array antenna, and A first summation signal is calculated by performing weighted addition of the received signals received by the sub-array using a weighting coefficient, and two divided apertures other than the two divided aperture planes among the four divided aperture planes are calculated. calculating a second summed signal by weighting and adding the received signals received by the sub-arrays of the plane; and calculating the first summed signal using the received signal used for calculating the first summed signal. A third addition signal is calculated by performing weighted addition by using a weighting factor different from the weighting factor used for the second addition signal, and the received signal used for calculating the second addition signal is added to the second addition signal. A fourth addition signal is calculated by performing weighted addition by using a weighting coefficient different from the weighting coefficient used to calculate the addition signal, and a fourth addition signal is calculated by the difference between the first addition signal and the second addition signal. a signal forming unit that calculates a first difference signal and calculates a second difference signal that is the difference between the third addition signal and the fourth addition signal;
When the sum signal is larger than the first difference signal and the second difference signal, an object in the pointing direction of the array antenna is detected based on the sum signal , and the first difference signal and the second difference signal are detected. a detection unit that does not detect the object when the sum signal is smaller than either or both of the difference signals ;
A radar device equipped with. The signal forming unit simultaneously calculates the first difference signal and the second difference signal each time the array antenna receives the signal.
The radar device according to claim 1. Each of the sub-arrays includes a plurality of antenna elements, a phase shifter provided for each antenna element, and an analog circuit that generates the received signal by adding and combining the outputs of the phase shifters.
The radar device according to claim 1 or 2. The weighting coefficient has a value such that the amplitude of either the first difference signal or the second difference signal is larger than the amplitude of the sum signal at any angle of the side lobe of the array antenna.
The radar device according to any one of claims 1 to 3.

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