JP7361648B2 - magnetic sensor device - Google Patents

Description

The present invention relates to a magnetic sensor device.

A magnetic sensor device using a Hall element requires a mechanism to adjust the magnetic field sensitivity to a desired value, and self-correction means have been proposed. A driving method is known in which the period of each of the two phases of the normal spinning operation is divided in half and allocated to the time for detecting the reference magnetic field signal, and the magnetic field sensitivity fluctuation of the measured magnetic field response signal is corrected according to the fluctuation. (See Patent Document 1). Furthermore, a spinning current method is known as a driving method for removing offset voltage in the absence of a magnetic field (see Patent Document 2).

Special Publication No. 2014-517919 Japanese Patent Application Publication No. 2008-309626

Patent Document 1 allocates half of the measurement target magnetic field response signal processing time to the reference magnetic field signal detection time due to time division processing, so the magnetic field detection time is less than when the entire time is used as the measurement target magnetic field response signal processing time. SN deteriorates. Further, Patent Document 1 requires a plurality of Hall elements, and mismatch between the Hall elements may result in a larger offset than that caused by the residual magnetic field.

The magnetic sensor device of the present invention includes a magnetic sensor circuit, a chopper modulation/demodulation circuit, a feedback circuit, a current input terminal, a voltage input terminal, and an output terminal, and the magnetic sensor circuits are connected singly or in parallel. The chopper modulation/demodulation circuit includes a first mixer whose output is connected to an amplifier, an amplifier whose output is connected to a second mixer, and a magnetic field generation circuit. The feedback circuit has a third mixer and a voltage-current conversion circuit, the current input terminal and the feedback circuit are connected to the magnetic sensor circuit, and the magnetic sensor circuit has a third mixer and a voltage-current conversion circuit. The voltage input terminal is connected to a chopper modulation/demodulation circuit, the chopper modulation/demodulation circuit is connected to the output terminal and the feedback circuit, and the voltage input terminal is connected to the feedback circuit.
Further, the magnetic sensor device of the present invention includes a magnetic sensor circuit, a chopper modulation/demodulation circuit, a feedback circuit, a current input terminal, a voltage input terminal, and an output terminal, and the magnetic sensor circuit is provided singly or in parallel. The chopper modulation/demodulation circuit includes a pair of connected Hall elements and a current supply circuit, and the chopper modulation/demodulation circuit includes a first mixer whose output is connected to an amplifier, an amplifier whose output is connected to a second mixer, the feedback circuit has a third mixer and a voltage-current conversion circuit, the current input terminal and the feedback circuit are connected to the magnetic sensor circuit, and the feedback circuit has a third mixer and a voltage-current conversion circuit; The voltage input terminal is connected to the chopper modulation/demodulation circuit, the chopper modulation/demodulation circuit is connected to the output terminal and the feedback circuit, and the voltage input terminal is connected to the feedback circuit.

An object of the present invention is to provide a magnetic sensor device that can constantly adjust magnetic field measurement sensitivity and output using one Hall element or a set of Hall elements connected in parallel.

FIG. 1 is a block diagram showing an example of a magnetic sensor device according to a first embodiment. FIG. 2 is a circuit diagram showing an example of a magnetic sensor circuit according to the first embodiment. It is a figure explaining the spinning current state of the Hall element concerning a first embodiment. FIG. 2 is a block diagram showing an example of a chopper modulation/demodulation circuit and a feedback circuit according to the first embodiment. FIG. 2 is a circuit diagram showing an example of a mixer circuit according to the first embodiment. 5 is a timing chart showing an example of each clock signal according to the first embodiment. It is a table showing an example of each clock signal according to the first embodiment. FIG. 1 is a circuit diagram showing an example of a low-pass filter (hereinafter referred to as LPF) according to the first embodiment. FIG. 3 is a circuit diagram showing another example of the LPF according to the first embodiment. FIG. 3 is a circuit diagram showing another example of the LPF according to the first embodiment. FIG. 3 is a circuit diagram showing another example of the LPF according to the first embodiment. FIG. 3 is a circuit diagram showing another example of the LPF according to the first embodiment. FIG. 2 is a circuit diagram showing an example of a gm amplifier according to the first embodiment. FIG. 3 is a diagram showing an example of a clock generator according to the first embodiment. FIG. 2 is a block diagram illustrating the operation of chopper modulation and demodulation on the I side according to the first embodiment. FIG. 2 is a block diagram illustrating the operation of chopper modulation and demodulation on the Q side according to the first embodiment. FIG. 3 is a diagram showing behavior in the frequency domain on the I side according to the first embodiment. FIG. 3 is a diagram showing behavior in the frequency domain on the Q side according to the first embodiment. FIG. 3 is a Bode diagram of a control loop according to the first embodiment. FIG. 2 is a block diagram showing an example of a digital circuit configuration of the feedback circuit according to the first embodiment. It is an example of a block diagram showing an example of a magnetic sensor device according to a second embodiment. FIG. 2 is a circuit diagram showing an example of a magnetic sensor circuit according to a second embodiment. It is a figure explaining the spinning current state of the Hall element concerning a second embodiment.

Hereinafter, the magnetic sensor device of the present invention will be explained with reference to the drawings.
(First embodiment)

Based on FIG. 1, a magnetic sensor device 1 according to a first embodiment of the present invention will be described. The magnetic sensor device 1 of the first embodiment includes a magnetic sensor circuit 2, a chopper modulation/demodulation circuit 3, a feedback circuit 4, a current Ibias input terminal, a reference voltage Vref input terminal, and five clock input terminals. and a voltage Vout output terminal. The magnetic sensor device 1 receives a current Ibias from a constant current source, a reference voltage Vref from a constant voltage source, and five clock signals to generate a signal corresponding to the magnetic field to be measured measured by the Hall element 21. is output from the voltage Vout output terminal.

The internal circuit of the magnetic sensor device 1 will be explained. The magnetic sensor circuit 2 measures the magnetic field applied to the Hall element 21 and outputs the measured signal to the chopper modulation/demodulation circuit 3. The chopper modulation/demodulation circuit 3 chopper-modulates, amplifies, and demodulates the input signal, and outputs a signal corresponding to the measured magnetic field to the voltage Vout output terminal. At the same time, the chopper modulation/demodulation circuit 3 branches the amplified signal and outputs it to the feedback circuit 4. The feedback circuit 4 generates a current Igm for adjusting the sensitivity of the Hall element 21 from the input signal and reference voltage Vref. The current Igm is added to the current Ibias and supplied to the magnetic sensor circuit 2.

The magnetic sensor circuit 2 that measures a magnetic field using the Hall element 21 will be described based on FIG. 2. The magnetic sensor circuit 2 includes a Hall element 21 that performs spinning current operation, a switch circuit 22, and a reference magnetic field generation circuit 23 that applies a reference magnetic field to the Hall element 21. The Hall element 21 is operated by the spinning current method by changing the connection with the switch circuit 22, and removes the offset voltage of the output signal when no magnetic field is applied. The switch circuit 22 has a current input terminal, a GND connection terminal, four Hall element connection terminals, a clock input terminal, and two output terminals.

An overview of the spinning current operation of the Hall element 21 is shown in FIG. The Hall element 21 has four terminals a to d, and is connected to the switch circuit 22. In the Hall element 21, when the spinning current state is 0, a current I is supplied to the terminal a and discharged from the terminal c. , the Hall element 21 generates a potential between the terminal b and the terminal d that is proportional to the magnetic field perpendicular to the Hall element 21 and proportional to the current I, and outputs it as a voltage Vb from the terminal b and a voltage Vd from the terminal d. do. When the Hall element 21 is in the spinning current state 1, a current I is supplied to the terminal b of the Hall element 21 and discharged from the terminal d. The Hall element 21 generates a potential proportional to the magnetic field perpendicular to the Hall element 21 and proportional to the current I, and outputs it as a voltage Va from a terminal a and a voltage Vc from a terminal c. Switching between the spinning current state 0 and the spinning current state 1 is performed by supplying the clock CPsp to the switch circuit 22. The switch circuit 22 outputs Vd to Va (hereinafter referred to as Vd/Va) from a first output terminal, and outputs Vb to Vc (hereinafter referred to as Vb/Vc) from a second output terminal. Although the Hall element 21 has been described using an example of one Hall element, it may be a plurality of Hall elements whose terminals are all connected in parallel, which is electrically equivalent to one Hall element. Parallel connection of a plurality of Hall elements is an effective method when a small Hall element lacks driving capability and there are restrictions on the shape of the space in which the Hall elements are installed.

The reference magnetic field generation circuit 23 will be further explained based on FIG. 2. The reference magnetic field generation circuit 23 includes a coil 24 , a constant current source 25 , inverters 26 and 27 , and a switch circuit 28 . Constant current source 25 is connected to switch circuit 28 . The switch circuit 28 has four switches 28-1 to 28-4, and is controlled by a clock CPplt supplied via inverters 26 and 27. The switch circuit 28 switches the direction of the current supplied from the constant current source 25 to the coil 24 in accordance with the clock CPplt. For example, when the clock CPplt is at high level H, the switches 28-1 and 28-2 are turned on, the first terminal of the constant current source 25 is connected to the first terminal of the coil 24, and the constant current source 25 is connected to the first terminal of the coil 24. A second terminal is connected to a second terminal of coil 24. When the clock CPplt is at low level L, the switches 28-3 and 28-4 are turned on, the first terminal of the constant current source 25 is connected to the second terminal of the coil 24, and the second terminal of the constant current source 25 is connected to the second terminal of the coil 24. is connected to a first terminal of the coil 24.

The reference magnetic field generation circuit 23 supplies the current generated by the constant current source 25 to the coil 24 to generate a reference magnetic field, and applies the magnetic field generated by the coil 24 to the Hall element 21. The coil 24 is arranged so that the generated magnetic field can be efficiently applied in the direction where the magnetic field sensitivity of the Hall element 21 is highest. The coil 24 may have any coil shape as long as it can efficiently generate a magnetic field. For example, if a large current can be supplied to the coil 24, the coil 24 may be a single conducting wire.

The reference magnetic field generated by the reference magnetic field generation circuit 23 is generated by supplying the clock CPplt to the switch circuit 28 via the inverters 26 and 27, and is an alternating current magnetic field whose magnitude is constant and the magnetic field direction alternately changes.

The reference magnetic field generation circuit 23 generates a reference magnetic field at the timing described below and applies it to the Hall element 21.

When the spinning current state is 0, the reference magnetic field generation circuit 23 switches the direction of the current supplied to the coil 24 using the clock CPplt, and switches the direction of the reference magnetic field applied to the Hall element 21 at a duty ratio of 50%. Similarly, in the spinning current state 1, the reference magnetic field generation circuit 23 switches the direction of the current supplied to the coil 24 using the clock CPplt, and switches the direction of the reference magnetic field applied to the Hall element 21 at a duty ratio of 50%. .

The magnetic sensor circuit 2 outputs signals Vd/Va and Vb/Vc output by the Hall element 21. The signal output by the magnetic sensor circuit 2 in response to the magnetic field applied from the reference magnetic field generation circuit 23 will be referred to as a pilot signal from now on.

The magnetic sensor circuit 2 outputs a signal in which a signal corresponding to the magnetic field to be measured passing through the Hall element 21 and a pilot signal are superimposed. Signals Vd/Va and Vb/Vc output by the magnetic sensor circuit 2 are output to the chopper modulation/demodulation circuit 3.

The chopper modulation/demodulation circuit 3 will be explained based on FIG. The chopper modulation/demodulation circuit 3 includes mixers 10 and 12, an amplifier 11, and an LPF circuit 13. Mixers 10, 12 are passive analog multipliers.

Signals Vd/Va and Vb/Vc output from the magnetic sensor circuit 2 are input to the mixer 10 of the chopper modulation/demodulation circuit 3. The output of mixer 10 is input to amplifier 11. The amplifier 11 uses a high input impedance amplifier using an FET or the like on the input side. The output of amplifier 11 is input to mixer 12 . The output of mixer 12 is input to LPF circuit 13. The output of the LPF circuit 13 is output from the voltage Vout output terminal. The flow of the signal from the mixer 12 to the LPF 13 circuit is called the in-phase component side (In-Phase side), hereinafter referred to as the I side.

The mixer 10 will be explained based on FIG. 5. Mixer 10 includes inverters 31 and 32, a switch circuit 33, an LO terminal, an input terminal, and an output terminal. A clock CPch0 signal is input to the LO terminal.

The input signals Vd/Va and Vb/Vc input to the mixer 10 are differential signals, and the positive side signal is denoted by MIXinP, the negative side signal is denoted by MIXinN, and the output signal is also a differential signal, and the positive side signal is denoted by MIXoutP, The negative side signal is indicated by MIXoutN. The mixer 12 has the same configuration as the mixer 10, and the clock CPch1 signal is input to the LO terminal. The output of the mixer 12 is input to the LPF 13, and only the low frequency component is output from the voltage Vout output terminal of the magnetic sensor device 1. The voltage Vout output from the voltage Vout output terminal is an output signal corresponding to the measured magnetic field applied to the Hall element 21.

The operation of the chopper modulation/demodulation circuit 3 will be explained. Signals Vd/Va and Vb/Vc output from the magnetic sensor circuit 2 are input to the mixer 10 of the chopper modulation/demodulation circuit 3. The input signal to the mixer 10 output from the magnetic sensor circuit 2 is a differential signal. The mixer 10 multiplies the differential input signal and the clock CPch0 signal by the switch circuit 33, and outputs the result as a differential output signal. The differential output signal output from the mixer 10 is input to the amplifier 11. The amplifier 11 amplifies the input differential input signal by a factor of A, and the amplified differential output signal is input to the mixer 12 .

The mixer 12 on the I side multiplies the differential signal output from the amplifier 11 and the clock CPch1 signal, and outputs a differential signal. The differential signal output from the mixer 12 is input to the LPF circuit 13, and the voltage Vout is output from the LPF circuit 13 to the voltage Vout output terminal. The clock CPch1 signal has the same frequency and the same phase as the clock CPch0 signal input to the mixer 10. The mixer 14 multiplies the differential signal output from the amplifier 11 and the clock CPch2 signal, and outputs a differential signal.

The LPF circuit 13 on the I side uses the response frequency band of the magnetic field to be measured as a passband, and eliminates switching noise of the chopping frequency caused by non-ideal operations such as delays in actual devices in chopper modulation and demodulation, and secondary noise of the magnetic field signal that arises in principle. Sideband waves are removed and a signal corresponding to the measured magnetic field is output to the voltage Vout output terminal.

FIG. 8 is a circuit example of the LPF circuit 13. The LPF circuit 13-1 in FIG. 8 includes a balanced/unbalanced conversion circuit 17 that converts a differential signal, which is a balanced signal, into an unbalanced signal, and a tertiary active LPF 13-2. Since the balanced/unbalanced conversion circuit 17 and the tertiary active LPF 13-2 have common circuit configurations, their explanations will be omitted. FIG. 9 is an example of another tertiary active type LPF circuit 13. The example LPF circuit in FIG. 9 includes a tertiary active LPF 13-3 that receives differential signals and outputs differential signals. When selecting differential signal output as the output signal, select this LPF circuit. The tertiary active type LPF 13-3 has a general circuit configuration, so a description thereof will be omitted.

The feedback circuit 4 will be explained based on FIG. Feedback circuit 4 includes a mixer 14, an LPF circuit 15, and a gm amplifier 16. Mixer 14 is a passive analog multiplier having the same circuit configuration as mixer 10. The gm amplifier 16 is a transconductance amplifier that outputs a current proportional to the potential difference between input terminals.

The output of the amplifier 11 and the clock CPch2 are input to the mixer 14. The output of mixer 14 is input to LPF circuit 15. The output of the LPF circuit 15 is input to the gm amplifier 16 as a voltage Vplt. The gm amplifier 16 converts the input voltage into a current and outputs the current. The signal flow from the mixer 14 to the LPF circuit 15 is called the quadrature phase component side (Quadrature side), and hereinafter expressed as the Q side.

The Q-side mixer 14 multiplies the differential signal output from the amplifier 11 and the clock CPch2 signal, and outputs the differential signal to the LPF circuit 15. The LPF circuit 15 is set to a very low cutoff frequency (for example, 10 Hz), separates a DC component (denoted as voltage Vplt) from the pilot signal output from the mixer 14, and outputs it to the gm amplifier 16.

The voltage Vplt output from the Q-side LPF circuit 15 corresponds to the magnetic field applied to the Hall element 21 from the reference magnetic field generation circuit 23. The gm amplifier 16 compares the voltage Vplt with a reference voltage Vref input from the outside, converts the comparison result into a current Igm, and outputs the current Igm. The gm amplifier 16 converts the error voltage between the voltage Vplt corresponding to the reference magnetic field and the reference voltage Vref, which is a control target, into an error current Igm and outputs the error current Igm.

To summarize the operation of the feedback circuit 4, the feedback circuit 4 demodulates the pilot signal from the output of the amplifier 11 using the mixer 14, compares the DC component of the pilot signal with an external reference voltage using the gm amplifier 16, and converts the comparison result into a current. Convert and output.

FIG. 10 is a circuit example of the LPF circuit 15. The LPF circuit 15-1 in FIG. 10 includes a balanced passive LPF 15-2 and a balanced-unbalanced conversion circuit 17. FIG. 11 shows another circuit example of the LPF circuit 15. The LPF circuit 15-3 in FIG. 11 includes a balanced/unbalanced conversion circuit 17 and an unbalanced passive LPF 15-4. FIG. 12 is a circuit example of an unbalanced gmC filter type LPF 15-5 that can be substituted for the unbalanced passive type LPF 15-4. The balanced passive type LPF 15-2, the unbalanced passive type LPF 15-4, and the unbalanced gmC filter type LPF 15-5 have common circuit configurations, so a description thereof will be omitted.

FIG. 13 is a circuit example of the gm amplifier 16. The gm amplifier 16 inputs an external reference voltage Vref to its In+ terminal, inputs the voltage Vplt output by the LPF circuit 15 to its In- terminal, and outputs a current Igm proportional to the voltage difference between the In+ and In- terminals. . Since the gm amplifier 16 has a general circuit configuration, a description thereof will be omitted.

FIG. 6 shows a timing chart of the clock CPplt signal, clock CPsp signal, clock CPch0 signal, clock CPch1 signal, and clock CPch2 signal, and FIG. 7 shows the frequency ratio and phase. In FIG. 6, the vertical axis represents logic level and the horizontal axis represents time. The clock CPsp signal, the clock CPch0 signal, and the clock CPch1 signal have the same frequency and the same phase. The clock CPch2 signal has the same frequency as the clock CPsp signal, but has a phase difference of +90°. The clock CPplt signal has twice the frequency of the clock CPsp signal. For example, the frequency of the clock CPsp signal is 2 MHz, and the frequency of the clock CPplt signal is 4 MHz. The five clock signals are clock signals that control the corresponding switch circuits, and the signal levels are such that the switch circuits can be controlled in accordance with the logic level. FIG. 14 is a circuit example of the clock generation circuit 18 that generates a clock signal. Since this is a general circuit, its explanation will be omitted.

The current Igm output from the feedback circuit 4 is added to the current Ibias supplied from an external constant current source, and is fed back to the magnetic sensor circuit 2 as a current I. Since the output voltage of the Hall element 21 in the magnetic sensor circuit 2 is proportional to the applied magnetic field and the current I, the sensitivity of the magnetic sensor circuit 2 can be adjusted by the current Igm.

To summarize the operation of the first embodiment, the signal flow is as follows. The magnetic sensor circuit 2 outputs to the chopper modulation/demodulation circuit 3 a signal in which a signal corresponding to the magnetic field to be measured that is proportional to the current I is superimposed with a pilot signal that is also proportional to the current I. The chopper modulation/demodulation circuit 3 chopper modulates the input signal with a mixer 10, amplifies it with an amplifier 11, demodulates a signal corresponding to the magnetic field to be measured with a mixer 12 and an LPF 13, and outputs the demodulated signal (I side). The feedback circuit 4 demodulates the DC component (voltage Vplt) of the pilot signal from the output of the amplifier 11 using the mixer 14 and LPF 15 (Q side), converts the difference from the reference voltage Vref into a current Igm using the gm amplifier 16, and outputs it. do. The current Igm is added to the current Ibias to obtain a current I, and the current I is fed back to the magnetic sensor circuit 2.

Next, a mechanism for separating a signal corresponding to the magnetic field to be measured (I side) and a pilot signal (Q side) that have been simultaneously subjected to chopper modulation and demodulation processing will be explained.

FIG. 15 is a block diagram schematically showing the main path chopper modulation/demodulation operation. Here, A is the amplification factor, x(t) is the signal corresponding to the magnetic field to be measured, y(t) is the output signal, z(t) is the output signal of the amplifier 11, p(t) is the pilot signal, and w( t) is a rectangular carrier wave and corresponds to clock CPch0 and clock CPch1, and wc/2π is a chopper modulation frequency. The signal x(t) corresponding to the magnetic field to be measured and the pilot signal p(t) are multiplied by a rectangular carrier wave w(t), amplified by A times, and multiplied again by w(t) to become y(t).

The rectangular wave w(t) is expressed as a Fourier series and can be written as in equation (4).

When w(t)^2 is calculated, it becomes equation (5), and when it is summarized using the product-sum formula of trigonometric functions, it becomes equation (6).

When formula (6) is rearranged term by term, it becomes formula (7).

If n → ∞ and m → ∞ in equation (7), then equations (8) and (9) are obtained.

From equations (8) and (9), an output y(t) obtained by linearly amplifying x(t)+p(t) is obtained as shown in equation (10).

Since p(t) is assumed to be outside the signal band corresponding to the magnetic field to be measured, x(t) may be discriminated by an LPF after this. Similarly, FIG. 16 is a block diagram schematically showing the chopper modulation/demodulation operation on the negative feedback side. Here, v(t) is a rectangular carrier wave (+90°) and corresponds to the clock CPch2. The input/output relationship is described as follows.

Further, the LO input for demodulation inputted to the mixer 14 is a rectangular wave v(t) advanced by 90° with respect to w(t), and is expressed by equation (15).

Here, when w(t)·v(t) is calculated, it becomes as shown in equation (16), and when summarized, it becomes as shown in equation (17).

When formula (17) is aligned for each term and 0 is complemented, formula (18) is obtained.

Equation (19) is obtained by rewriting the coefficients of Equation (18) for each term into Leibniz series and its odd fraction.

Equations (18) and (19) can be seen as equivalent since the error converges to 0 as the coefficients of each term are accumulated downward. Therefore, equation (19) is factorized to obtain equation (20).

That is, the multiplication of the rectangular waves w(t) and v(t) results in v(2t), which is obtained by compressing the time axis t by half. Since the pilot signal p(t) is originally assumed to be a rectangular wave that is twice the chopper modulation frequency advanced by 90 degrees and generated from the reference magnetic field generation circuit 23, if the Hall element sensitivity coefficient is B, then the following equation is obtained.

Although the process is omitted, the right side of equation (22) is calculated in the same way as w(t)^2 and becomes 1.

Therefore, as shown on the right side of equation (24), the output y(t) includes a sideband wave obtained by up-converting x(t) linearly amplified by A times to twice the chopper frequency, and a DC signal multiplied by AB. A signal appears.

The above considerations will be extended to the frequency domain as well. The Fourier transform of each time signal is expressed as follows.

Furthermore, since the delta function can be used to write equations (29) and (30), equations (31) and (32) can be obtained from equations (14) and (15).

17 and 18 show the angular frequency on the X-axis, the imaginary number on the Y-axis, and the real number on the Z-axis, with the Z-axes superimposed. Each signal processing process of the feedback signal is expressed. The spectrum of the signal corresponding to the magnetic field to be measured is assumed to be X(ω), which exists only on the positive side of the cosine phase. The spectral sequence P(ω) of the pilot signal p(t) is a rectangular wave with a sine phase that is twice the chopper frequency ωc and advanced by 90°, so 2ωc is decomposed into the fundamental wave and its odd harmonics, and Y It is placed on the axis. Since the chopper modulated carrier wave W(ω) has a cosine phase and formula (31) becomes a spectral sequence, chopper modulation is performed by independently converting the frequency of X(ω) + P(ω) using the same spectral sequence and adding them together to obtain each A. The multiplied value becomes the chopper modulated wave Z(ω). The steps up to this point are common to both the I side and Q side.

Regarding chopper demodulation on the I side in Figure 17, the chopper demodulated carrier wave is the same W(ω) as during modulation, so if frequency convolution is performed in the same way as frequency conversion during modulation, a spectrum like Y(ω) will be obtained. It can be estimated. Originally, in the time domain, the final result is equation (10) in which the output signal y(t) has no relation to the chopper modulated carrier wave w(t), so combining it with the following equation (33) yields equation (34).

Therefore, when formula (34) is Fourier transformed, formula (35) is obtained, which coincides with Y(ω) shown in the figure.

Since V(ω/2) in equation (35) is obtained by expanding the frequency axis of V(ω) by twice, it can be easily separated from the band of X(ω). That is, it can be seen that on the I side, by adding an LPF of the required band, the influence of the pilot signal p(t) superimposed in advance can be removed. Regarding the chopper demodulation on the Q side in FIG. 18, since the chopper demodulated carrier wave is V(ω) with a sine phase, Y(ω) can be estimated by similar convolution. Similarly, when formula (24) is Fourier transformed, formula (36) is obtained.

As a result, a DC signal derived from a pilot signal that has undergone the same chopper modulation signal processing as the magnetic field response signal to be measured appears on the Q side.

From the above, the Q-side node, that is, the voltage Vplt in FIG. 1, can be used as a monitor of the magnetic field sensitivity that depends on the electrical characteristics of the main path, that is, from the magnetic sensor circuit 2 in FIG. 1 to the mixer 10 and amplifier 11. 12 and the mixer 14 have the same characteristics, and if the characteristics of the LPF 13 and the LPF 15 are sufficiently predictable, it can be seen that the magnetic field sensitivity can be monitored depending on the electrical characteristics up to the output of the voltage Vout.

Note that it is desirable that the operating speed of the amplifier 11 be significantly higher than the chopper demodulation frequency, but if phase lag is a factor, an equivalent delay should be added to the chopper demodulation clock path of the LO inputs of the mixer 12 and mixer 14 by connecting multiple inverters, etc. All you have to do is insert a delay circuit to generate it. Further, if the spinning offset of the Hall element 21 of the main body is larger than expected, or if the DC offset of the amplifier 11 cannot be ignored, a high-pass filter HPF may be inserted before and after the amplifier 11.

Next, a control method using the magnetic field sensitivity monitor, that is, the voltage Vplt on the Q side in FIG. 1 will be described. From the above description, the baseband voltages of voltage Vout and voltage Vplt exclude the influence of the mixer, so the magnetic sensor device 1 in FIG. 1 can be regarded as a linear system. Here, the output of the reference magnetic field generator 23 is B [T], the magnetic field sensitivity of the Hall element 21 is S [V/AT],
Assuming that the LPF 15 is first-order, the transfer function is 1/(sτ1+1), and the transconductor of the gm amplifier 16 is C(s) [A/V], the loop transfer from the reference voltage Vref to the voltage Vplt in the loop including the feedback circuit 4 is performed. The function G(s) is expressed as equation (37).

The transfer function of the transconductor C(s) of the gm amplifier 16 is expressed by equation (38), but the first order is such that the pole closest to 0 -1/τ2 is left out and the others can be ignored. Rewriting equation (37) as having a dominant characteristic results in equation (39).

A Bode diagram based on this is shown in FIG. 19.
From this, the stability condition for this system is expressed as equation (40).

Normally, the design parameter is τ1 such that τ1>>τ2, and ABCS may be determined to an extent that can suppress unnecessary secondary sideband waves.
This time, for the sake of simplicity, LPF15 was used as a primary filter at the expense of control speed, but it is not limited to a primary filter as long as precise stability can be ensured.
As a result, the voltage Vplt can be controlled using the reference voltage Vref as a control target, which simultaneously controls the output voltage Vout.

Further, if digital processing is used, it is possible to process faster than the above-mentioned control. That is, it is also possible to replace the gm amplifier 16 as shown in FIG. 20, for example. Here, the gm amplifier 16 is replaced with a subtracter 40, an AD converter (ADC) 41, a digital signal processor (DSP) 42, and a current output DA converter (DAC) 43. The subtracter 40 and ADC 41 may be combined or may be a simple 1-bit latch comparator, but it is desirable that the operating speed be four times or more the chopper modulation frequency ωc. The DSP 42 is a band rejection filter (BRF) for removing secondary sidebands superimposed on the chopper demodulated signal, and the DAC 43 is a current output type DAC with relatively high resolution.

As can be understood from FIG. 18, for the voltage Vplt, the secondary sideband wave of the magnetic field response signal to be measured is the dominant interference wave, so if it is strongly attenuated by the DSP 42, the load on the LPF 15 will be lightened, and the cutoff It becomes possible to increase the frequency, that is, decrease τ1, and widen the closed loop band. Particularly if hunting is a concern due to the coarse resolution of the DAC 43, one means of stabilization is to set a dead condition to stop the operation.

(Second embodiment)
A magnetic sensor device 5 according to a second embodiment of the present invention will be described based on FIG. 21. The magnetic sensor device 5 of the second embodiment includes a magnetic sensor circuit 6, a chopper modulation/demodulation circuit 3, a feedback circuit 4, a current Ibias input terminal, a reference voltage Vref input terminal, and five clock input terminals. and a voltage Vout output terminal. The difference between the second embodiment and the first embodiment is that the magnetic sensor circuit 2 in FIG. 1 is replaced by a magnetic sensor circuit 6. Circuits that are the same as those in the first embodiment are given the same numbers, and descriptions thereof will be omitted.

The magnetic sensor circuit 6 that measures the magnetic field with the Hall element 21 will be described based on FIG. 22. The magnetic sensor circuit 6 includes a Hall element 21 that performs spinning current operation, a switch circuit 56, two resistors 57 and 58, and a current supply circuit 50 that supplies current to the Hall element 21. The Hall element 21 is operated by a spinning current method in which the connection is switched by the switch circuit 56, and removes the offset voltage of the output signal when no magnetic field is applied. The switch circuit 56 has two current input terminals, a GND connection terminal, four Hall element connection terminals, a clock input terminal, and two output terminals. The Hall element 21 has four terminals a to d, and is connected to a switch circuit 56. Here, the first resistor 57 is connected in series between the terminal c of the Hall element 21 and the switch circuit 56, and the second resistor 58 is connected between the terminal d of the Hall element 21 and the switch circuit 56. connected in series.

The operation of the spinning current supplied to the Hall element 21 will be explained based on FIG. 23. The spinning current operation is similar to the operation described in the first embodiment and FIG. 3, but differs from the first embodiment in the addition of two resistors and the input/output of the current i. Since the subsequent amplifier 11 that amplifies the output of the Hall element 21 is a high input impedance amplifier, the current i flows through the two resistors and the Hall element 21 to GND.

In the Hall element 21, when the spinning current state is 0, a current I is supplied to the terminal a, is discharged from the terminal c, and flows through the resistor 58 to GND. The Hall element 21 generates a potential between the terminal b and the terminal d that is proportional to the magnetic field perpendicular to the Hall element 21 and proportional to the current I, and a voltage Vb is applied from the terminal b and a voltage is applied from the terminal d through the resistor 57. Output as Vd. On the other hand, the product of the resistance value of the resistor 57 and the current i is added to the output of the Hall element 21 as an offset voltage. The voltage Vd is the sum of the Hall voltage generated by the Hall element 21 and the offset voltage. At the same time, a voltage drop occurs across the resistor 58 due to the current i. Since the output of the Hall element 21 is a differential signal, the voltage drop occurring across the resistor 58 does not affect the output of the Hall element 21.

When the Hall element 21 is in the spinning current state 1, a current I is supplied to the terminal b, is discharged from the terminal d, and flows through the resistor 57 to GND. The Hall element 21 generates a potential between terminals a and c that is proportional to the magnetic field perpendicular to the hall element 21 and proportional to the current I, and a voltage Va is applied from the terminal a and a voltage is applied from the terminal c through the resistor 58. Output as Vc. On the other hand, the product of the resistance value of the resistor 58 and the current i is added to the output of the Hall element 21 as an offset voltage. The voltage Vc is the sum of the Hall voltage generated by the Hall element 21 and the offset voltage. At the same time, a voltage drop occurs across the resistor 57 due to the current i. Since the output of the Hall element 21 is a differential signal, the voltage drop occurring across the resistor 57 does not affect the output of the Hall element 21.

The voltage drop that occurs in the resistor 57 and the resistor 58 is set to be approximately the same as the Hall voltage that occurs in the Hall element 21 in response to the magnetic field to be measured. In order to adjust the voltage drop occurring at the resistor 57 and the resistor 58, the resistance values of the resistor 57 and the resistor 58 may be adjusted by an external signal or trimming. If the current i is made proportional to the current I and the resistors 57 and 58 are manufactured using the same process recipe as the Hall element 21, the voltage generated at the resistors 57 and 58 will be correlated with the Hall voltage generated in the Hall element 21 due to the magnetic field to be measured. do. The voltage generated at the resistor 57 and the resistor 58 can be used as a sensitivity monitor of the Hall element 21.

Switching between the spinning current state 0 and the spinning current state 1 is performed by supplying the clock CPsp to the switch circuit 56. The switch circuit 56 outputs Vd/Va from a first output terminal and Vb/Vc from a second output terminal.

The current supply circuit 50 will be explained based on FIG. 22. The current supply circuit 50 includes two current mirror circuits 51 and 52, a switch circuit 53, two inverters 54 and 55, a current input terminal, and two current output terminals. Current mirror circuit 51 has three MOS transistors 60, 61, and 62. Current mirror circuit 52 has three MOS transistors 63, 64, and 65. The mirror ratio between MOS transistor 60 and MOS transistor 61 is 1:1. The mirror ratio between MOS transistor 63 and MOS transistor 64 is 1:1. A current I input from the current input terminal flows into the MOS transistor 60 and is current mirrored as the current I into the MOS transistor 61. The current I that is current-mirrored to the MOS transistor 61 flows to the MOS transistor 63, and is current-mirrored to the MOS transistor 64 as the current I. The current I current-mirrored by the MOS transistor 64 is supplied to the Hall element 21 via the switch circuit 56.

MOS transistor 65 current mirrors current I with MOS transistor 63 at a predetermined mirror ratio to generate current i. The mirror ratio is determined so that the voltage drop between the current i and the resistor 57 or the resistor 58 is approximately the same as the Hall voltage generated by the Hall element 21, as described above. The MOS transistor 65 may be configured by connecting a plurality of MOS transistors in parallel, and the mirror ratio may be adjusted by separating some of the MOS transistors using an external signal or trimming. Similar to MOS transistor 65, MOS transistor 62 current mirrors current I with MOS transistor 60 at a predetermined mirror ratio to generate current -i. The MOS transistor 62 may also be configured by connecting in parallel a plurality of MOS transistors that can be separated by external signals or trimming.

One of the current i and the current -i is selected by the clock CPplt, and is supplied to the Hall element 21 via the switch circuit 56 and further via the resistor 57 or 58. The current supply circuit 50 switches the current i and the current -i supplied to the Hall element 21 with a duty ratio of 50% using the clock CPplt when the spinning current state is 0, and similarly switches between the current i and the current -i supplied to the Hall element 21 using the clock CPplt when the spinning current state is 1. Current i and current -i supplied to element 21 are switched at a duty ratio of 50%. The signals output by the magnetic sensor circuit 6 corresponding to the currents i and -i are called pilot signals, as in the first embodiment.

The magnetic sensor circuit 6 outputs a signal in which a signal corresponding to the magnetic field to be measured passing through the Hall element 21 and a pilot signal are superimposed. Signals Vd/Va and Vb/Vc output by the magnetic sensor circuit 6 are output to the chopper modulation/demodulation circuit 3. The circuit configuration and operation after the chopper modulation/demodulation circuit 3 are the same as in the first embodiment.

According to the magnetic sensor device of this embodiment, by utilizing the frequency orthogonality of the signal corresponding to the magnetic field to be measured (I side) and the pilot signal (Q side), the Q side signal is always transmitted regardless of time division operation. Accordingly, the magnetic sensor device can be operated while adjusting the magnetic field measurement sensitivity of the I-side signal.

1, 5 Magnetic sensor device 2, 6 Magnetic sensor circuit 3 Chopper modem device 4 Feedback circuit 4
10, 12, 14 Mixer 11 Amplifier 16 GM amplifier 21 Hall element 23 Reference magnetic field generation circuit 41 AD converter 42 Digital signal processor 43 DA converter 50 Current supply circuit

Claims (5)

Comprising a magnetic sensor circuit, a chopper modulation/demodulation circuit, a feedback circuit, a current input terminal, a voltage input terminal, and an output terminal,
The magnetic sensor circuit includes one Hall element or a set of Hall elements connected in parallel, and a magnetic field generation circuit,
The chopper modulation/demodulation circuit has a first mixer whose output is connected to an amplifier, an amplifier whose output is connected to a second mixer, and a second mixer,
The feedback circuit has a third mixer and a voltage-current conversion circuit,
the current input terminal and the feedback circuit are connected to the magnetic sensor circuit;
the magnetic sensor circuit is connected to the chopper modulation/demodulation circuit;
the chopper modulation/demodulation circuit is connected to the output terminal and the feedback circuit;
A magnetic sensor device characterized in that the voltage input terminal is connected to the feedback circuit. Comprising a magnetic sensor circuit, a chopper modulation/demodulation circuit, a feedback circuit, a current input terminal, a voltage input terminal, and an output terminal,
The magnetic sensor circuit includes one Hall element or a set of Hall elements connected in parallel, and a current supply circuit,
The chopper modulation/demodulation circuit has a first mixer whose output is connected to an amplifier, an amplifier whose output is connected to a second mixer, and a second mixer,
The feedback circuit has a third mixer and a voltage-current conversion circuit,
the current input terminal and the feedback circuit are connected to the magnetic sensor circuit;
the magnetic sensor circuit is connected to the chopper modulation/demodulation circuit;
the chopper modulation/demodulation circuit is connected to the output terminal and the feedback circuit;
A magnetic sensor device characterized in that the voltage input terminal is connected to the feedback circuit. 3. The magnetic sensor device according to claim 1, wherein the first to third mixers are passive multipliers configured with switches. 3. The magnetic sensor device according to claim 1, wherein the voltage-current conversion circuit is a transconductance amplifier. 3. The magnetic sensor device according to claim 1, wherein the voltage-current conversion circuit includes an AD converter, a digital signal processor, and a current output DA converter.