JP7148372B2 - motor drive - Google Patents

Description

The present invention relates to a motor drive device.

Electric transportation equipment such as EV (Electric Vehicle) mainly uses a three-phase synchronous motor such as a brushless motor as a power source. A three-phase synchronous motor (hereinafter abbreviated as "motor") applies a so-called rotating magnetic field to the windings, in which the magnetic poles of the windings are switched in the direction of rotation by applying a three-phase AC-like voltage to the windings. A rotor of the motor, which is made up of permanent magnets, rotates following the generated rotating magnetic field, thereby functioning as a power source.

Since the power source for EVs and the like is a secondary battery (battery) that is a DC power source, a FET (Field Effect Transistor) or IGBT is required to generate a three-phase AC-like voltage to be applied to the windings of the motor from the DC voltage of the battery. (Insulated Gate Bipolar Transistor) or other switching elements are used as inverters. By PWM (Pulse Width Modulation) that turns on and off (switches) the switching elements of the inverter, the DC voltage supplied from the battery is converted into a three-phase AC-like voltage.

However, by switching the switching element, a current ripple may occur in which the current of the battery fluctuates in small steps. A motor drive circuit including an inverter is often provided with a smoothing capacitor having one end connected to the positive terminal of the inverter (the terminal connected to the positive terminal of the battery) and the other end connected to the negative terminal. . In general, a capacitor does not conduct direct current, but conducts alternating current. Therefore, the smoothing capacitor absorbs and eliminates current fluctuations such as current ripple generated on the positive electrode side of the battery by charging and discharging.

A smoothing capacitor that short-circuits a positive terminal and a negative terminal is substantially equivalent to a series resonance circuit in which a capacitor and a coil (inductor) are connected in series. A series resonant circuit has a minimum impedance at its own resonant frequency. Therefore, if the frequency of the current ripple matches the resonance frequency of the series resonance circuit, the current ripple will not be eliminated and resonance will be amplified instead.

In order to suppress the resonance of the smoothing capacitor, it is effective to increase the capacity of the smoothing capacitor.

Patent Literature 1 discloses an invention of a rotary electric machine drive device that forms a current ripple outside the resonance frequency range of a smoothing capacitor by increasing the switching frequency of an inverter.

In Patent Document 2, a plurality of inductors are connected in series to the positive electrode side of a battery, and a switch connected in parallel with one inductor is turned on or off to control energization of the one inductor to adjust the resonance frequency. An invention is disclosed for a rotating electric machine drive that changes.

Japanese Patent No. 6004086 Japanese Patent No. 6004087

However, the rotary electric machine drive device disclosed in Patent Document 1 frequently operates the switching elements in order to increase the switching frequency of the inverter, so the power of the battery is consumed by the switching operation, resulting in the power consumption of the electric transportation equipment. There was a problem that the volume increased.

In addition, the rotary electric machine drive device disclosed in Patent Document 2 requires mounting an inductor that can withstand current ripple in addition to battery voltage reaching several hundred volts. Since the inductor is generally expensive and huge, there is a problem that not only does the manufacturing cost of the rotary electric machine drive device increase, but also the downsizing of the rotary electric machine drive device is hindered. Furthermore, there is a problem that power consumption of the electric transportation equipment increases due to power consumption by the inductor.

SUMMARY OF THE INVENTION It is an object of the present invention to provide a motor drive device capable of suppressing current ripple caused by inverter switching with a simple and inexpensive configuration.

The motor drive device according to claim 1 comprises a drive circuit for generating a voltage applied to the windings of the motor by pulse width modulation of power supplied from a DC power supply, and a positive electrode and a negative electrode on the DC power supply side of the drive circuit. and a second resonant circuit having an inductor and a second capacitor connected in series to short-circuit the positive terminal and the negative terminal of the DC power supply. and a control unit for controlling the drive circuit to perform the pulse width modulation at a frequency different from the resonance frequency at which the impedance of the circuit including the first resonance circuit and the second resonance circuit is minimized. .

According to the motor drive device of claim 1, by providing the second resonance circuit in addition to the first resonance circuit, the resonance frequency is set to a frequency different from that in the case of providing only the first resonance circuit, and the pulse width modulation is performed. The drive circuit can be controlled so that the frequency does not match the set resonance frequency.

A motor drive device according to claim 2 is the motor drive device according to claim 1, further comprising a switch for turning on and off connection between a positive terminal and a negative terminal of the DC power supply of the second resonance circuit, When the switch is on, the resonant frequency includes a first resonant frequency and a second resonant frequency higher than the first resonant frequency, and when the switch is off, the resonant frequency is higher than the first resonant frequency. and a third resonance frequency lower than the second resonance frequency, and the control unit generates the pulse when the rotation speed of the motor is within a predetermined range and the torque of the output shaft of the motor is equal to or greater than a predetermined torque. The switch is turned off when the width modulation frequency is within a predetermined frequency range including the second resonance frequency.

According to the motor drive device of claim 2, when the pulse width modulation frequency matches the set resonance frequency, the switch is turned off, and the drive circuit is turned off so that the pulse width modulation frequency does not match the resonance frequency. can be controlled.

The motor drive device according to claim 3 is the motor drive device according to claim 2, wherein each of the capacitance C _Inv of the first capacitor, the capacitance C _Bat of the second capacitor, and the inductance L _Bat of the inductor is: It is calculated by the following formula A based on the second resonance frequency f2.

According to the motor drive device of claim 3, each of the capacitance C _Inv of the first capacitor, the capacitance C _Bat of the second capacitor, and the inductance L _Bat of the inductor is adjusted based on the second resonance frequency different from the first resonance frequency. By setting, a frequency between the first resonance frequency and the second resonance frequency can be used as a frequency for pulse width modulation.

The motor drive device according to claim 4 is the motor drive device according to claim 3, wherein the second resonance frequency f2 is the base speed, which is the maximum rotation speed at which the motor can maintain maximum torque. It is greater than the larger one of f2-1 and f2-2 calculated using the following equations B and C based on the number of pole pairs and the carrier frequency of the carrier wave related to the pulse width modulation.
f2-1 = basal velocity / 60 x pole log number x 6 ... B
f2-2 = carrier frequency + base velocity / 60 x number of pole pairs x 3...C

According to the motor drive device of claim 4, by setting the second resonance frequency f2 based on the base speed, which is the maximum number of rotations at which the motor can maintain the maximum torque, the motor is rotated at a speed equal to or lower than the base speed. Each of the capacitance C _Inv of the first capacitor, the capacitance C _Bat of the second capacitor and the inductance L _Bat of the inductor can be set accordingly.

The motor drive device according to claim 5 is the motor drive device according to claim 3, wherein the second resonance frequency f2 is the maximum machine rotation speed, which is the maximum rotation speed of the motor, and the poles of the motor. It is larger than the larger one of f2-1 and f2-2 calculated using the following equations D and E based on the logarithm and the carrier frequency of the carrier wave related to the pulse width modulation.
f2-1=maximum machine speed/60×number of pole pairs×6 …D
f2-2 = carrier frequency + maximum machine speed / 60 x pole logarithm x 3...E

According to the motor drive device of claim 5, by setting the second resonance frequency f2 based on the maximum machine rotation speed, which is the maximum rotation speed of the motor, the motor can be rotated at the maximum machine rotation speed or less. Each of the capacitance C _Inv of the first capacitor, the capacitance C _Bat of the second capacitor, and the inductance L _Bat of the inductor can be set corresponding to the case where the voltages are applied.

According to the motor drive device of claim 1, by controlling the drive circuit so that the pulse width modulation frequency does not match the set resonance frequency, it is possible to suppress the current ripple caused by the switching of the inverter.

According to the motor drive device of claim 2, by turning off the switch and controlling the drive circuit so that the frequency of the pulse width modulation does not match the resonance frequency, it is possible to suppress the current ripple caused by the switching of the inverter. It has the effect of

According to the motor drive device of claim 3, the frequency between the first resonance frequency and the second resonance frequency is the frequency of the pulse width modulation, thereby suppressing the current ripple caused by the switching of the inverter. Effective.

According to the motor drive device of claim 4, each of the capacitance C _Inv of the first capacitor, the capacitance C _Bat of the second capacitor, and the inductance L _Bat of the inductor is adjusted to correspond to the case where the motor is rotated below the base speed. By setting, it is possible to suppress the current ripple caused by switching of the inverter until the rotation speed of the motor reaches the base speed.

According to the motor drive device of claim 5, the capacitance C _Inv of the first capacitor, the capacitance C _Bat of the second capacitor, and the inductance L _Bat of the inductor correspond to the case where the motor is rotated at a maximum machine speed or less. By setting each, it is possible to suppress the current ripple caused by the switching of the inverter.

It is a block diagram showing an example of composition of a motor drive concerning a 1st embodiment of the present invention. FIG. 5 shows changes in the DC side current gain with respect to the switching frequency of the transistor of the inverter in the motor drive device according to the first embodiment of the present invention; FIG. 1A is a block diagram showing an example of the configuration of a general motor drive device, and FIG. 1B is a block diagram showing an example of the configuration of a motor drive device according to a first embodiment of the present invention; FIG. FIG. 4 is an explanatory diagram showing changes in the DC side current gain of a general motor drive device and changes in the DC side current gain of the motor drive device according to the first embodiment of the present invention with respect to the transistor switching frequency; be. 5 is an explanatory diagram showing an example of a motor torque, a motor phase current, a U-phase interphase voltage, and a battery current at the frequency f3 shown in FIG. 4; FIG. It is the schematic which showed an example of an electric frequency 6th component and a carrier 1st-order component. (A) is a schematic diagram showing an example of a sinusoidal waveform having a phase and amplitude according to the position of the magnetic poles of a rotor, and (B) is a schematic diagram showing an example of a carrier wave waveform in PWM. , (C) are schematic diagrams showing an example of a pulsed PWM control signal generated from the sine wave and the carrier wave; 4 is a schematic diagram showing an example of changes in motor torque with respect to motor rotation speed; FIG. When the specifications of each of the smoothing capacitor, the battery unit inductor, and the battery unit capacitor are determined based on the second resonance frequency calculated using the base speed, and based on the second resonance frequency calculated using the maximum machine rotation speed FIG. 4 is an explanatory diagram showing changes in DC side current gain when specifications of a smoothing capacitor, a battery section inductor, and a battery section capacitor are determined; It is a block diagram which shows an example of a structure of the motor drive device which concerns on the 2nd Embodiment of this invention. FIG. 9 is an explanatory diagram showing changes in DC side current gain with respect to the switching frequency of the transistor of the inverter in the motor drive device according to the second embodiment of the present invention; (A) is an explanatory diagram showing changes in the DC side current gain with respect to the switching frequency of the transistors of the motor drive device according to the second embodiment of the present invention, and (B) is an explanatory diagram showing changes in the motor torque with respect to the motor rotation speed. It is the schematic which showed an example of a change. 8 is a flow chart showing an example of control of the motor drive device according to the second embodiment of the present invention;

[First embodiment]
A motor drive device 100 according to the present embodiment will be described below with reference to FIGS. 1 to 9. FIG. FIG. 1 is a block diagram showing an example of the configuration of a motor drive device 100 according to this embodiment. A motor drive device 100 shown in FIG. 1 is a device that generates electric power for driving a motor 60 of a three-phase synchronous motor that powers electric transportation equipment.

The motor driving device 100 includes a DC power supply unit 10 including a battery 12 which is a DC power supply, and a DC voltage supplied from the DC power supply unit 10, which is applied to a coil 62 which is a winding of a motor 60 by PWM by an inverter 40. An inverter section 30 that generates a phase alternating voltage, and a motor control device 80 that outputs PWM signals for controlling switching of transistors 42U, 42V, 42W, 44U, 44V, and 44W, which are switching elements that constitute the inverter 40. and a motor drive circuit 90 that amplifies the PWM signal output by the motor control device 80 to such an extent that the transistors 42U, 42V, 42W, 44U, 44V, and 44W can be turned on, and outputs the amplified signal to the inverter 40 .

Motor 60 is, for example, a three-phase synchronous motor driven by a three-phase AC-like voltage, but it may also be a motor with brushes driven by DC. In such a case, instead of inverter 40, four switching elements and a drive circuit having an H-bridge circuit composed of:

The battery 12 of the DC power supply unit 10 is a rechargeable secondary battery such as a lithium-ion battery or a nickel-metal hydride battery. A voltage of about 100 V can be output.

The battery 12 is equivalently represented by a circuit model in which a battery internal resistance 14 and a battery internal inductor 16 are connected in series. As shown in FIG. 1 , the battery internal resistance 14 is represented by a model in which one end is connected to the positive electrode of the battery 12 and the other end is connected to one end of the battery internal inductor 16 . The battery internal inductor 16 is represented by a model in which one end is connected to the other end of the battery internal resistor 14 and the other end is connected to the inverter unit 30 as described above. A terminal portion of the battery 12 is connected to a battery portion inductor 18 and a battery portion capacitor 20 that short-circuit the positive electrode and the negative electrode. The battery section inductor 18 and the battery section capacitor 20 may be arranged closer to the body of the battery 12 than the terminal section of the battery 12 .

The inverter section 30 is composed of a three-phase (U-phase, V-phase, W-phase) inverter 40 . As shown in FIG. 1, the inverter 40 includes transistors 42U, 42V, 42W, 44U, 44V and 44W, which are switching elements. The transistors 42U, 42V, 42W, 44U, 44V, and 44W are, for example, IGBTs that can handle high voltages and have relatively low on-resistance, but may be N-channel FETs.

In inverter 40, each of transistors 42U, 42V and 42W functions as an upper stage switching element, and each of transistors 44U, 44V and 44W functions as a lower stage switching element. Transistors 42U, 42V, 42W and transistors 44U, 44V, 44W are collectively referred to as "transistor 42" and "transistor 44" when it is not necessary to distinguish them individually. , "U", "V", and "W".

When the transistors 42U, 42V, 42W, 44U, 44V, 44W are IGBTs, the emitter of the transistor 42U and the collector of the transistor 44U are connected to the terminal of the coil 62U, and the emitter of the transistor 42V and the collector of the transistor 44V are connected to the coil. 62V terminal, and the emitter of transistor 42W and the collector of transistor 44W are connected to the terminal of coil 62W.

When the transistors 42U, 42V, 42W, 44U, 44V, 44W are FETs, the source of the transistor 42U and the drain of the transistor 44U are connected to the terminal of the coil 62U, and the source of the transistor 42V and the drain of the transistor 44V are connected to The source of transistor 42W and the drain of transistor 44W are connected to the terminal of coil 62W.

A smoothing capacitor 32 short-circuits the positive terminal and the negative terminal of the inverter 40 .

Transistors 42U, 42V, 42W, 44U, 44V, 44W (whether IGBTs or FETs) are turned on when a positive charge drive signal is applied to their gates. For example, when transistor 42U is on and transistor 44U is off, coil 62U is energized by battery 12, and when transistor 42U is off and transistor 44U is on, coil 62U is grounded. Similarly, when the transistor 42V is on and the transistor 44V is off, the power of the battery 12 is applied to the coil 62V, and when the transistor 42V is off and the transistor 44V is on, the coil 62V is grounded and the transistor 42W is turned on. When the transistor 44W is on and the transistor 44W is off, the power of the battery 12 is applied to the coil 62W, and when the transistor 42W is off and the transistor 44W is on, the coil 62W is grounded.

The inverter 40 energizes the coils 62U, 62V and 62W by turning on and off the transistors 42U, 42V, 42W, 44U, 44V and 44W as described above. Further, the inverter 40 can change the voltage applied to the coils 62U, 62V, 62W by PWM that turns on and off the transistors 42U, 42V, 42W, 44U, 44V, 44W in small increments.

The motor 60 is a three-phase synchronous motor in which a rotor 64 made of permanent magnets rotates following magnetic fields generated in energized coils 62U, 62V, and 62W. Each terminal (one end) of the coils 62U, 62V, 62W is connected to the transistors 42U, 42V, 42W, 44U, 44V, 44W of the inverter 40 as described above, and the other end of each of the coils 62U, 62V, 62W is connected to , 1 to the neutral point 62N. The coils 62U, 62V, 62W may be, for example, delta-connected instead of star-connected.

The coils 62U, 62V, and 62W are collectively referred to as "coils 62" when there is no need to distinguish them individually, and "U", "V", and "W" when they need to be distinguished individually. is referred to with the sign of

Motor drive device 100 according to the present embodiment switches energization of coils 62U, 62V, and 62W in time series by inverter 40 to generate a so-called rotating magnetic field in coils 62U, 62V, and 62W. The rotor 64 rotates due to attraction to or repulsion from the rotating magnetic field generated in the coils 62U, 62V, 62W.

In the three-phase synchronous motor, it is necessary to energize the coils 62U, 62V, 62W according to the position of the magnetic poles of the rotor 64. It detects the position of the magnetic pole of

Based on the magnetic pole position of the rotor 64 detected by the motor magnetic pole position detector 70 and the value of the current output by the inverter 40 detected by the inverter current sensor 50, the motor control device 80 controls the coils 62U, 62V and 62W. The waveform phase and amplitude (range of voltage change) of the voltage to be applied are calculated, and a PWM control signal, which is a pulse signal corresponding to the phase and amplitude, is output to the motor drive circuit 90 . The phase of the voltage waveform is determined mainly in response to changes in the position of the magnetic poles of rotor 64 , and the upper limit of the amplitude is determined based on the value of the current output by inverter 40 . For example, when the current value exceeds a predetermined value, the motor 60 may be overloaded, so the amplitude of the voltage waveform is suppressed.

The motor drive circuit 90 amplifies the PWM control signal output from the motor control device 80 to such an extent that the transistors 42 and 44 can be turned on, and applies the generated drive signal to the gates of the transistors 42 and 44 . Since the drive signal is composed of pulses in accordance with the phase and amplitude according to the position of the magnetic poles of the rotor 64, like the PWM control signal, the inverter 40 causes the coil 62 to correspond to the position of the magnetic poles of the rotor 64. A voltage can be generated that causes a corresponding rotating magnetic field.

Further, motor drive device 100 according to the present embodiment is provided with current sensor 24 between the other end of battery internal inductor 16 and one end of battery section inductor 18 . The current sensor 24 is, for example, a sensor that detects the current value in the conductor based on the induced current generated by the current in the conductor.

In addition to the current value detected by the current sensor 24, Ibat, and the current value detected by the inverter current sensor 50, etc., the positive electrode side terminal of the inverter 40 calculated based on the signals of the transistors 42U, 42V, 42W, 44U, 44V, and 44W. Assuming that the current value is Idc, the DC side current gain Ig in the motor driving device 100 is calculated by the following equation (1).
Ig=Ibat/Idc (1)

FIG. 2 shows changes in the DC side current gain Ig with respect to the switching frequency of the transistors 42 and 44 of the inverter 40, that is, the PWM frequency, in the motor drive device 100 according to the present embodiment. A circuit (first resonance circuit) formed by the smoothing capacitor 32 and a circuit (second resonance circuit) formed by the battery section inductor 18 and the battery section capacitor 20 each act as a resonance circuit. As a result, the curve 102 showing the change in the DC side current gain Ig in FIG. , the impedance of the circuit including the first resonant circuit and the second resonant circuit is minimized. As a result, at the first resonance frequency f1 and the second resonance frequency f2, resonance occurs in which the current ripple caused by the switching of the transistors 42 and 44 is amplified.

However, resonance does not occur at frequencies different from the first resonance frequency f1 or the second resonance frequency f2, such as the inverter switching frequency range between the first resonance frequency f1 and the second resonance frequency f2. Therefore, it is possible to switch the transistors 42 and 44 at frequencies in this range.

FIG. 3A is a block diagram showing an example of the configuration of a general motor drive device, and FIG. 3B is a block diagram showing an example of the configuration of the motor drive device 100 according to this embodiment. Motor drive device 100 according to the present embodiment shown in FIG. It is different from a typical motor drive device.

FIG. 4 shows changes in the DC side current gain Ig of a general motor drive device and changes in the DC side current gain Ig of the motor drive device 100 according to the present embodiment with respect to the switching frequency of the transistors 42 and 44. showing.

A curve 104 in FIG. 4 shows changes in the DC side current gain Ig of a general motor drive device. The DC side current gain Ig shows a maximum value.

A curve 102 in FIG. 4 shows a change in the DC side current gain Ig of the motor drive device 100 according to the present embodiment. Although Ig exhibits a maximum value, the maximum value at the first resonance frequency f1 (<the third resonance frequency f1') is more relaxed than the maximum value at the third resonance frequency f1' of a general motor drive device. there is As a result, at the frequency f3, the curve 104 is above the reference value 105, while the curve 102 is below the reference value 105.

FIG. 5 is an explanatory diagram showing an example of the motor torque, motor phase current, U-phase interphase voltage, and battery current at the frequency f3 shown in FIG. In FIG. 5, the motor torque fluctuates within a certain range, but the effective voltage value of the voltage applied to the coil 62 of the motor 60 generated by PWM is the U-phase (V-phase and W-phase are the same) in FIG. , a certain amount of variation may occur in both the motor driving device 100 according to the present embodiment and a general motor driving device.

Further, in the motor phase currents shown in FIG. 5, all of the U-phase current iu, the V-phase current iv, and the W-phase current iw change in a substantially sinusoidal shape under the influence of the interphase voltage change. Such changes are the same in the motor drive device 100 according to the present embodiment and in a general motor drive device.

However, there is a difference of about ΔA in the maximum amplitude of the battery current between the curve 112 indicating the battery current of the motor driving device 100 according to the present embodiment and the curve 110 indicating the battery current of a general motor driving device. there is ΔA is a current value equivalent to approximately 50% less than the battery current of a general motor drive device, although it depends on the specifications of the motor drive device 100 . Therefore, according to the motor drive device 100 according to the present embodiment, it is possible to suppress the current ripple caused by the small fluctuations of the battery current, as compared with a general motor drive device.

However, in the motor drive device 100 according to the present embodiment, in order to secure the inverter switching frequency range as shown in FIG. It is important to properly determine the specification of 20. In particular, when the first resonance frequency f1 matches the frequency of the current fluctuation component on the power supply side (battery 12 side), the current ripple increases. After setting, the specifications of the smoothing capacitor 32, the battery section inductor 18, and the battery section capacitor 20 are determined so that the frequency of the set current fluctuation component does not match the second resonance frequency f2 shown in FIGS.

Calculation of the frequency of the current fluctuation component on the power supply side will be described below. The current fluctuation component includes an electric frequency sixth-order component f2-1 (Hz) and carrier first-order components f2-2A and f2-2B (Hz). 2), the first-order carrier components f2-2A and f2-2B are calculated by the following equation (3).
Electric frequency 6th order component f2-1
= Motor rotation speed (rpm) ÷ 60 × number of pole pairs × 6 (2)
Carrier primary components f2-2A, f2-2B
= carrier frequency ± motor rotation speed (rpm) ÷ 60 × number of pole pairs × 3 (3)

FIG. 6 is a schematic diagram showing an example of the electric frequency sixth-order component f2-1 and carrier first-order components f2-2A and f2-2B. The electric frequency sixth-order component f2-1, the carrier first-order component f2-2A, and the carrier first-order component f2-2B do not match each other.

In the motor drive device 100 according to the present embodiment, it is necessary to set the frequency of the current fluctuation component so as not to coincide with the first resonance frequency f1. and carrier primary components f2-2A and f2-2B, the one with the largest value is adopted.

The carrier frequency in equation (3) is the frequency of the carrier used in generating the PWM control signal having pulses according to the phase and amplitude according to the magnetic pole position of the rotor 64 . FIG. 7A shows an example of a sine wave waveform having a phase and amplitude corresponding to the position of the magnetic poles of the rotor 64, FIG. 7B shows an example of a carrier wave waveform in PWM, and FIG. ) is an example of a pulsed PWM control signal generated from the sine wave and the carrier wave.

Motor control device 80 of motor driving device 100 according to the present embodiment generates a sine wave signal as shown in FIG. Then, the motor control device 80 uses a circuit such as a comparator to compare the amplitude of the sine wave with the amplitude of the carrier wave shown in FIG. If it is less than the amplitude, a pulse indicating ON is generated, and if the amplitude of the sine wave is equal to or greater than the amplitude of the carrier wave, no pulse is generated (indicating OFF), and the PWM shown in FIG. Generate control signals.

As shown in FIG. 7, the carrier frequency directly affects the frequency of the PWM control signal, and the PWM control signal directly affects the switching frequency of the transistors 42 and 44 that make up the inverter 40. Therefore, for convenience, The carrier frequency may be considered approximately the same as the switching frequency of transistors 42,44.

Since the equations (2) and (3) include a motor rotation speed term, the electric frequency sixth-order component f2-1 and the carrier first-order components f2-2A and f2-2B increase according to the motor rotation speed. Therefore, how to define the number of rotations of the motor is an important factor in calculating the electric frequency sixth-order component f2-1 and the carrier first-order components f2-2A and f2-2B. If the motor rotation speed to be substituted into equations (2) and (3) is large, the electric frequency sixth-order component f2-1 and the carrier first-order components f2-2A and f2-2B can become higher than the first resonance frequency f1. This is because

FIG. 8 is a schematic diagram showing an example of changes in motor torque with respect to motor rotation speed. As shown in FIG. 8, the motor 60 can maintain maximum torque until the motor speed reaches the base speed of 120, but when the motor speed exceeds the base speed of 120, the motor torque gradually falls below the maximum torque. Thus, when the motor rotation speed exceeds the maximum machine rotation speed of 122, the motor 60 cannot continue rotating.

Therefore, the base speed 120 is the maximum motor speed at which maximum torque can be maintained, and the maximum machine speed 122 is the maximum motor speed at which the motor 60 can be rotated in practice. The value of the second resonance frequency f2 differs depending on whether the base speed 120 or the maximum machine rotation speed 122 is substituted into the equations (2) and (3).

From the above, in order to calculate the second resonance frequency f2 using the base velocity 120, f2-1 and f2-2 calculated using the following equations (4) and (5), whichever is larger, is adopted as the second resonance frequency f2.
f2-1 = basal velocity / 60 x pole log number x 6 (4)
f2-2 = carrier frequency + base velocity / 60 x pole logarithm x 3 (5)

In order to calculate the second resonance frequency f2 using the maximum machine rotation speed 122, f2-1 and f2-2 calculated using the following formulas (6) and (7) are calculated using A high frequency is adopted as the second resonance frequency f2.
f2-1 = maximum machine speed/60 x number of pole pairs x 6 (6)
f2-2 = carrier frequency + maximum machine rotation speed / 60 x pole logarithm x 3 (7)

Then, the specifications of the smoothing capacitor 32, the battery section inductor 18, and the battery section capacitor 20 are determined by applying the employed second resonance frequency f2 to the following equation (8). In the following equation (8), L _Bat is the inductance (H) of the battery section inductor 18, C _Inv is the capacity (F) of the smoothing capacitor 32, and C _Bat is the capacity (F) of the battery section capacitor 20. be.

As a result, the DC side current gain Ig with respect to the switching frequency of the transistors 42 and 44 changes as shown in FIG. A curve 102A in FIG. 9 represents the DC side current gain Ig when the specifications of the smoothing capacitor 32, the battery section inductor 18 and the battery section capacitor 20 are determined based on the second resonance frequency f2 calculated using the base speed 120. , and curve 102B is the direct current when the specifications of each of smoothing capacitor 32, battery section inductor 18, and battery section capacitor 20 are determined based on second resonance frequency f2 calculated using maximum machine rotation speed 122. It shows changes in the side current gain Ig.

The curve 102A occurs at a frequency where the second resonance frequency f2A is relatively close to the first resonance frequency f1A. Ripple can be effectively suppressed. Since the frequency between the first resonance frequency f1A and the second resonance frequency f2A is a relatively low frequency, the switching operations of the transistors 42 and 44 can be reduced, and the power consumption due to the switching operations can be suppressed.

Paradoxically, the fact that the current ripple can be effectively suppressed means that the current ripple can be suppressed to the extent that there is no practical problem even if the capacities of the smoothing capacitor 32 and the battery section capacitor 20 are reduced. As a result, the motor drive device 100 can be made compact and low cost.

In curve 102B, the DC side current gain Ig does not decrease so much between the first resonance frequency f1B and the second resonance frequency f2B. f2B occurs at a frequency significantly different from the first resonance frequency f1B.

As shown in equations (4) to (7), the second resonance frequency f2 depends on the motor speed. Therefore, in the case of the curve 102A in which the second resonance frequency f2A occurs at a frequency relatively close to the first resonance frequency f1A, the current ripple is effective at the motor rotation speed up to the base speed 120 used to calculate the second resonance frequency f2A. can be effectively suppressed.

In the case of the curve 102B in which the second resonance frequency f2B occurs at a frequency significantly different from the first resonance frequency f1B, the current ripple is suppressed at the motor rotation speed up to the maximum machine rotation speed 122 used to calculate the second resonance frequency f2B. can.

As described above, according to the motor drive device 100 of the present embodiment, by connecting in series the battery section inductor 18 that short-circuits the positive and negative terminals of the battery 12 and the battery section capacitor 20, the second A current ripple can be effectively suppressed between the second resonance frequency f2 different from the first resonance frequency f1.

Compared to a general motor drive device, only the battery section inductor 18 and the battery section capacitor 20 are added to the motor drive device 100 according to the present embodiment with a simple and inexpensive configuration.

Since the second resonance frequency f2 changes depending on the motor rotation speed, the second resonance frequency f2 is calculated according to the rotation speed at which the motor 60 is used, and based on the calculated second resonance frequency f2, the smoothing capacitor 32, the specifications of each of the battery section inductor 18 and the battery section capacitor 20 can be determined appropriately.

[Second embodiment]
Next, a motor drive device 200 according to a second embodiment will be described with reference to FIGS. 10 to 13. FIG. FIG. 10 is a block diagram showing an example of the configuration of motor drive device 200 according to the present embodiment. The motor drive device 200 shown in FIG. 10 is related to the first embodiment in that a switch 26 for turning on/off a resonance circuit composed of a battery inductor 18 and a battery capacitor 20 is mounted in the DC power supply unit 10. It differs from the motor driving device 100 . However, since other configurations are the same as those of the motor drive device 100 according to the first embodiment, the other configurations are denoted by the same reference numerals as those of the motor drive device 100 according to the first embodiment. , detailed description is omitted.

FIG. 11 shows changes in the DC side current gain Ig with respect to the switching frequency of the transistors 42 and 44 of the inverter 40 in the motor drive device 200 according to this embodiment. A curve 102 in FIG. 11 indicates a change in the DC side current gain Ig of the motor drive device 200 according to the present embodiment when the switch 26 is turned on. A curve 102 is similar to the curve 102 representing changes in the DC side current gain Ig of the motor driving device 100 according to the first embodiment shown in FIGS.

A curve 104 in FIG. 11 indicates a change in the DC side current gain Ig of the motor drive device 200 according to the present embodiment when the switch 26 is turned off. A curve 104 is similar to the curve 104 showing changes in the DC side current gain Ig of the general motor drive device shown in FIG.

A curve 102 shows a maximum value that greatly exceeds the reference value 105 at the first resonance frequency f1 and the second resonance frequency f2, and resonance occurs in which the current ripple due to the switching of the transistors 42 and 44 is amplified. Since no resonance occurs between the resonance frequency f1 and the second resonance frequency f2, it is possible to switch the transistors 42 and 44 within that range of frequencies. However, it is not suitable for operating the transistors 42, 44 at frequencies higher than the second resonant frequency f2.

When the transistors 42 and 44 are operated at a frequency higher than the second resonance frequency f2, the switch 26 is turned off in the motor driving device 200 according to the present embodiment. When the switch 26 is turned off, as shown by the curve 104 in FIG. 11, the DC side current gain Ig becomes equal to or less than the reference value 105 in the high frequency range. becomes less likely to occur.

Therefore, by controlling the on/off of the switch 26, the wide range of frequencies above the first resonance frequency f1 shown in FIG.

FIG. 12(A) shows changes in the DC side current gain Ig with respect to the switching frequency of the transistors 42 and 44 in the motor drive device 200 according to this embodiment, and FIG. 1 is a schematic diagram showing an example of a change in motor torque with respect to .

As described in the first embodiment, the second resonance frequency f2 depends on the motor speed. Therefore, resonance can occur over a range of motor speeds.

As shown in FIG. 12A, when the switch 26 is turned on, the DC side current gain Ig exceeds the reference value 105 between the frequency f4 and the frequency f5. In such a case, the DC side current gain Ig exceeding the reference value 105 also affects changes in the torque of the motor 60 with respect to the motor rotation speed.

Specifically, when the motor rotation speed is increased with the switch 26 turned on, as shown in the area 222 in FIG. 12B, between the motor rotation speed fm4 and the motor rotation speed fm5 It becomes necessary to reduce the torque and keep the current ripple within a predetermined value. However, in curve 104 when switch 26 is turned off, DC side current gain Ig does not exceed reference value 105 between frequency f4 and frequency f5.

The motor drive device 200 according to the present embodiment turns off the switch 26 when the motor rotation speed is within a predetermined range and a torque of a predetermined value or more is required with the switch 26 turned on. control.

FIG. 13 is a flowchart showing an example of control of motor drive device 200 according to the present embodiment. The process shown in FIG. 13 is executed when the motor 60 starts rotating, and at step 130 the switch 26 is turned on.

At step 132, it is determined whether or not the motor rotation speed is within a predetermined range as indicated by the rotation speeds fm4 and fm5 in FIG. 12(B). However, if the motor rotation speed is not within the predetermined range, the procedure proceeds to 130 .

At step 134, it is determined whether or not the torque of the output shaft of the motor 60 is equal to or greater than a predetermined value. The output shaft torque is calculated based on the current value detected by the inverter current sensor 50 . Further, when the torque is equal to or higher than the predetermined value, for example, the motor rotation speed of the motor 60 and the torque of the output shaft are within the region 222 in FIG. 12(B). If in step 134 the torque of the output shaft of the motor 60 is greater than or equal to the predetermined value, the procedure proceeds to step 136 . If the torque of the output shaft of the motor 60 is less than the predetermined value in step 134, that is, if the motor rotation speed and the torque of the output shaft of the motor 60 are within the region 220 in FIG. .

At step 136, the switch 136 is turned off, and after that, when the rotation of the motor 60 is stopped, the processing ends.

As described above, according to the motor drive device 200 according to the present embodiment, the resonance circuit configured by the battery section inductor 18 and the battery section capacitor 20 is turned on and off by the switch 26 to set the first resonance frequency f1 to the first resonance frequency f1. The above wide frequency range can be the inverter switching frequency range in which the transistors 42 and 44 can be operated.

As described in the first embodiment, when determining the specifications of each of the smoothing capacitor 32, the battery section inductor 18, and the battery section capacitor 20, the formula for calculating the second resonance frequency f2 using the base speed 120 When (4) and (5) are used, the second resonant frequency f2A occurs at a frequency relatively close to the first resonant frequency f1A, narrowing the inverter switching frequency range in which the transistors 42 and 44 can be operated. However, in the present embodiment, the switch 26 is turned off to invalidate the resonant circuit composed of the battery section inductor 18 and the battery section capacitor 20, thereby expanding the inverter switching frequency range and widening the motor 60. It is possible to rotate with high torque within the range of motor rotation speed.

In addition, among the structures of claims, the DC power source is the battery 12, the drive circuit is the inverter 40, the first capacitor is the smoothing capacitor 32, the inductor is the battery section inductor 18, and the second capacitor is the battery section capacitor 20. In addition, the control section corresponds to the motor control device 80, and the switch corresponds to the switch 26, respectively.

10 DC power supply unit 12 battery 14 battery internal resistance 16 battery internal inductor 18 battery unit inductor 20 battery unit capacitor 26 switch 30 inverter unit 32 smoothing capacitor 40 inverters 42, 44 transistor 60 motor 62 coil 64 rotor 80 motor control device 100 motor drive Device 120 Base speed 122 Maximum mechanical rotation speed 200 Motor drive device 220, 222 Regions f1, f1A, f1B First resonance frequency f2, f2A, f2B Second resonance frequency f1' Third resonance frequency

Claims (5)

a driving circuit for generating a voltage applied to the windings of the motor by pulse width modulation of the power supplied from the DC power supply;
a first resonance circuit having a first capacitor that short-circuits a positive electrode and a negative electrode on the DC power supply side of the drive circuit;
a second resonant circuit having an inductor and a second capacitor connected in series with each other and short-circuiting a positive terminal and a negative terminal of the DC power supply;
a control unit for controlling the drive circuit to perform the pulse width modulation at a frequency different from the resonance frequency at which the impedance of the circuit including the first resonance circuit and the second resonance circuit is minimized;
a motor drive including; including a switch that turns on and off the connection between the positive terminal and the negative terminal of the DC power supply of the second resonant circuit,
When the switch is on, the resonant frequencies include a first resonant frequency and a second resonant frequency higher than the first resonant frequency, and when the switch is off, the resonant frequency is higher than the first resonant frequency. including a third resonance frequency that is higher and lower than the second resonance frequency;
The controller controls the frequency of the pulse width modulation to fall within a predetermined frequency range including the second resonance frequency when the rotation speed of the motor is within a predetermined range and the torque of the output shaft of the motor is equal to or greater than a predetermined torque. 2. The motor drive device according to claim 1, wherein the switch is turned off when 3. The method according to claim 2, wherein each of the capacitance C _Inv of the first capacitor, the capacitance C _Bat of the second capacitor, and the inductance L _Bat of the inductor is calculated by the following equation A based on the second resonance frequency f2. motor drive.

The second resonance frequency f2 is based on the base speed, which is the maximum rotational speed at which the motor can maintain the maximum torque, the polar logarithm of the motor, and the carrier frequency of the carrier wave related to the pulse width modulation, and the following equations B and C 4. The motor driving device according to claim 3, wherein the larger value of f2-1 and f2-2 calculated using
f2-1 = basal velocity / 60 x pole log number x 6 ... B
f2-2 = carrier frequency + base velocity / 60 x number of pole pairs x 3...C The second resonance frequency f2 is determined by the following equations D and E based on the maximum mechanical rotation speed, which is the maximum rotation speed of the motor, the pole logarithm of the motor, and the carrier frequency of the carrier wave related to the pulse width modulation. 4. The motor driving device according to claim 3, wherein the greater value of f2-1 and f2-2 calculated using the f2-1 and f2-2 is larger.
f2-1=maximum machine speed/60×number of pole pairs×6 …D
f2-2 = carrier frequency + maximum machine speed / 60 x pole logarithm x 3...E

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