JP7113095B2 - power switching circuit - Google Patents

Description

The present invention relates to power switching circuits, and more particularly to power switching circuits for electric propulsion vehicles.

Generally, an electric propulsion vehicle, such as a battery electric vehicle (BEV), has a power switching circuit that includes a three-phase inverter. A common DC bus based dual inverter has been proposed as one form of this three-phase inverter. In addition, switched batteries, which switch between series and parallel connections of two batteries, have been proposed as another power switching circuit for electric propulsion vehicles.

The low energy density (kWh/kg) of batteries is well known as an inherent drawback of BEVs. BEV weight and manufacturing costs increase as battery weight increases for extended range. Additionally, the weight of the traction motor and the power consumption of the inverter are increased due to this weight increase of the BEV.

A high density of fast charging spots is a very good solution for this problem. A common fast charging scheme is the off-board current control DC charging scheme, where the fast charging spot adjusts the DC charging current according to the BEV battery's demand. However, construction of this high-density fast charging network consisting of a large number of current-controlled DC charging spots is economically difficult due to high spot costs.

A grid-connected on-board integrated charger that uses the inductance of the traction motor and the inverter driving this traction motor as three boost rectifier choppers for regulating the battery charging current, to economically build a high-density fast-charging network. is an excellent proposal. For example, US Pat. No. 6,300,001 proposes a grid-connected on-board integrated charger using dual inverters based on a common DC bus. A dual inverter allows the elimination of the neutral release contactor required by typical on-board integrated chargers.

However, conventional grid-connected on-board integrated chargers have many problems that need to be resolved. The first problem is that a grid-connected on-board integrated charger must incorporate a PFC (Power Factor Correction Circuit) to improve the power factor. However, a 3-phase PFC circuit formed by a 3-phase boost chopper consisting of an open-ended 3-phase coil and a common DC bus-based dual inverter has not yet been proposed.

The second problem is that many off-board current-controlled DC charging fast charging spots have already been built. For this reason, a grid-connected on-board integrated charger must have the ability to both accept AC power from a three-phase grid and DC power from an off-board current-controlled DC charging spot. It is possible for a grid-connected on-board integrated charger to accept DC power. However, the three-phase coil of the traction motor as part of the boost chopper produces high resistive losses. For example, when the equivalent resistance of the 3-phase coil is 0.15 ohms and the DC charging current is 100A, a resistive loss of 1500W occurs.

A third problem is that in the near future it is expected that BEV batteries will be charged by external DC power sources such as wind power plants, solar power plants, fuel cell power plants and hybrid plants thereof. Power losses occur when the power of these external DC sources is converted to grid power. It is possible to directly charge the BEV's battery with an external DC power supply. However, the external DC power supply requires a DCDC converter for charging current control for each charging spot. As a result, the construction cost of the fast-charging network will skyrocket. After all, conventional grid-connected on-board integrated chargers are not yet intended to accept current-controlled DC power or current-uncontrolled DC power.

A fourth problem is that the BEV's fast charging time is extended when the battery is hot. For example, in quick charging after driving on a highway in summer, the quick charging time is extended to avoid high temperature deterioration of the battery. This problem is ameliorated by reducing the BEV's motor loss and inverter loss. However, conventional traction motor systems do not fully anticipate shortening the fast charging time by reducing these power losses.

Known boost choppers and known switched batteries reduce losses in traction motor systems. However, these boost choppers and switched batteries cause the DC link voltage applied to the inverter to fluctuate greatly. As a result, a step-down DCDC converter for low voltage battery charging that steps down this DC link voltage is adversely affected by this DC link voltage variation. Conventional step-down DCDC converters for low-voltage battery charging have not yet taken into account this DC link voltage variation.

A fifth problem is that the BEV's fast charging time is extended by low battery temperatures. For example, the fast charging time in severe winter must be extended to avoid permanent deterioration of the lithium-ion battery.

USP8415904

SUMMARY OF THE INVENTION It is an object of the present invention to provide a power switching circuit for an electric propulsion vehicle with greater convenience.

According to a first aspect of the invention, a three-phase grid-connected on-board integrated charger using a common DC bus based dual inverter has a three-phase grid charging mode and a current controlled DC charging mode. In both modes, the dual inverters are connected to the 3-phase grid or external DC power supply through a common plug. In the three-phase grid charging mode, the battery-side three-phase inverter of the dual inverter performs step-up rectification operation. A three-phase grid has a lower phase-to-phase voltage than the DC link voltage, which is the voltage of the DC bus to which the dual inverters are connected. The battery-side three-phase inverter, which operates as a boost rectifier in three-phase grid charging mode, can control the battery charging current according to the state of charge of the battery.

This allows the batteries to be charged by both existing current-controlled DC fast-charging spots and three-phase grid outlets, improving the convenience of electric propulsion vehicles while avoiding increased manufacturing costs. Furthermore, in DC charging mode, the traction motor's 3-phase coils do not consume power. In addition, the plug can also be connected to a non-current-controlled external DC power supply when the DC link voltage, which is the voltage of the DC bus to which the dual inverter is connected, is higher than the DC voltage of the external DC power supply. A three-phase inverter on the battery side as a boost chopper can regulate the battery charging current.

In one aspect, the battery-side 3-phase inverter is driven as a 3-phase phase correction (PFC) circuit in a 3-phase grid charging mode. This eliminates the need for the external DC power supply to incorporate a three-phase phase correction (PFC) circuit. As a result, a three-phase grid-connected on-board fast charging network with good power factor can be economically constructed.

In another aspect, a battery-side three-phase inverter driven as a three-phase PFC circuit in three-phase grid charging mode has a buck phase period driven as a buck chopper in addition to a buck phase period driven as a boost chopper. have Thereby, a high power factor can be realized. This three-phase PFC circuit with alternating boost and buck phase periods can be employed in other applications that require rectifying a three-phase grid voltage.

In another aspect, the battery is connected to the DC bus through a front end boost converter that boosts the battery voltage. This front-end boost converter regulates battery charging current in three-phase grid charging mode and current-uncontrolled DC charging schemes. This allows the battery to be charged even if the externally applied grid voltage or DC voltage is higher than the battery voltage.

For example, a deeply discharged battery has a voltage lower than its rated voltage. In addition, Chinese and European three-phase grid voltages have high phase-to-phase voltage values. Additionally, wind power plants, solar power plants, and fuel cell plants can have a higher DC voltage than the electric vehicle battery voltage for loss reduction. The step-down charging operation of the front-end boost converter allows fast charging under these conditions.

In another aspect, when the phase-to-phase voltage of the three-phase grid is lower than the battery voltage in the three-phase grid charging mode, the step-down charging operation of the front end boost converter is deactivated. Thereby, the loss of the front end boost converter can be reduced.

In another aspect, the front end boost converter comprises a single reactor boost switched battery. As a result, losses in the inverter and battery can be reduced. This single reactor boost switched battery can be employed by other electric propulsion vehicles.

In another aspect, the front end boost converter has a voltage balance mode that reduces the voltage difference between the two batteries before parallel connecting the two batteries if the voltage difference between the two batteries exceeds a predetermined value. . Thereby, the loss of the front end boost converter can be reduced.

In another aspect and a second aspect of the invention, the smoothing capacitor that applies the DC link voltage to the inverter is connected to the low voltage battery through an isolated bi-directional DCDC converter operable as a chopper. This converter has two H-bridges that are separately connected to the two coils of the transformer. The H-bridge on the primary side connected to the smoothing capacitor has a pre-charge mode for charging the smoothing capacitor in addition to a step-down charging mode for charging the low-voltage battery. This can prevent the so-called inrush current of the smoothing capacitor.

In one aspect, the H-bridge on the primary side and the leakage inductance of the primary coil are driven as a buck chopper in buck charging mode. This makes it possible to absorb fluctuations in the DC link voltage with a simple circuit topology.

In another aspect, the H-bridge on the primary side and the leakage inductance of the primary coil are driven as a boost chopper in precharge mode. This makes it possible to prevent inrush current with a simple circuit topology.

In another aspect and a third aspect of the invention, the inverter supplies single-phase alternating current from the battery to the three-phase coil when the battery temperature is low. A three-phase coil of a stationary three-phase motor can be regarded as a simple inductive load. As a result, the battery alternately repeats the charging operation and the discharging operation, and is efficiently heated. As a result, the electrolyte in the battery heats up and the battery resistance drops rapidly. This single-phase alternating current forms two rotating magnetic fields that rotate in opposite directions. Therefore, the three-phase motor does not generate starting torque.

In another aspect, each of the three H-bridges of the common DC bus based dual inverter consists of a fixed potential leg and a PWM leg. The fixed potential leg and PWM leg of each H-bridge alternate every 180 electrical degrees. This nearly halves the switching losses of the dual inverter. Furthermore, the upper arm transistor of the fixed potential leg is always turned on. This PWM control of dual inverter is called upper arm conduction single PWM method. As a result, inverter loss is reduced due to ringing surge voltage reduction.

In another preferred embodiment, the H-bridge outputting the phase voltages of minimum amplitude is deactivated and is called the dormant H-bridge. The two legs of the dormant H-bridge each have their respective upper and lower arm switches turned off. This PWM control method of dual inverter is called double H-bridge mode. Thereby, battery loss can be reduced. This double H-bridge mode can be employed by other dual inverters driving 3-phase motors.

In another aspect, H-bridges that output phase voltages of maximum amplitude are referred to as voltage-fixed H-bridges. This potential fixed H-bridge consists of two fixed potential legs that output voltages opposite to each other. In addition, the front end boost converter connecting the battery and the dual inverter outputs a DC link voltage equal to the phase voltage of maximum amplitude. This DC link voltage is applied to one phase coil through a potential clamp H-bridge. Furthermore, the output voltages of the two fixed potential legs of the fixed potential H-bridge are switched periodically. This inverter control method is called boost double H-bridge mode. Thereby, the power loss of the battery and the inverter can be reduced. This boost double H-bridge mode can be employed by other dual inverters driving 3-phase motors.

In another aspect, the double H-bridge mode and the boosted double H-bridge mode are performed together. This mode is called single H-bridge mode. According to this single H-bridge mode, only one H-bridge of the dual inverter is PWM controlled. As a result, battery and inverter power losses are further reduced. This single H-bridge mode can be employed by other dual inverters driving 3-phase motors.

In another aspect, the three H-bridges of the dual inverter are each controlled by a space vector PWM method (SVPWM), and the current supply periods of the three H-bridges are arranged so as not to overlap as much as possible. Thereby, battery loss can be reduced. This current sharing method can be employed by other dual inverters driving three-phase motors.

FIG. 1 is a circuit diagram showing a power switching circuit of an embodiment. FIG. 2 is a waveform diagram of three-phase grid currents. FIG. 3 is a timing chart showing power factor correction operation of a dual inverter operating as a three-phase PFC circuit. FIG. 4 is a circuit diagram showing the current flow in the dual inverter during phase periods P1 and P2. FIG. 5 is a circuit diagram showing current flow in the dual inverter during phase period P3. FIG. 6 is a circuit diagram showing current flow in the dual inverter during phase period P4. FIG. 7 is a circuit diagram showing the current flow in the dual inverter during phase periods P5 and P6. FIG. 8 is a circuit diagram showing current flow in the dual inverter during phase period P7. FIG. 9 is a circuit diagram showing current flow in the dual inverter during phase period P8. FIG. 10 is a circuit diagram showing current flow in the dual inverter during phase periods P9 and P10. FIG. 11 is a circuit diagram showing current flow in the dual inverter during phase period P11. FIG. 12 is a circuit diagram showing current flow in the dual inverter during phase period P12. FIG. 13 is a circuit diagram showing a modified power switching circuit. FIG. 14 is a block diagram showing an electromagnetic brake device. FIG. 15 is a flow chart showing control of the electromagnetic brake device. FIG. 16 is a circuit diagram showing an isolated bidirectional DCDC converter. FIG. 17 is a block circuit diagram for explaining the single-phase battery heating method. FIG. 18 is a timing chart showing waveforms of fundamental frequency components of single-phase AC current and single-phase AC voltage employed in the single-phase battery heating method. FIG. 19 is a diagram showing the state of the dual inverter in the upper arm conduction type single PWM method. FIG. 20 is a diagram showing the output voltage of the H-bridge driven by the upper arm conduction type single PWM method. FIG. 21 is a schematic circuit diagram showing the current supply period of the H-bridge driven by the double PWM method. FIG. 22 is a schematic circuit diagram showing the current regeneration period of the H-bridge driven by the double PWM method. FIG. 23 is a schematic circuit diagram showing the current supply period of the H-bridge driven by the single PWM method. FIG. 24 is a schematic circuit diagram showing the freewheeling period of the H-bridge driven by the single PWM method. FIG. 25 is a timing chart showing the fundamental frequency components of the U-phase voltage applied to the U-phase coil and the U-phase current supplied to the U-phase coil. FIG. 26 is a schematic circuit diagram showing the U-phase current flowing through the U-phase coil during the phase period P1. FIG. 27 is a schematic circuit diagram showing the U-phase current flowing through the U-phase coil during the phase period P2. FIG. 28 is a schematic circuit diagram showing the U-phase current flowing through the U-phase coil during the phase period P3. FIG. 29 is a schematic circuit diagram showing the U-phase current flowing through the U-phase coil during the phase period P4. FIG. 30 is a vector diagram showing the phase domain in which the synthesized rotating voltage vector formed by the double H-bridge mode rotates. FIG. 31 is a diagram showing the state of the dual inverters in double H-bridge mode. FIG. 32 is a diagram showing the waveform of the fundamental frequency component of the 3-phase grid voltage in the boost double H-bridge mode. FIG. 33 is a diagram showing the state of the dual inverter in the boost double H-bridge mode. FIG. 34 is a diagram showing the state of each leg in single H-bridge mode. FIG. 35 is a timing chart showing the arrangement of current supply periods for each phase given to the dual inverter driven by the current distribution method. FIG. 36 is a timing chart showing a modification of the current spreading method.

A preferred embodiment of a power switching circuit used for a battery electric vehicle (BEV) is described with reference to the drawings. This power switching circuit can be used in other electric propulsion vehicles.
POWER SWITCHING CIRCUIT TOPOLOGY FIG. 1 is a circuit diagram showing a power switching circuit that doubles as an on-board integrated charger. This power switching circuit, called a three-phase motor drive, has a single reactor step-up switched battery 100, a common DC bus-based dual inverter 200, and an isolated bidirectional DCDC converter 300. The controller 9, smoothing capacitor 13, and Further includes a 3-phase plug 400 .

The step-up switched battery 100 is connected to a high-level DC bus 81 through a high-side main relay 61 and connected to a low-level DC bus 82 through a low-side main relay 62 . A smoothing capacitor 13 connected to DC buses 81 and 82 has a DC link voltage VC. Main relays 61 and 62 are safety switches for isolating boost switched battery 100 from other circuit parts of the power switching circuit. Each switch of boost switched battery 100 and dual inverter 200 consists of an IGBT with an anti-parallel diode. Controller 9 controls boost switched battery 100 , dual inverter 200 , and DCDC converter 300 .

Topology of Boost Switched Battery 100 A boost switched battery 100 , which is a DC power supply circuit, consists of a low-side battery 1 , a high-side battery 2 , and a front-end boost converter 10 . A front-end boost converter 10 called a connection switching circuit has both a connection switching function for the batteries 1 and 2 and a bidirectional DCDC converter function. The converter 10 has a series switch 3, a low side parallel switch 4, a high side parallel switch 5, an output switch 6, a reactor 7, a surge absorption diode, and three safety fuses (501-503). High level DC bus 81 is connected to low level DC bus 82 through output switch 6 , battery 2 , reactor 7 , series switch 3 and battery 1 . A connection point between the negative electrode of the battery 2 and the reactor 7 is connected to the negative electrode of the battery 1 through the parallel switch 4 .

A connection point between the positive electrode of the battery 1 and the series switch 3 is connected to the positive electrode of the battery 2 through the parallel switch 5 . A connection point between the reactor 7 and the series switch 3 is connected to the cathode electrode of the surge absorption diode, and the anode electrode of the surge absorption diode is connected to the negative electrode of the battery 1 . Batteries 1 and 2 have rated voltages equal to each other. The operation modes of front-end boost converter 10 include parallel mode, series mode, transient mode, voltage balance mode, boost discharge mode, and buck charge mode.

Parallel and Series Modes Parallel and series modes can be employed in both discharging and charging operations of the batteries 1 and 2 . In parallel mode, series switch 3 is turned off and parallel switches 4 and 5 are turned on. Batteries 1 and 2 are thereby connected in parallel. The DC link voltage VC will be equal to the voltage of one battery. In series mode, series switch 3 is turned on, parallel switches 4 and 5 are turned off, and batteries 1 and 2 are connected in series. Therefore, the DC link voltage VC is the sum of the voltages of the two batteries.

TRANSIENT MODES Transient modes, which are executed during the short transition period between parallel mode and series mode, consist of transient discharge mode and transient charge mode. In transient discharge mode, parallel switches 4 and 5 are turned off and series switch 3 is PWM controlled. The PWM duty ratio of the series switch 3 is slowly changed from 0 to 1 in the parallel-to-series transient discharge mode where switching from parallel mode to series mode is performed. In the series-to-parallel transient discharge mode where switching from series mode to parallel mode takes place, the PWM duty ratio of series switch 3 is slowly changed from 1 to 0.

During the ON period when the series switch 3 is ON, the DC link voltage VC is the sum of the voltages of the batteries 1 and 2 minus the back electromotive force of the reactor 7 . When the series switch 3 is turned off, the magnetic energy of the reactor 7 is consumed by current flowing through the surge absorption diode, the reactor 7, the battery 2, and the output switch 6. Ultimately, the transient discharge mode slowly changes the DC link voltage VC.

In transient charging mode, parallel switches 4 and 5 are synchronously PWM controlled. In parallel-to-series transient charging mode, where switching from parallel mode to series mode occurs, the PWM duty ratio of parallel switches 4 and 5 is slowly changed from 1 to 0. In the series-to-parallel transient charging mode where switching from series mode to parallel mode takes place, the PWM duty ratio of parallel switches 4 and 5 is slowly changed from 0 to 1.

During the off period when the parallel switches 4 and 5 are turned off, the magnetic energy of the reactor 7 is increased by the charging current charging the batteries 1 and 2 in series. In other words, reactor 7 suppresses an increase in charging current flowing through batteries 1 and 2 . During the ON period when the parallel switches 4 and 5 are turned on, the DC link voltage VC is applied to the batteries 1 and 2 respectively. This causes the charging current of batteries 1 and 2 to surge.

However, the magnetic energy of the reactor 7 causes a freewheeling current to flow through the antiparallel diode of the series switch 3 , the parallel switch 5 and the battery 2 . Similarly, the magnetic energy of the reactor 7 causes a freewheeling current to flow through the antiparallel diode of the series switch 3 , the battery 1 and the parallel switch 4 . These freewheeling currents suppress the surge in charging current of batteries 1 and 2 described above. After all, in the transient charging mode, sudden changes in charging current are avoided.

Voltage Balance Mode Voltage balance mode is implemented when batteries 1 and 2 have different voltages. A short circuit current circulates through the two parallel switches 4 and 5 in the batteries 1 and 2 when the voltage difference between the batteries 1 and 2 exceeds a predetermined value in parallel mode. As a result, the higher voltage battery charges the lower voltage battery. When this short circuit current exceeds a predetermined value, the power loss of batteries 1 and 2 increases. According to this voltage balance mode, the voltage difference between the two batteries 1 and 2 is detected before the parallel mode is started. When this voltage difference exceeds a predetermined value, the transistors of at least one parallel switch are inhibited from turning on, at least during the initial period of parallel mode. For example, the case where battery 1 is higher than battery 2 by a predetermined voltage value or more before starting the parallel mode will be described below.

Turning on of the transistor of the parallel switch 4 is prohibited in the battery discharge mode. Thus, only battery 1 supplies current to dual inverter 200 . After the voltages of batteries 1 and 2 are approximately equal due to the voltage drop of battery 1, the transistor of parallel switch 4 is turned on. However, the transistors of the parallel switch consisting of IGBTs do not have to be turned on because of the parasitic diodes.

Turning on of the transistor of the parallel switch 5 is prohibited in the battery charging mode. The battery 2 is thereby charged through the parallel switch 4 . After the voltages of batteries 1 and 2 are approximately equal due to the voltage rise of battery 2, the transistor of parallel switch 5 is turned on. The voltages of batteries 1 and 2 are thereby equalized. If the battery 2 is higher than the battery 1 by a predetermined value or more before starting the parallel mode, the control opposite to the above control is executed. According to this voltage balance mode, wasteful circulating current (short-circuit current) flowing immediately after starting the parallel mode can be prevented.

Boost Discharge Mode In the boost discharge mode in which the boost switched battery 100 applies a boost voltage to the dual inverter 200, the series switch 4 is turned on and the parallel switches 4 and 5 are synchronously PWM-controlled. Batteries 1 and 2 respectively supply circulating current to reactor 7 when parallel switches 4 and 5 are turned on. Thereby, the magnetic energy of the reactor 7 increases. When the parallel switches 4 and 5 are turned off, the DC link voltage VC is the sum of the voltage sum of the batteries 1 and 2 and the reactor 7 voltage. When battery 1 has voltage V1 and battery 2 has voltage V2, the boost ratio (VC/(V1+V2)) is controlled by adjusting the PWM duty ratio of parallel switches 4 and 5. FIG.

When the voltage difference between batteries 1 and 2 exceeds a predetermined value in boost discharge mode, a short circuit current circulates through battery 1, parallel switch 5, battery 2, and parallel switch 4 immediately after parallel switches 4 and 5 are turned on. do. As a result, the higher voltage battery charges the lower voltage battery. To inhibit this circulating current, a voltage balance mode is implemented in the boost discharge mode when the voltage difference between batteries 1 and 2 exceeds a predetermined value. For example, when the voltage of battery 2 is higher than the voltage of battery 1 by a predetermined value or more, the PWM switching of parallel switch 5 is suspended. As a result, only the battery 1 is discharged.
After the voltages of batteries 1 and 2 are approximately equal, parallel switches 4 and 5 are synchronously PWM controlled.

Step-Down Charging Mode In the step-down charging mode for charging the batteries 1 and 2, the series switch 4 is turned on, the parallel switches 4 and 5 are turned off, and the output switch 6 is PWM controlled. When output switch 6 is turned on, charging current flows through battery 2, reactor 7, series switch 3, and battery 1, and batteries 1 and 2 are charged in series. The reactor 7 accumulates magnetic energy. When the output switch 6 is turned off, the magnetic energy of the reactor 7 circulates the charging current through the reactor 7, the series switch 3, the parallel switch 5, and the battery 2. In addition, this magnetic energy circulates the charging current through reactor 7, series switch 3, battery 1, and parallel switch 4. FIG. Consequently, the magnetic energy of reactor 7 charges batteries 1 and 2 in parallel. The step-down ratio ((V1+V2)/VC) is adjusted by the PWM duty ratio of the output switch 6.

Advantages of Boost Switched Battery 100 In battery electric vehicle (BEV) applications, the parallel mode is employed in the low speed range, the series mode is employed in the medium speed range, and the boost discharge mode and the buck charge mode are employed in the high speed range. be. The boost discharge mode achieves an increase in the number of turns of the three-phase coil 50 and a reduction in inverter current. As a result, inverter loss is significantly reduced compared to non-boosted switched batteries. For example, when the maximum boost ratio is 2, the inverter current is half that of a non-boost type switched battery, and the inverter loss is 1/4.

Additionally, this inverter current reduction reduces battery losses in parallel and series modes. For example, when the maximum step-up ratio is 2, the amplitude of the power supply current supplied from batteries 1 and 2 to the inverter is halved. However, in the low-speed region where the back electromotive force of the three-phase coil 50 is low, the supply time of the power supply current is almost doubled due to the reduction of the DC link voltage VC. As a result, the boost switched battery 100 can have half the battery loss in the non-boost region consisting of parallel mode and series mode compared to the non-boost switched battery.

Furthermore, the parallel mode of a switched battery has approximately 1/4 the battery resistance compared to its series mode. However, the supply time of the power supply current supplied from the batteries 1 and 2 to the inverter is almost doubled due to the reduction of the DC link voltage VC. As a result, parallel mode can have about half the battery loss compared to series mode.

Furthermore, reactor 7 of boost switched battery 100 shown in FIG. 1 does not generate resistive losses in parallel mode. Further, the single reactor boost switched battery 100 shown in FIG. 1 can have about 70% weight and loss compared to reactors compared to a conventional double reactor boost switched battery with two reactors. can.

Topology of Dual Inverter 200 A traction motor consisting of a three-phase synchronous motor has open-ended three-phase coils 50 as stator coils. The three-phase coil 50 consists of a U-phase coil 5U, a V-phase coil 5V, and a W-phase coil 5W. The common DC bus based dual inverter 200 consists of a grid side 3-phase inverter 30 and a battery side 3-phase inverter 40 . The inverter 30 consists of a U-phase leg 3U, a V-phase leg 3V, and a W-phase leg 3W. The inverter 40 consists of a U-phase leg 4U, a V-phase leg 4V, and a W-phase leg 4W. In normal operation, the upper and lower arm switches of each leg are switched complementarily. The three AC terminals of inverter 30 are separately connected to the three terminals (X, Y, Z) of plug 400 for grid connection.

The inverter 40 is individually connected to one end of each of the three phase coils (5U, 5V, and 5W), and the inverter 30 is individually connected to each other end of the three phase coils (5U, 5V, and 5W). . The inverter 30 consists of three upper arm switches (31, 33, 35) and three lower arm switches (32, 34, 36). The inverter 40 consists of three upper arm switches (41, 43, 45) and three lower arm switches (42, 44, 46). A U-phase H-bridge consisting of legs 3U and 4U controls the U-phase current IU supplied to the phase coil 5U. A V-phase H-bridge consisting of legs 3V and 4V controls the V-phase current IV supplied to phase coil 5V. A W-phase H bridge consisting of legs 3W and 4W controls the W-phase current IW supplied to the W-phase coil 5W. A torque control operation of dual inverter 200 for generating motor torque will be described later.

Onboard Integrated Charger Topology Boost switched battery 100, dual inverter 200, and three-phase coil 50 form an onboard integrated charger. The mode of operation of this on-board integrated charger is described below. The modes of operation include current controlled DC charging mode, current non-controlled DC charging mode, grid charging mode, and reverse transmission mode.

Current-controlled DC charging mode Conventional current-controlled DC fast charging spots incorporate a DCDC converter that can control the DC charging current according to the BEV's requirements . This current controlled DC charging mode is implemented when the two terminals (X, Y) of the plug 400 are connected to a current controlled DC fast charging spot. Series switch 3 and output switch 6 of boost switched battery 100 are turned on, and batteries 1 and 2 are charged in series. The switching action of the two three-phase inverters 30 and 40 of dual inverter 200 is essentially stopped. However, synchronous rectification operation is possible.

A DC charging current flows from the terminal X of the charging plug 400 through the upper arm switch 31 to the high level DC bus 81 and then charges the two batteries 1 and 2 . DC charging current returned from battery 1 to low level DC bus 82 returns to terminal Y of charging plug 400 through lower arm switch 34 . The DCDC converter of the current controlled DC fast charging spot controls the DC charging current according to the controller 9 request.

The advantage of this current controlled DC charging mode is that no DC charging current flows through the three-phase coil 50 . As a result, it is possible to improve the DC charging efficiency and suppress the temperature rise of the three-phase coil 50 . For example, when the average resistance of the 3-phase coil 50 is 0.15 ohms and the average charging current is 100A, the power loss of the 3-phase coil 50 is 1500W.

High Voltage Current Uncontrolled DC Charging Mode The various power plants previously described can apply a DC voltage between the two terminals (X, Y) of the charging plug 400 . When the DC voltage is higher than the sum of the voltages of batteries 1 and 2, the DC voltage is applied to boost switched battery 100 through upper arm switch 31 and lower arm switch 34 . The step-up switched battery 100 performs a step-down charging mode, and the charging current of batteries 1 and 2 is controlled by PWM control of output switch 6 . The advantage of this current-uncontrolled DC charging mode is that no DC charging current flows through the three-phase coil 50 . As a result, it is possible to improve the DC charging efficiency and suppress the temperature rise of the three-phase coil 50 .

Low Voltage Current Uncontrolled DC Charging Mode This low voltage current uncontrolled DC charging mode is operated when the DC voltage of the current uncontrolled power plant is lower than the sum of the voltages of batteries 1 and 2 . This DC voltage is applied to boost switched battery 100 after being boosted by a boost chopper comprising U-phase coil 5U and leg 4U. When the lower arm switch 42 is turned on, the U-phase current IU circulating in the U-phase coil 5U, the lower arm switch 42, the lower arm switch 34, and the power plant increases, and the magnetic energy of the U-phase coil 5U increases. When the lower arm switch 42 is turned off, the U-phase current IU returns to the power plant through the upper arm switch 41, the boost switched battery 100 and the lower arm switch 34. Batteries 1 and 2 are accordingly charged. The charging current is adjusted by controlling the PWM duty ratio of the lower arm switch 42 .

This low-voltage, current-uncontrolled DC charging mode is suitable for charging BEVs from low-voltage DC power plants, such as domestic solar panels. On the other hand, the high voltage current uncontrolled DC charging mode is suitable for charging BEVs from high voltage DC power plants such as large scale solar plants.

3-Phase Grid Charging Mode When a 3-phase grid voltage is applied to the plug 400, a 3-phase grid charging mode is performed. The three-phase sinusoidal grid voltage applied to plug 400 consists of U-phase voltage VU1, V-phase voltage VV1, and W-phase voltage VW1. Batteries 1 and 2 are connected in series. Three phase-to-phase voltages (VU1-VV1, VV1-VW1, and VW1-VU1) are lower than the DC link voltage VC. Dual inverter 200 and open-ended three-phase coil 50 form a three-phase power factor correction circuit (PFC circuit).

3-Phase PFC Mode in 3-Phase Grid Charge Mode The 3-phase PFC mode, which is the power factor correcting operation of the 3-phase PFC circuit formed by the dual inverter 200, is described with reference to FIGS. 2-12. FIG. 2 shows waveforms of a U-phase current IU, a V-phase current IV, and a W-phase current IW flowing from the plug 400 to the three-phase coil 50 during one cycle period of the three-phase grid voltage. U-phase current IU, V-phase current IV, and W-phase current IW are called charging phase currents.

Each charging phase current (IU, IV, and IW) generally has a different amplitude value than each phase grid voltage (VU1, VV1, and VW1) of the three-phase grid voltage. However, in FIG. 2 each charging phase current is shown to have an amplitude equal to each phase grid voltage. A U-phase current IU having a sinusoidal waveform in phase with the U-phase voltage VU1 flows through the U-phase coil 5U. A V-phase current IV having a sinusoidal waveform in phase with the V-phase voltage VV1 flows through the V-phase coil 5V. A W-phase current IW having a sinusoidal waveform in phase with the W-phase voltage VW1 flows through the W-phase coil 5W.

This one cycle period is divided into 12 phase periods P1-P12. Inverter 40 forms a chopper circuit together with three-phase coil 50 . Legs 4U, 4V, and 4W of inverter 40 have step-up mode B, rest mode OFF, and step-down mode D in three-phase PFC mode. FIG. 3 is a diagram illustrating mode selection for legs (4U, 4V, and 4W) in phase periods P1-P12. 4-12 are diagrams showing phase current flows in the phase periods P1-P12. In sleep mode OFF, leg PWM switching is stopped.

FIG. 4 shows the current flow during the phase periods (P1, P2). U-phase coil 5U and U-phase leg 4U form a U-phase boost chopper. The switch 42 is PWM controlled. The U-phase current IU is supplied to the smoothing capacitor 13 while the switch 42 is off. U-phase current IU returning from smoothing capacitor 13 returns to the grid through V-phase lower arm switch 34 . Similarly, W-phase coil 5W and W-phase leg 4W form a W-phase boost chopper. The switch 46 is PWM controlled. The W-phase current IW is supplied to the smoothing capacitor 13 while the switch 46 is off. W-phase current IW returning from smoothing capacitor 13 returns to the grid through V-phase lower arm switch 34 .

FIG. 5 shows the current flow during phase period P3. Switch 42 continues PWM control. A W-phase coil 5W and a W-phase leg 4W form a W-phase step-down chopper. The switch 45 is PWM controlled. The W-phase current IW is supplied from the smoothing capacitor 13 to the grid through the W-phase coil 5W while the switch 45 is on. The magnetic energy of the W-phase coil 5 W supplies a freewheeling current through the switch 46 during the off period of the switch 45 .

FIG. 6 shows the current flow during phase period P4. Switch 42 continues PWM control. A V-phase coil 5V and a V-phase leg 4V form a V-phase step-down chopper. The switch 43 is PWM controlled. The V-phase current IV is supplied from the smoothing capacitor 13 to the grid through the V-phase coil 5V while the switch 43 is on. During the OFF period of the switch 43 , the magnetic energy of the V-phase coil 5V supplies a freewheeling current through the switch 44 .

FIG. 7 shows the current flow during the phase periods (P5, P6). Switch 42 continues PWM control. A V-phase coil 5V and a V-phase leg 4V form a V-phase boost chopper. The switch 44 is PWM controlled. The V-phase current IV is supplied to the smoothing capacitor 13 while the switch 44 is off. V-phase current IV returning from smoothing capacitor 13 returns to the grid through W-phase lower arm switch 36 .

FIG. 8 shows the current flow during phase period P7. Switch 44 continues PWM control. U-phase coil 5U and U-phase leg 4U form a U-phase step-down chopper. The switch 41 is PWM controlled. The U-phase current IU is supplied to the grid from the smoothing capacitor 13 through the U-phase coil 5U while the switch 41 is on. The magnetic energy of the U-phase coil 5U supplies a freewheeling current through the switch 42 during the OFF period of the switch 41 .

FIG. 9 shows the current flow during phase period P8. Switch 44 continues PWM control. A W-phase coil 5W and a W-phase leg 4W form a W-phase step-down chopper. The switch 45 is PWM controlled. The W-phase current IW is supplied to the grid from the smoothing capacitor 13 through the W-phase coil 5W while the switch 45 is on. The magnetic energy of the W-phase coil 5 W supplies a freewheeling current through the switch 46 during the off period of the switch 45 .

FIG. 10 shows the current flow during the phase periods (P9, P10). Switch 44 continues PWM control. A W-phase coil 5W and a W-phase leg 4W form a W-phase boost chopper. The switch 46 is PWM controlled. The W-phase current IW is supplied to the smoothing capacitor 13 while the switch 46 is off. W-phase current IW returning from smoothing capacitor 13 returns to the grid through U-phase lower arm switch 32 .

FIG. 11 shows the current flow during phase period P11. Switch 46 continues PWM control. A V-phase coil 5V and a V-phase leg 4V form a V-phase step-down chopper. The switch 43 is PWM controlled. The V-phase current IV is supplied from the smoothing capacitor 13 to the grid through the V-phase coil 5V while the switch 43 is on. During the OFF period of the switch 43 , the magnetic energy of the V-phase coil 5V supplies a freewheeling current through the switch 44 .

FIG. 12 shows the current flow during phase period P12. Switch 46 continues PWM control. U-phase coil 5U and U-phase leg 4U form a U-phase step-down chopper. The switch 41 is PWM controlled. The U-phase current IU is supplied to the grid from the smoothing capacitor 13 through the U-phase coil 5U while the switch 41 is on. The magnetic energy of the U-phase coil 5U supplies a freewheeling current through the switch 42 during the OFF period of the switch 41 . After all, according to this 3-phase PFC circuit, it is understood that two phase coils of the 3-phase coil 50 are used as reactors of the boost chopper or the step-down chopper in each phase period.

In the three-phase grid charging mode, the charging currents charging batteries 1 and 2 are regulated by the boost chopper operation of the three-phase inverter 40 or the step-down charging mode of the boost switched battery 100 . The step-down charging mode of boost switched battery 100 can regulate this charging current when the sum of the voltages of batteries 1 and 2 is lower than the phase-to-phase voltage of the three-phase grid (VU1-VV1, VV1-VW1, VW1-VU1). preferred. In this step-down charging mode, the charging current adjusted by PWM control of the output switch 6 preferably matches the step-up current output by the three-phase inverter 40 . As a result, ripples in the DC link voltage VC can be reduced. This charge current regulation by the boost switched battery 100 is preferred in a three-phase grid charging mode using a class 400V three-phase grid.

Chopper operation of the three-phase inverter 40 preferably regulates this charging current when the sum of the voltages of batteries 1 and 2 is higher than the phase-to-phase voltages (VU1-VV1, VV1-VW1, VW1-VU1) of the three-phase grid. be. The boost switched battery 100 is operated in series mode and the DC link voltage VC is equal to the sum of the voltages of batteries 1 and 2. This charging current regulation by the 3-phase inverter 40 is preferred in the 3-phase grid charging mode using a class 200V 3-phase grid.

Single Phase Grid Charging Mode The single phase grid charging mode will now be described. A single-phase grid voltage is applied between the two terminals (X and Y) of plug 400 . This single phase grid voltage is lower than the DC link voltage VC. Dual inverter 200 and three-phase coil 50 form a single-phase PFC circuit. The front end boost converter 10, which implements the buck charging mode, controls the charging current. The step-down operation of front-end boost converter 10 is the same in three-phase grid charging mode and single-phase grid charging mode.

Single-Phase PFC Mode A single-phase PFC mode operated by a single-phase PFC circuit is described. Phase voltage VU1 is higher than phase voltage VV1 during the positive half cycle of the single-phase grid voltage applied across terminals (X, Y). Conversely, phase voltage VU1 is lower than phase voltage VV1 during the negative half cycle of the single-phase grid voltage. During the positive half cycle period, the U-phase boost chopper consisting of the U-phase coil 5U and the U-phase leg 4U is operated, and the lower arm switch 42 is PWM-controlled. When switch 42 is turned on, current IU flows through U-phase coil 5U, switch 42, and switch 34, and U-phase coil 5U accumulates magnetic energy. When switch 42 is turned off, current IU flows through U-phase coil 5U, switch 41, smoothing capacitor 13 and switch 34, and smoothing capacitor 13 is charged. PWM control of switch 42 causes the waveform of the charging current to be a sinusoidal waveform that matches the positive half-cycle sinusoidal waveform of the single-phase grid voltage.

During the negative half cycle period, a V-phase boost chopper consisting of a V-phase coil 5V and a V-phase leg 4V is operated, and the lower arm switch 44 is PWM-controlled. When switch 44 is turned on, current IV flows through V-phase coil 5V, switch 44 and switch 32, and V-phase coil 5V accumulates magnetic energy. When switch 44 is turned off, current IV flows through V-phase coil 5U, switch 43, smoothing capacitor 13, and switch 32, and smoothing capacitor 13 is charged. PWM control of switch 44 causes the waveform of the charging current to be a sinusoidal waveform that matches the negative half-cycle sinusoidal waveform of the single-phase grid voltage.

3-Phase Reverse Transmission Mode A 3-phase reverse transmission mode for transmitting power from boost switched battery 100 back to the 3-phase grid is described. The DC link voltage VC is higher than the peak value of the phase-to-phase voltage of the 3-phase grid. The front end boost converter 10 performs a boost discharge mode when the sum of the voltages of batteries 1 and 2 is below this peak value, and a series mode when the sum of the voltages of batteries 1 and 2 is above this peak value. . A 3-phase step-down chopper comprising a 3-phase coil 50 and a 3-phase inverter 40 supplies 3-phase AC current through a plug 400 to the 3-phase grid. Three-phase inverter 30 is stopped. This three-phase step-down chopper also serves as a PFC circuit.

Leg 4U and U-phase coil 5U form a U-phase step-down chopper, leg 4V and V-phase coil 5V form a V-phase step-down chopper, and leg 4W and W-phase coil 5W form a W-phase step-down chopper. The operation of the 3-phase buck chopper is explained with reference to FIG. The 3-phase buck chopper supplies 3-phase sinusoidal currents (IU, IV, IW) to the plug 400 in phase opposition to the 3-phase grid voltage. In other words, the 3-phase currents (IU, IV, IW) are out of phase with the 3-phase grid voltages (VU1, VV1, VW1) by an electrical angle of 180 degrees.

In the phase period (P12, P1-P3) shown in FIG. 2, the lower arm switch 44 of the V-phase leg 4V is turned on, and the upper arm switch 41 of the U-phase leg 4U and the upper arm switch 45 of the W-phase leg 4W are PWM. controlled. During the phase period (P4-P7), the lower arm switch 46 of the W-phase leg 4W is turned on, and the upper arm switch 41 of the U-phase leg 4U and the upper arm switch 43 of the V-phase leg 4V are PWM-controlled. During the phase period (P8-P11), the lower arm switch 42 of the U-phase leg 4U is turned on, and the upper arm switch 43 of the V-phase leg 4V and the upper arm switch 45 of the W-phase leg 4W are PWM-controlled.

Single-Phase Reverse Power Transmission Mode A single-phase reverse power transmission mode for powering the boost switched battery 100 back to the single-phase grid will be described. The DC link voltage VC selects a suitable discharge mode according to the single-phase grid voltage. The terminals (X, Y) of plug 400 are connected to the single-phase grid. U-phase coil 5U and U-phase leg 4U form a U-phase step-down chopper. A V-phase coil 5V and a V-phase leg 4V form a V-phase step-down chopper. The three-phase inverter 30 and the W-phase leg 4W are stopped. The U-phase step-down chopper and the V-phase step-down chopper also serve as a single-phase PFC circuit.

During the positive half cycle period of the single-phase grid voltage in which the potential of the terminal X is higher than the potential of the terminal Y, the U-phase boost chopper consisting of the U-phase coil 5U and the U-phase leg 4U is operated, and the upper arm switch 41 is PWM-controlled. be done. The lower arm switch 44 of the V-phase leg 4V is turned on. During the negative half cycle period of the single-phase grid voltage in which the potential of the terminal X is lower than the potential of the terminal Y, the V-phase boost chopper consisting of the V-phase coil 5V and the V-phase leg 4V is operated, and the upper arm switch 43 is PWM-controlled. be done. The lower arm switch 42 of the U-phase leg 4U is turned on.

MODIFICATION OF ON-BOARD INTEGRATED CHARGER A modification of the on-board integrated charger shown in FIG. 1 will be described with reference to FIG. The integrated on-board charger shown in FIG. 13 uses one battery 1 instead of the boost switched battery 100 of the integrated on-board charger shown in FIG. The integrated onboard charger of FIG. 13 performs essentially the same operation as the integrated onboard charger of FIG. To achieve the three-phase grid charging mode, the three-phase grid phase-to-phase voltages (VU1-VV1, VV1-VW1, and VW1-VU1) are lower than the battery 1 voltage Vb. The PWM duty ratio of the battery-side three-phase inverter 40 is controlled to adjust the charging current flowing into the battery 1 in the three-phase grid charging mode and the current-uncontrolled DC charging mode. Similarly, the DC voltage applied to the plug 400 is lower than the voltage Vb of the battery 1 in current uncontrolled DC charging mode.

Advantages of On-Board Integrated Charger This on-board integrated charger allows the dual inverter 200 to operate as a PFC circuit in three-phase and single-phase grid charging modes. Moreover, this on-board integrated charger can be connected to both current-controlled DC fast charging spots and non-current-controlled DC fast charging spots. In addition, this on-board integrated charger can have zero resistive loss in the 3-phase coil 50 when connected to a DC fast charging spot. For example, when a current-uncontrolled DC charging scheme is employed, a highly efficient charging of the battery from the solar panel is possible. Moreover, according to this on-board integrated charger, the grid-side three-phase inverter is turned off when a short circuit accident occurs at the DC fast charging spot. This prevents the battery of the DC power supply and the semiconductor switch from being destroyed by the short-circuit current.

Topology of Electric Brake Device 90 FIG. 14 is a block diagram showing the electromagnetic brake device 90 used in the grid charging mode. The controller 9 controls an electric brake device 90 that restrains the wheels 91 . A brake control routine executed by the controller 9 will now be described with reference to FIG. First, it is determined whether or not the battery charging mode has been commanded (S100), and when it is determined that the command has been commanded, the wheels 91 are restrained by the electric braking device 90 (S102). When the battery charging mode is not commanded, it is determined whether or not the 3-phase grid voltage is applied from the plug 400 to the dual inverter 200 (S104), and when it is determined that the 3-phase grid voltage is applied, the electric brake device The wheel 91 is restrained by 90 (S102). When it is determined that the three-phase grid voltage is not applied, the restraint of the wheels 91 by the electric brake device 90 is released (S106).

In the 3-phase grid charging mode, the 3-phase coil 50 generates a rotating magnetic field synchronous with the grid frequency. As a result, the stationary rotor of the three-phase synchronous motor generates acceleration torque in the first half of the cycle period of the grid frequency and deceleration torque in the second half. Consequently, a three-phase synchronous motor produces vibrations that are synchronous to the grid frequency. This vibration is reduced when the electric brake device 90 restrains the wheel 91 .

Topology of Isolated Bidirectional DCDC Converter 300 The isolated bidirectional DCDC converter 300 consists of a primary H-bridge 301 , a transformer 302 and a secondary H-bridge 303 . Details of converter 300 are described with reference to FIGS. The H-bridge 301 consists of a first leg and a second leg. The first leg consists of an upper arm switch 311 and a lower arm switch 312 . The second leg consists of upper arm switch 313 and lower arm switch 314 . H-bridge 303 consists of a third leg and a fourth leg. The third leg consists of upper arm switch 315 and lower arm switch 316 . The fourth leg consists of upper arm switch 317 and lower arm switch 318 .

A transformer 302 with primary coil C1 and secondary coil C2 has a primary side leakage inductance L1 and a secondary side leakage inductance L2. Since the primary coil C1 has much more turns than the secondary coil C2, the leakage inductance L1 is much higher than the leakage inductance L2. H-bridge 301 is connected to primary coil C1 through leakage inductance L1. H-bridge 303 is connected to secondary coil C2 through leakage inductance L2. H-bridge 301 is connected in parallel with smoothing capacitor 13 through DC buses 81 and 82 . H-bridge 303 is connected in parallel with a low voltage battery 304 which is connected to a low voltage electrical load. The switches 311-318 consist of MOS transistors. Converter 300 has a step-down charging mode and a pre-charging mode.

Step-Down Charging Mode In the step-down charging mode for charging the low voltage battery 304, the H-bridge 301 is PWM-controlled and an AC voltage is applied to the primary coil C1. The positive half-cycle period of this alternating voltage is described. Lower arm switch 314 is turned on. Switches 311 and 312 are complementary PWM switched. During the period when the switch 311 is turned on, the primary current flowing into the primary coil C1 through the leakage inductance L1 increases. During the period when switch 311 is turned off, the magnetic energy of leakage inductance L1 causes a freewheeling current to circulate through switch 312, leakage inductance L1, primary coil C1, and lower arm switch 314. FIG. This freewheeling current decays rapidly. A secondary voltage induced in the secondary coil C2 by this change in primary current is rectified by the H bridge 303 and charges the battery 304 . The charging current is controlled by controlling the PWM duty ratio of switch 311 .

The negative half-cycle period of this alternating voltage is described. Lower arm switch 312 is turned on. Switches 313 and 314 are complementary PWM switched. During the period in which the switch 313 is turned on, the primary current flowing into the leakage inductance L1 through the primary coil C1 increases. During the period when switch 313 is turned off, the magnetic energy of leakage inductance L 1 causes a freewheeling current to circulate through switch 314 , primary coil C 1 , leakage inductance L 1 , and lower arm switch 312 . This freewheeling current decays rapidly. A secondary voltage induced in the secondary coil C2 by this change in primary current is rectified by the H bridge 303 and charges the battery 304 . The charging current is controlled by controlling the PWM duty ratio of switch 313 . Ultimately, H-bridge 301 and leakage inductance L1 act as an isolation step-down chopper. As a result, this isolated step-down chopper allows variation of the DC link voltage VC.

It is known that a high inrush current flows into the smoothing capacitor 13 when the precharge mode main relays 61 and 62 are turned on. This precharge mode reduces this inrush current.

In this precharge mode, the two legs of H-bridge 303 are complementary PWM-controlled at a predetermined frequency. This causes an alternating current to flow through the secondary coil C2 and induce a secondary voltage in the primary coil C1. The lower arm switch 312 is PWM controlled during the positive half cycle period of this secondary voltage. When the lower arm switch 312 is turned on, a secondary current circulates through the primary coil C1, the leakage inductance L1, the lower arm switch 312, and the lower arm switch 314, increasing the magnetic energy of the leakage inductance L1. When the lower arm switch 312 is turned off, this magnetic energy supplies a precharge current through the upper arm switch 311, the smoothing capacitor 13, the lower arm switch 314 and the primary coil C1.

During the negative half cycle period of this secondary voltage, the lower arm switch 314 is PWM controlled. When the lower arm switch 314 is turned on, a secondary current circulates through the primary coil C1, the lower arm switch 314, the lower arm switch 312 and the leakage inductance L1, increasing the magnetic energy of the leakage inductance L1. This magnetic energy supplies a precharge current through leakage inductance L1, primary coil C1, upper arm switch 313, smoothing capacitor 13, and lower arm switch 312 when lower arm switch 314 is turned off. Thereby, the smoothing capacitor 13 is precharged. Ultimately, H-bridge 301 and leakage inductance L1 act as an isolation boost chopper.

Advantages of Isolated Bidirectional DCDC Converter 300 This converter 300 can absorb the change of DC link voltage VC during parallel mode, series mode and boost discharge mode in charging low voltage battery 304 . Furthermore, this converter 300 can prevent an inrush current from flowing into smoothing capacitor 13 .

Single Phase Battery Heating Method A single phase battery heating method for rapidly heating the cold batteries 1 and 2 will now be described. Conventional external electric heaters to solve this problem have high power losses. A single-phase battery heating method to solve this problem is described.

FIG. 17 is a block circuit diagram showing the current flow through dual inverter 200. As shown in FIG. A dual inverter 200 connected to the three-phase coil 50 is powered by the battery 1 through DC buses 81 and 82 . The dual inverter 200 consists of a U-phase H bridge 710 , a V-phase H bridge 720 and a W-phase H bridge 730 . Controller 9 commands dual inverter 200 to perform a single-phase battery heating operation under cold conditions in which the battery temperature is less than a predetermined value. In this single-phase current heating, the H bridge 710 applies the U-phase voltage VU to the U-phase coil 5U, and the H bridge 720 applies the U-phase voltage VU to the V-phase coil 5V. The H bridge 730 applies a -U phase voltage (-VU) to the W phase coil 5W. The fundamental frequency components of the U-phase voltage VU and the -U-phase voltage (-VU) are sinusoidal voltages. The -U phase voltage (-VU) has the opposite phase to the U phase voltage VU. As a result, a single-phase AC voltage is applied to the three-phase coil 50 .

A U-phase current IU flows through each of the U-phase coil 5U and the V-phase coil 5V, and a -U-phase current (-IU) flows through the W-phase coil 5W. The -U phase current (-IU) has the opposite phase to the U phase current IU. Therefore, the 3-phase coil 50 forms a single-phase alternating magnetic field within the 3-phase motor. In this embodiment, the single-phase alternating magnetic field is a U-phase alternating magnetic field. A three-phase motor does not generate motor torque due to this single-phase alternating magnetic field. Ultimately, the three-phase coil 50 supplied with single-phase alternating current can essentially be regarded as an inductor. The battery 1 supplies the dual inverter 200 with a U-phase current IU having triple the amplitude.

FIG. 18 shows the waveforms of the fundamental wave components of the U-phase voltage VU and the U-phase current IU applied to the U-phase coil 5U. When the electrical resistance of the three-phase coil 50 is ignored, the phase difference between the U-phase current IU and the U-phase voltage VU is an electrical angle of 90 degrees. The fact that the U-phase alternating current IU, which is a single-phase alternating current, flows through the battery 1 means that the battery 1 repeats charging and discharging periodically. As a result, battery 1 generates heat due to its internal resistance. When the battery 1 is cold, the electrolyte resistance becomes the main resistance component of the battery 1 . Therefore, the electrolyte is rapidly heated and the battery resistance is rapidly reduced.

The dual inverter 200 can supply a non-sinusoidal single-phase AC current to the three-phase coil 50 . However, sinusoidal current is effective in reducing iron loss. The two phase coils (5U, 5V) form a magnetic field in the opposite direction to the phase coil 5W. This prevents the three phase magnetic fields formed by the three phase coils (5U, 5V, and 5W') from canceling each other out in the magnetic circuit of the three-phase motor.

The dual inverter 200 can supply other alternating current components to the three-phase coil 50 in addition to the single phase alternating current. However, since single-phase alternating current is suitable for heating the battery 1 , the single-phase alternating current should be the main component of the current that the dual inverter 200 supplies to the three-phase coil 50 .

This single-phase battery heating method can also be implemented immediately after the start of driving on cold winter mornings. Dual inverter 200 supplies both the torque generating current and the single-phase AC current to three-phase coil 50 after the vehicle starts running. As a result, the battery 1 can be rapidly warmed up without generating excessive motor torque.

Important variants are described. In this variant, the single-phase battery heating method is performed by a single three-phase inverter connected to a conventional star-connected three-phase coil. This single three-phase inverter, consisting of a U-phase leg, a V-phase leg, and a W-phase leg, is connected to a star three-phase coil consisting of a U-phase coil, a V-phase coil, and a W-phase coil.

The U-phase leg applies the U-phase voltage VU to the U-phase coil, the V-phase leg applies the -U-phase voltage (-VU) to the V-phase coil, and the W-phase leg applies the -U-phase voltage (-VU) to the W-phase coil. VU). As a result, double the U-phase voltage is applied to the star-shaped three-phase coil, and double the U-phase current is supplied to the star-connected three-phase coil. The U-phase voltage VU and the -U-phase voltage (-VU) are sinusoidal voltages and have phases opposite to each other. As a result, the U-phase current IU, which is a single-phase alternating current, flows through the battery 1, and the electrolyte of the battery 1 is heated. This single 3-phase inverter is essentially equivalent to U-phase H-bridge 710, and this star 3-phase coil is essentially equivalent to U-phase coil 5U. Ultimately, this variant supplies the battery 1 with single-phase alternating current by means of a conventional single three-phase inverter connected to a star-shaped three-phase coil.

Advantages of single-phase battery heating method Rapid charging of low-temperature batteries accelerates battery deterioration. Therefore, conventional quick chargers cannot perform quick charging when the battery 1 is cold. According to this embodiment, the electrolyte of a cold battery can be warmed up quickly before the start of fast charging. As a result, the total charging time can be shortened. In addition, conventional external electric heaters indirectly heat the battery electrolyte through a plurality of heat-conducting members. Therefore, the temperature rise of the electrolytic solution is delayed. According to this single-phase battery heating method, the electrolyte is directly heated without heat loss, so the electrolyte can be heated quickly.

Multiple Torque Control Methods for Dual Inverter 200 Torque control methods for dual inverter 200 for producing motor torque will now be described. A drawback of the grid-connected on-board integrated charger using the dual inverter 200 as a three-phase PFC circuit as described above is that the dual inverter 200 requires a more complex circuit topology than a typical single three-phase inverter. For this reason, the dual inverter 200 is required to realize a new advantage over this drawback, such as a reduction in inverter loss.

Upper Arm Conduction Single PWM Method The upper arm conduction single PWM method as the first torque control method will be described with reference to FIG. Leg 3U applies U-phase voltage VU1 to phase coil 5U, and leg 4U applies U-phase voltage VU2 to phase coil 5U. A U-phase voltage VU (=VU1-VU2) is applied to the phase coil 5U. Leg 3V applies V-phase voltage VV1 to phase coil 5V, and leg 4V applies U-phase voltage VV2 to phase coil 5V. A V-phase voltage VV (=VV1-VV2) is applied to the phase coil 5V. Leg 3W applies W-phase voltage VW1 to phase coil 5W, and leg 4W applies W-phase voltage VW2 to phase coil 5W. A W-phase voltage VW (=VW1-VW2) is applied to the phase coil 5W. The phase difference between the three phase voltages (VU, VV, and VW) is 120 electrical degrees. Legs 3U and 4U form a U-phase H-bridge that applies phase voltage VU to phase coil 5U. Legs 3V and 4V form a V-phase H-bridge that applies phase voltage VV to phase coil 5V. Legs 3W and 4W form a W-phase H-bridge that applies phase voltage VW to phase coil 5W.

According to this upper arm conducting single PWM method, each H-bridge consists of a PWM leg and a fixed potential leg. A PWM leg is a leg that is PWM controlled. A fixed potential leg is a leg that outputs either of the two DC link potentials. FIG. 19 is a timing chart showing the states of the six legs of the dual inverter 200 in triple H-bridge mode in which the three H-bridges are controlled by the upper arm conductive single PWM method. According to this triple H-bridge mode, the three phase voltage vectors applied to the three-phase coils 50 form a combined rotating voltage vector for forming a rotating magnetic field within the three-phase motor.

The three H-bridges of dual inverter 200 have essentially the same operation. For this reason only the PWM control of the U-phase H-bridge consisting of legs 3U and 4U will be described. FIG. 20 shows U-phase voltages VU1 and VU2. The U-phase voltage VU1 is the fundamental frequency component of the output voltage of leg 3U, and the U-phase voltage VU2 is the fundamental frequency component of the output voltage of leg 4U. As a result, the U-phase voltage VU (=VU1-VU2) is applied to the U-phase coil 5U. One period of the U-phase voltage VU is divided into a positive half-wave period PA and a negative half-wave period PB each equal to an electrical angle of 180 degrees. According to this upper arm conduction type single PWM method, in the positive half-wave period PA, leg 4U becomes a PWM leg, and leg 3U becomes a fixed potential leg that outputs a high level DC link voltage (1). In the negative half-wave period PB, the leg 3U becomes the PWM leg, and the leg 4U becomes the fixed potential leg that outputs the high level DC link voltage (1).

In other words, according to the upper arm conduction single PWM method, the upper arm switch of the fixed potential leg is turned on steadily. The fixed potential leg and the PWM leg alternate every 180 electrical degrees. When the lower arm switch of the PWM leg is turned on, the PWM leg outputs a low level DC link voltage (0), and when the upper arm switch of the PWM leg is turned on, the PWM leg outputs a high level DC link voltage (1). to output

Advantages of upper arm conduction single PWM method First, the fixed potential leg does not generate switching loss. For this reason, the upper arm conduction single PWM method has half the switching losses compared to the conventional double PWM method in which the two legs of the H-bridge are PWM legs. Then the upper arm switch of the fixed potential leg is always turned on. Therefore, the ringing surge voltage caused by turning off the upper arm transistor of the inverter can be reduced. As a result, switching losses can be reduced by shortening the switching transient period.

This upper arm conduction single PWM method and the conventional double PWM method are compared with reference to FIGS. 21-24. 21-24 show the phase period during which the U-phase current IU flows through the U-phase coil 5U. Figures 21 and 22 show a conventional double PWM method in which legs 3U and 4U are complementary PWM controlled. Figures 23 and 24 show the upper arm conduction single PWM method where only leg 4U is PWM controlled. The upper and lower arm switches of the PWM leg are switched complementarily.

In FIG. 21, switches 31 and 42 are turned on. U-phase current IU consists of power supply current IP flowing from battery 1 to phase coil 5U. In FIG. 22, switches 32 and 41 are turned on. The U-phase current IU becomes a regenerative current IR that flows from the phase coil 5U to the battery 1. According to this double PWM method, the U-phase voltage VU applied to the U-phase coil 5U is VU1-VU2, and the U-phase current IU is IP-IR. The regenerative current IR returning to batteries 1 and 2 during each PWM cycle increases the high frequency resistive losses of battery 1 .

In FIG. 23, which is the same as FIG. 21, switches 31 and 42 are turned on. U-phase current IU consists of power supply current IP flowing from battery 1 to phase coil 5U. In FIG. 24, switches 31 and 41 are turned on. The U-phase current IU becomes a freewheeling current If that circulates through the phase coil 5U, the switch 41, and the switch 31. According to this upper arm conduction single PWM method, the U-phase voltage VU applied to the U-phase coil 5U is VU1-VU2, and the U-phase current IU is IP+If. Regenerative current IR is prevented from returning to battery 1 . As a result, the loss of battery 1 is reduced.

The top arm conduction single PWM method is further described with reference to FIGS. 25-29. FIG. 25 shows fundamental frequency components of U-phase voltage VU and U-phase current IU. A U-phase voltage VU is a phase voltage applied to the U-phase coil 5U, and a U-phase current IU is a phase current flowing through the U-phase coil 5U. One cycle period corresponding to an electrical angle of 360 degrees is divided into a positive half-wave period PA and a negative half-wave period PB. The amplitude value of the U-phase voltage VU is positive during the period PA and negative during the period PB.

The leg 3U becomes a fixed potential leg during the period PA and becomes a PWM leg during the period PB. The leg 4U becomes the PWM leg during the period PA and becomes the fixed potential leg during the period PB. The upper arm switch of the fixed potential leg is turned on and the lower arm switch of the fixed potential leg is turned off. The U-phase current IU lags the U-phase voltage VU by a predetermined phase period due to the inductance of the U-phase coil 5U. Period PA is divided into phase periods P1 and P2, and period PB is divided into phase periods P3 and P4. In phase periods P1 and P3, U-phase voltage VU has an opposite direction to U-phase current IU. In phase periods P2 and P4, U-phase voltage VU has the same direction as U-phase current IU.

FIG. 26 shows phase period P1. When the upper arm switch 41 of the PWM leg 4U is turned on, a freewheeling current If flows, and when the upper arm switch 41 is turned off, the regenerated current IP charges the battery 1 . A ringing voltage is generated due to the inductance of the high level internal DC bus 810 connecting the two upper arm switches 31 and 41 when the upper arm switch 41 is turned off. However, this high level internal DC bus 810 has a lower inductance value than the high level DC bus 81 . As a result, this ringing voltage is reduced. FIG. 27 shows phase period P2. When the lower arm switch 42 of the PWM leg 4U is turned on, the power supply current IP is supplied to the U phase coil 5U, and when the lower arm switch 42 is turned off, the freewheeling current If circulates through the U phase coil 5U. .

FIG. 28 shows phase period P3. When the upper arm switch 31 of the PWM leg 3U is turned on, a freewheeling current If flows. When the upper arm switch 31 is turned off, a ringing voltage is generated due to the inductance of the high level internal DC bus 810 . However, high level DC internal DC bus 810 has a lower inductance value than high level DC bus 81 . As a result, this ringing voltage is reduced. FIG. 29 shows phase period P4. When the lower arm switch 32 of the PWM leg 3U is turned on, the power supply current IP is supplied to the U phase coil 5U, and when the lower arm switch 32 is turned off, the freewheeling current If circulates through the U phase coil 5U. .

Double H-Bridge Mode The double H-bridge mode as the second torque control method is described with reference to FIGS. 30 and 31. FIG. According to this double H-bridge mode, one of the three H-bridges of dual inverter 200 is deactivated in turn every 60 electrical degrees. A dormant H-bridge is called a dormant H-bridge. Each upper arm switch and lower arm switch of the two legs of the dormant H-bridge are turned off. This double H-bridge mode, which deactivates one of the three H-bridges, reduces dual inverter 200 losses.

Dual inverter 200 applies a resultant rotating voltage vector to three-phase coil 50 to generate motor torque. This resultant rotating voltage vector forms a resultant rotating current vector, and this resultant rotating current vector forms a rotating magnetic field vector. The combined rotational voltage vector is a vector of a U-phase voltage vector VU applied to the U-phase coil 5U, a V-phase voltage vector VV applied to the V-phase coil 5V, and a W-phase voltage vector VW applied to the W-phase coil 5W. It is harmony.

In double H-bridge mode, the resultant rotating voltage vector is formed by the vector sum of the phase voltage vectors of the two adjacent phases. The double H-bridge mode is adopted in the low voltage operating region where the maximum amplitude of the combined rotating voltage vector is less than or equal to the maximum amplitude of each phase voltage vector. Therefore, the double H-bridge mode is suitable for a medium-low speed range of a three-phase motor in which the back electromotive force Vemf of the three-phase coil 50 is relatively low. In the double H-bridge mode, one of the three H-bridges is deactivated in turn every 60 electrical degrees and the remaining two H-bridges are PWM controlled.

FIG. 30 is a vector diagram for showing the existence region of the composite rotation voltage vector in the double H-bridge mode. The region of presence of the resultant rotating voltage vector is divided into seven phase regions Z0-Z6. In the phase region Z1, the combined rotating voltage vector is the vector sum of the U-phase voltage vector (VU) and the -W-phase voltage vector (-VW). The V-phase H-bridge is quiesced. In the phase region Z2, the combined rotational voltage vector is the vector sum of the V-phase voltage vector (VV) and the -W-phase voltage vector (-VW). The U-phase H-bridge is deactivated.

In the phase region Z3, the combined rotational voltage vector is the vector sum of the V-phase voltage vector (VV) and the -U-phase voltage vector (-VU). The W-phase H-bridge is deactivated. In the phase region Z4, the combined rotational voltage vector is the vector sum of the W-phase voltage vector (VW) and the -U-phase voltage vector (-VU). The V-phase H-bridge is quiesced. In the phase region Z5, the combined rotational voltage vector is the vector sum of the W-phase voltage vector (VW) and the -V-phase voltage vector (-VV). The U-phase H-bridge is deactivated. In the phase region Z6, the combined rotational voltage vector is the vector sum of the U-phase voltage vector (VU) and the -V-phase voltage vector (-VV). The W-phase H-bridge is deactivated.

A U-phase H-bridge consisting of legs 3U and 4U outputs phase voltage vectors VU and -VU. A V-phase H-bridge consisting of legs 3V and 4V outputs phase voltage vectors VV and -VV. A W-phase H-bridge consisting of legs 3W and 4W outputs phase voltage vectors VW and -VW. In FIG. 30, six phase voltage vectors (VU, -VW, VV, -VU, VW, -VV) each have a maximum amplitude value.

The maximum amplitude of the resultant rotating voltage vector, equal to the radius of the inner circle indicated by the dashed line in FIG. 30, is equal to the maximum amplitude of one phase voltage vector in this double H-bridge mode. Therefore, the double H-bridge mode is executed inside this inner circle. A triple H-bridge mode in which each of the three H-bridges is PWM-controlled is executed in the phase region Z0 between this inner circle and the outer circle indicated by the solid line.

In triple H-bridge mode, each of the three H-bridges forms a phase voltage vector, and the vector sum of the three phase voltage vectors is the resultant rotating voltage vector. Disabling PWM control of the dormant H-bridge reduces battery loss. Switching between each phase region can be performed gradual to suppress abrupt changes in phase current. In other words, the triple H-bridge mode may be briefly performed during the transition period when switching between each phase domain is performed.

FIG. 31 is a timing chart showing this double H bridge mode. Each of the three H-bridges is controlled by a single PWM with upper arm conduction. The U-phase H-bridge is quiescent during quiescent periods of 330-30 electrical degrees and 150-210 electrical degrees. The V-phase H-bridge is quiescent during quiescent periods of 90-150 electrical degrees and 270-330 electrical degrees. The W-phase H-bridge is quiescent during quiescent periods of 30-90 electrical degrees and 210-270 electrical degrees. Finally, according to the double H-bridge mode, the H-bridge that outputs the phase voltage with the smallest amplitude becomes the dormant H-bridge. The upper and lower arm switches of the dormant H-bridge are turned off and the boost switched battery 100 does not supply power current to the dormant H-bridge.

Boost Double H-Bridge Mode The boost double H-bridge mode as the third torque control method will be described with reference to FIGS. 32 and 33. FIG. In this boost double H-bridge mode, the PWM switching of the fixed potential H-bridge is stopped. A potential-fixed H-bridge outputs phase voltages of maximum amplitude. This double H-bridge mode, which deactivates one of the three H-bridges, reduces dual inverter 200 losses.

Boost switched battery 100 shown in FIG. 1 applies a boosted DC link voltage VC to dual inverter 200 . The U-phase H bridge of dual inverter 200 applies U-phase voltage VU to U-phase coil 5U. The V-phase H bridge of dual inverter 200 applies V-phase voltage VV to V-phase coil 5U. The W-phase H bridge of the dual inverter 200 applies the W-phase voltage VW to the W-phase coil VW. The two legs of the potential fixed H-bridge outputting the phase voltages of maximum amplitude are both fixed potential legs. These two fixed potential legs output opposite voltages. One of the two fixed potential legs outputs a high level DC link voltage and the other outputs a low level DC link voltage. The output potentials of the two fixed potential legs alternate every 180 electrical degrees.

FIG. 32 is a timing chart showing fundamental frequency components of three phase voltages VU, VV, and VW. In the U-phase fixed phase period (electrical angles of 60 degrees to 120 degrees and 240 degrees to 300 degrees) in which the amplitude of the U-phase voltage VU is maximum, the U-phase H bridge becomes a potential fixed H bridge. During the V-phase fixed phase period (electrical angle 180 degrees to 240 degrees and 0 degrees to 60 degrees) in which the amplitude of the V-phase voltage VV is maximum, the V-phase H bridge becomes a potential fixed H bridge. The W-phase H bridge becomes a potential-fixed H bridge in the W-phase potential-fixed phase period (300-360 electrical degrees and 120-180 degrees electrical angle) in which the amplitude of the W-phase voltage is maximum. The length of the fixed phase period of each phase arranged in order is 60 electrical degrees.

The upper arm switch 31 and the lower arm switch 42 are turned on during the U-phase fixed phase period in which the U-phase voltage VU has a positive value. The upper arm switch 41 and the lower arm switch 32 are turned on during the U-phase fixed potential period in which the U-phase voltage VU has a negative value. During the V-phase fixed phase period in which the V-phase voltage VV has a positive value, the upper arm switch 33 and the lower arm switch 44 are turned on.

During the V-phase fixed potential period in which the V-phase voltage VV has a negative value, the upper arm switch 43 and the lower arm switch 34 are turned on. During the W-phase fixed phase period in which the W-phase voltage VW has a positive value, the upper arm switch 35 and the lower arm switch 46 are turned on. During the W-phase fixed potential period in which the W-phase voltage VW has a negative value, the upper arm switch 45 and the lower arm switch 36 are turned on.

Further, the front end boost converter 10 boosts the battery voltage. The boosted DC link voltage VC is equal to the U-phase voltage VU during the U-phase fixed-phase period, equal to the V-phase voltage VV during the V-phase fixed-phase period, and equal to the W-phase voltage VW during the W-phase fixed-phase period. . Ultimately, front-end boost converter 10 applies a DC link voltage VC having a waveform equal to the rectified three-phase sinusoidal voltage to dual inverter 200 . In other words, the front end boost converter 10 is PWM controlled instead of a potential fixed H-bridge. The DC link voltage VC has ripples. Therefore, the PWM duty ratios of the other two H-bridges, excluding the fixed potential H-bridge, are corrected to eliminate the effect of this DC link voltage VC ripple.

FIG. 33 is a timing chart showing the states of three H-bridges. In the phase periods P1 and P4, the U-phase H-bridge becomes a potential fixed H-bridge. In phase periods P2 and P5, the V-phase H bridge becomes a potential fixed H bridge. In phase periods P3 and P6, the W-phase H bridge becomes a potential fixed H bridge. The other two H-bridges, except the voltage fixed H-bridge, are operated with the upper arm conduction single PWM method already described. This boost double H-bridge mode using the boost operation of converter 10 is executed in a high-speed operation region where the output voltage of the potential-fixed H-bridge is higher than the battery voltage.

Single H-Bridge Mode The single H-bridge mode as a fourth torque control method is described with reference to FIG. In this single H-bridge mode, the double H-bridge mode and boost double H-bridge mode are executed together. FIG. 34 is a timing chart showing the state of each leg of dual inverter 200 driven in single H-bridge mode. In FIG. 33, a high level period 'H' is a phase period during which each leg outputs a high level potential, and a low level period 'L' is a phase period during which each leg outputs a low level potential. During idle period 'A', both the upper arm switch and the lower arm switch of each leg are turned off. A phase period 'PWM' is a period in which each leg is PWM-controlled. Each H-bridge is controlled by the upper arm conduction single PWM method in each phase period 'PWM'.

In this single H-bridge mode, the boost switched battery 100 applies a DC link voltage VC equal to the full amplitude phase voltage across the potential clamp H-bridge. In addition, a double H-bridge mode is implemented, in which the dormant H-bridge that should output the phase voltages of minimum amplitude is quiesced. Ultimately, only one of the three H-bridges and converter 10 is PWM controlled. This reduces switching losses in a three-phase motor drive.

Current Spreading Method The current spreading method as the fifth torque control method will be described with reference to FIGS. 35 and 36. FIG. According to this current distribution method, dual inverter 200 is controlled by a space vector PWM method (SVPWM method). This current spreading method reduces battery losses. The three H-bridges of dual inverter 200 are PWM controlled based on a PWM carrier signal with a predetermined frequency value. For example, when the frequency of the PWM carrier signal is 20 kHz, the common PWM cycle period TC has a length of 50 µs.

According to the SVPWM method, each H-bridge has a current supply period and a freewheeling period that are placed within a common PWM cycle period TC. This current supply period is a period during which the boost switched battery 100 supplies power supply current to the H bridge. The freewheeling period is the period during which the freewheeling current circulates between the H-bridge and the phase coils. The U-phase voltage vector VU is approximately proportional to the length of the current supply period of the U-phase H bridge, the V-phase voltage vector VV is approximately proportional to the length of the current supply period of the V-phase H-bridge, and the W-phase voltage vector VW is Almost proportional to the length of the current supply period of the W-phase H-bridge.

An important feature of dual inverter 200 is that each H-bridge can be independently PWM controlled. In other words, when the SVPWM method is adopted, the current supply periods of each phase can be arranged freely relative to each other within the common PWM cycle period TC. In this current distribution method, the current supply periods of each phase are arranged within a common PWM cycle period TC so as not to overlap each other as much as possible. Thereby, the loss of boost switched battery 100 can be reduced.

FIG. 35 is a timing chart showing one PWM cycle period TC in triple H bridge mode in which three H bridges are PWM controlled. A current supply period TXV for the V-phase H bridge, a current supply period TXW for the W-phase H bridge, and a current supply period TXU for the U-phase H bridge are arranged in order within a common PWM cycle period TC. This essentially avoids overlapping of the three current supply periods TXV, TXW, TXU. The boost switched battery 100 supplies the U-phase power supply current IUP to the U-phase H bridge during the period TXU, supplies the V-phase power supply current IVP to the V-phase H bridge during the period TXV, and supplies the W-phase power supply current IVP to the W-phase H bridge during the period TXW. Provides phase power current IWP. Therefore, the overlap of the three phase power supply currents IUP, IVP, and IWP is minimized and resistive losses of boost switched battery 100 are reduced.

Furthermore, the current supply period for each phase is arranged continuously. As a result, the high-frequency current component contained in the power supply current supplied from DC power supply 100 to dual inverter 200 is reduced. Furthermore, the longest current supply period TXW is sandwiched by two other current supply periods TXV and TXU. As a result, the high frequency current component flowing through boost switched battery 100 is reduced. Furthermore, two adjacent current supply periods overlap each other by a short transition period Tt. As a result, the high frequency current component flowing through boost switched battery 100 is reduced.

FIG. 36 is a timing chart showing one PWM cycle period TC in double H-bridge mode or boost double H-bridge mode. The resistive loss of boost switched battery 100 is reduced. Furthermore, the shorter current supply period is placed immediately after the longer current supply period. This suppresses a rapid decrease in the power supply current and reduces the ringing surge voltage. Ultimately, the current spreading method reduces resistive losses in boost switched battery 100 .

Switching Torque Control Modes Each of the torque control methods described above is preferably switched in accordance with the motor speed. It is also possible to select the optimum torque control method according to other parameters such as torque command value and efficiency. For example, the current distribution method is very effective in low speed regions where the back electromotive force (back EMF) of the three-phase coil 50 is low. The PWM duty ratio of each phase is reduced by this low back EMF. Therefore, the current supply period of each phase can be arranged within each PWM cycle period without overlapping each other.

Modified Embodiment When each of the boost switched battery 100, the dual inverter 200, and the isolated bidirectional DCDC converter 300, which are the main elements of the power switching circuit of the present invention, is used separately, the merits specific to each element are realized. can be done.

Claims (17)

A three-phase inverter connected to three-phase coils of a three- phase motor , a DC power supply including a battery connected to the three-phase inverter through a DC bus, and one end of each of the three phase coils of the three-phase coils connected to an external power supply. In a power switching circuit comprising a connecting plug and a controller for controlling the three-phase inverter,
The three-phase inverter is a common inverter composed of a grid-side three-phase inverter (30) and a battery-side three-phase inverter (40) connected to the grid-side three-phase inverter (30) through an open-end three-phase coil (50). consists of a DC bus based dual inverter (200),
The controller (9) operates in a three-phase grid charging mode for driving the three-phase coil (50) and the battery-side three-phase inverter (40) as a three-phase boost rectifier, and the grid-side three-phase inverter (40) as a three-phase rectifier. 30) and a DC charging mode to drive
The three-phase boost rectifier converts a three-phase grid voltage applied from the external power supply through the plug (400) into a boost rectified voltage for application to the DC bus;
A power switching circuit, wherein the three-phase rectifier converts a DC voltage applied from the external power supply through the plug (400) into a rectified voltage to be applied between the DC buses (81, 82). . The power switching circuit of claim 1, wherein said controller (9) has a 3-phase PFC mode for driving said battery-side 3-phase inverter (40) as a 3-phase phase correction (PFC) circuit in said 3-phase grid charging mode. The controller (9) controls, in the three-phase PFC mode, a step-up phase period in which the battery-side three-phase inverter (40) is driven as a step-up chopper and a step-down phase period in which the battery-side three-phase inverter (40) is driven as a step-down chopper. 3. The power switching circuit of claim 2, having a phase period. The power switching circuit of claim 1, wherein said DC power supply comprises a front end boost converter (10) for stepping down a DC link voltage (VC) applied across said DC bus (81, 82) in said three-phase grid charging mode. . The power of claim 4, wherein the front-end boost converter (10) steps down the DC link voltage (VC) for charging current control in the DC charging mode when the external power supply is a non-current controlled DC power supply. switching circuit. 5. The front end boost converter (10) according to claim 4, wherein said front end boost converter (10) stops stepping down said DC link voltage (VC) when said three phase grid voltage is lower than said DC link voltage (VC) in said three phase grid charging mode. power switching circuit. The front end boost converter (10) has a series transistor (3), two parallel transistors (4, 5), an output transistor (6), a reactor (7) and a surge absorption diode ,
The series transistor (3) and the reactor (7) connected in series connect the positive electrode of the battery (1) on the low side to the negative electrode of the battery (2) on the high side,
the two parallel transistors (4, 5) connect the two batteries (1, 2) in parallel;
the output transistor (6) connects the two batteries (1, 2) to the inverter (200);
5. The power switching circuit according to claim 4, wherein the surge absorption diode connects the reactor (7) to at least one of the negative electrode of the battery (1) on the low side and the positive electrode of the battery (2) on the high side. Said controller (9) controls said two batteries (1, 2) before connecting said two batteries (1, 2) in parallel if the voltage difference between said two batteries (1, 2) exceeds a predetermined value. 8. The power switching circuit of claim 7, having a voltage balancing mode to reduce the voltage difference between , 2). said DC bus (81, 82) is connected to a low voltage battery (304) through an isolated bi-directional DCDC converter (300);
Said converter (300) comprises a transformer (302) having a primary coil (C1) and a secondary coil (C2), and a primary side H-1 connecting said primary coil (C1) to said DC bus (81, 82). a bridge (301) and a secondary side H-bridge (303) connecting said secondary coil (C2) to said low voltage battery (304);
The controller (9) has a step-down charging mode for charging the low voltage battery (304) and a pre-charging mode for charging a smoothing capacitor (13) connected to the DC bus (81, 82). 2. A power switching circuit according to claim 1. 10. A power switching circuit according to claim 9, wherein the primary side H-bridge (301) and the leakage inductance (L1) of the primary coil (C1) are substantially driven as a step-down chopper reactor in the step-down charging mode. 10. The power switching circuit of claim 9, wherein the primary side H-bridge (301) and the leakage inductance (L1) of the primary coil (C1) are substantially driven as a boost chopper reactor in the precharge mode. Power switching according to claim 1, wherein said controller (9) has a single-phase battery heating mode for supplying single-phase alternating current from said battery to said three-phase coil (50) when the temperature of said battery is below a predetermined value. circuit. Each of the three H-bridges of the dual inverter (200) consists of a PWM-controlled PWM leg and a fixed potential leg that outputs either a high-level DC voltage or a low-level DC voltage,
The controller (9) controls the motor torque by the upper arm conduction single PWM mode,
the fixed potential leg and the PWM leg of the H bridge are alternated every 180 electrical degrees in the upper arm conduction single PWM mode;
2. The power switching circuit according to claim 1, wherein the upper arm transistor of said fixed potential leg is always turned on in said upper arm conduction single PWM mode. Power switching according to claim 1, wherein the controller (9) controls the motor torque by a double H-bridge mode in which one of the three H-bridges of the dual inverter (200) is deactivated in turn every 60 electrical degrees. circuit. one of the three H-bridges of the dual inverter (200) consists of a fixed potential H-bridge that constantly outputs the DC link voltage (VC);
The controller (9) operates in a boost double H-bridge mode in which the DC link voltage (VC) formed by the boosting operation of the front-end boost converter (10) is applied to the fixed potential H-bridge. control torque,
5. The fixed potential H-bridge according to claim 4, wherein in the boost double H-bridge mode, the boosted voltage is applied to one phase coil (5U or 5V or 5W) of the three-phase coil (50) without PWM control. power switching circuit. The three H-bridges of the dual inverter (200) are connected to the three phase coils (50) during current supply periods (TXU, TXV, and TXW) of each phase formed within a common PWM cycle period (TC). Separately supply each phase power supply current (IUP, IVP, IWP) to three phase coils (5U, 5V, and 5W),
When the sum of the phase current supply periods (TXU, TXV, TXW) is shorter than a predetermined time, the controller (9) controls the motor torque by a current distribution method that does not temporally overlap at least two of the phase current supply periods. 2. The power switching circuit of claim 1, wherein the power switching circuit controls the In a power switching circuit comprising a controller that controls an inverter that converts DC power supplied from a DC power supply through a DC bus into three-phase AC power and applies it to three-phase coils of a three-phase motor,
The controller (9) supplies a single-phase alternating current from the batteries (1, 2) to the three-phase coil (50) when the temperature of the batteries (1, 2) of the DC power supply is lower than a predetermined value. A power switching circuit characterized by having a phase battery heating mode.

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